4. TEST CASE

Fourth, if we want to compute the electric field E៮ only on a certain spatial window of the entire xy-domain, whether centered or not, we can easily modify the convolution algorithm by providing as input the function E៮ 1 on the desired xy-window, as long as a sufficient decay of E៮ 1 is ensured. The Green’s function can also be considered on a window smaller than the entire xy-domain, provided that such a window is centered at the origin and a decay of at least 60 dB from the peak is ensured. By doing that, the computational time for the convolution drastically diminishes. Fifth, it is noted that the singularity extraction technique provides the correct aperture field also for nonsingular spectral components and the technique can thus be applied to all components of the PWS without a priori considerations on the absence or presence of the singularity.

5.

6. 7. 8.

9.

4. TEST CASE

To illustrate the effect of the singularity extraction technique presented in Section 3, we investigate here an array of 5 y-oriented electric Hertzian dipoles displaced along the x-axis at a distance of 2␭ from each others, see Figure 2. The excitations of the five dipoles are P, P/2, P/5, P/8, and P/10, respectively, with P being the dipole moment of the dipole at the origin. The exact PWS is first computed from Eq. (2) on the [⫺2k : 2k] kxky-domain with 91 sampling points in both directions, see Figure 3. The PWS clearly shows the singularity in both the x- and y-components. In the computation of the aperture field from this PWS, only the visible region is taken into account since this is most often the case in practice, while the invisible region is zero-padded. Figure 4a shows the analytical x- and y-components of the electric field on the z ⫽ 0.1␭ plane while Figure 4(b) shows the result of a straightforward IFFT of the singular PWS without the use of the singularity extraction technique. It is evident that while all five dipoles are seen in Figure 4(a), only the first two are clearly distinguished in Figure 4(b). The last three dipoles, having a weaker excitation, can not be correctly detected since the singularity is not properly taken into account. Figure 4(c) then shows the result of applying the singularity extraction technique. In this case all five dipoles are clearly detected and the difference in their excitations can also be seen. The slightly wider extensions of these compared to the analytical result is due to the truncation of the PWS to the visible region and the truncation of the two functions involved in the convolution on the finite xy-plane. 5. CONCLUSIONS

An effective technique to extract the singularity of plane wave spectra in the computation of antenna aperture fields has been presented. The algorithm is based on the Inverse Fast Fourier Transform and Weyl-identity and allows the accurate computation of the aperture field when a dense sampling in the spectral domain is not possible. The detection of sources of very weak amplitude has been verified by a numerical example and the evident advantages compared to the Inverse Fast Fourier Transform of the PWS without the singularity extraction have been underlined. REFERENCES 1. A.J. Devaney, A filtered backpropagation algorithm for diffraction tomography, Ultrason Imag 4 (1982), 336 –360. 2. T.B. Hansen and P.M. Johansen, Inversion scheme for ground penetrating radar that takes into account the planar air-soil interface, IEEE Trans Geosci Remote Sens 38(2000), 496 –506. 3. J.J.H. Wang, An examination of the theory and practices of planar near/field measurement, IEEE Trans Antenna Propagat 36 (1988), 746 –753. 4. T.B. Hansen and A.D. Yaghjian, Plane wave theory of time-domain

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fields, near-field scanning applications, IEEE Press, New York, NY, 1999. A.D. Yaghjian, Upper-bound errors in far-field antenna parameters determined from planar near-field measurements, Part I: Analysis, Nat Bur Stand Tech Note 667, Boulder, CO, (1975). A.D. Yaghjian, Electric dyadic Green’s functions in the source region, Proc IEEE 68 (1980), 248 –263. M.G. Duffy, Quadrature over a pyramid or cube of integrands with a singularity at a vertex, SIAM J Numer Anal 19 (1982), 1260 –1262. C. Cappellin, A. Frandsen, and O. Breinbjerg, Application of the SWE-to-PWE antenna diagnostics technique to an offset reflector antenna, AMTA Antenna Measurements Technique Association Symposium, St. Louis, USA, November 2007. C. Cappellin, A. Frandsen, S. Pivnenko, G. Lemanczyk, and O. Breinbjerg, Diagnostics of the SMOS radiometer antenna system at the DTU-ESA spherical near-field antenna test facility, EuCAP European Conference on Antennas and Propagation, Edinburgh, UK, November 2007. A. Papoulis, Signal analysis, McGraw-Hill, 1977.

© 2008 Wiley Periodicals, Inc.

DESIGN AND ANALYSIS OF WIDEBAND DOUBLE-SIDED PRINTED SPIRALDIPOLE ANTENNA WITH CAPACITIVE COUPLING Hatem Rmili1 and Jean-Marie Floc’h2 1 ISSAT, Sidi Messaoud, 5111 Mahdia, Tunisie; Corresponding author: [email protected] 2 IETR, UMR CNRS 6164, INSA, 20 avenue Buttes des Coe¨smes 35043 Rennes, France Received 8 October 2007 ABSTRACT: A compact and wideband printed dipole antenna is presented for C- and X-applications. The proposed balun-fed antenna consists of a double-sided printed dipole structure with a microstrip-fed modified spiral-dipole printed on the top side of the substrate, and a capacitive-coupling rectangular-patch, a connecting strip line and the ground plane printed on the bottom side. For the proposed antenna, a wide resonant band which extends from 6.27 to 12.47 GHz, with a fractional bandwidth of 66%, is observed. Details of the experimental and simulation results are presented and discussed. © 2008 Wiley Periodicals, Inc. Microwave Opt Technol Lett 50: 1312–1317, 2008; Published online in Wiley InterScience (www.interscience.wiley. com). DOI 10.1002/mop.23341 Key words: printed spiral; dipole antennas; wideband; capacitive coupling 1. INTRODUCTION

Microstrip or patch antennas are widely used for microwave applications in a market driven by device miniaturization, as they are small and easily fabricated. In addition to their well-known advantages (low profile, low cost of fabrication, easy integration into planar arrays, light weight, and compatibility with microwave integrated-circuit technologies), broadening frequency bandwidth and sharing multifrequency bands are also desirables properties, especially with the rapid progress in wireless communication. In particular, communication systems that operate in the C and X-bands are normally designed using separate antennas for each band. Since it is becoming more and more important to use such systems in one setting, it is desirable to design a single antenna that operates in both frequency bands [1]. To comply with this require-

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In this letter, a compact and wideband double-sided printed antenna with spiral configuration is designed to operate in both C and X-bands. The antenna was fabricated and characterized by determining the impedance matching condition, the impedance bandwidth, and the radiation patterns. The compact size of the antenna was obtained by optimizing the spiral length, by using a circular capacitive loading (central disk), and by using the doublesided structure, whereas the broadband behavior was achieved by a capacitive coupling (a rectangular patch is connected to the ground plane), and by using a modified-CPW feeding technique. It is shown that, for a fixed spiral length, the impedance bandwidth can be tuned especially by the width of the rectangular couplingpatch. 2. ANTENNA DESIGN

Figure 1 Printed dipole antenna (Dimensions in mm: LSub ⫽ 50, WSub ⫽ 40, L ⫽ 35, W ⫽ 12, L1 ⫽ 15.5, W1 ⫽ 6, L2 ⫽ 15, W2 ⫽ 1, W3 ⫽ 5, W4 ⫽ 3, D ⫽ 5)

ment compact high-performance broadband/multiband planar antennas are needed. Recently, various types of printed antennas have been studied to meet the increasing trend for wideband antennas [2, 3], and several techniques for size reduction and bandwidth enhancement have been proposed [4 –7]. For the size reduction [4, 5, 7], the main miniaturizing techniques used are loading the antenna with lumped elements, highdielectric materials, or with conductors; using ground planes and short circuits; optimizing the geometry; and using the antenna environment to reinforce the radiation; whereas, several techniques [4, 6, 7], such as using substrate of low permittivity, or of high thickness; mounting reactive loading; use of coupled resonators, and parasitic elements; feeding from rear slot, or feeding by coplanar feeding line, have been developed as bandwidth enhancement techniques. Here, both miniaturization and bandwidth enhancement techniques are combined in order to realize a novel compact and broadband antenna. Miniaturization is provided by the optimization of the antenna geometry, and bandwidth enhancement is obtained by using a capacitive coupling. One of the effective techniques to reduce the patch size and to increase the impedance bandwidth of microstrip antennas is the use of spiral antennas [8]. Spiral antennas have an easy impedance matching, a superior radiation efficiency, an ultra-wideband characteristic and radiate circularly polarized waves, thereby making them good candidates for mobile and satellite communications [9].

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The proposed antenna was etched on Fr4 substrate with thickness h ⫽ 0.8 mm and permittivity ␧r ⫽ 4.4. The design parameters of the optimized structure are summarized in the caption of Figure 1. The dimension of the antenna is 50 by 40 mm2. The antenna is fed by a balun. The printed patch on the top side of the substrate (Wsub ⫻ Lsub), is obtained from a circular-spiral, with an inner radius L2 and a width W2, by adding a central disk of diameter D as shown in Figure 1. This modified-spiral patch is fed by a 50-⍀ strip line of width W2. On the bottom side of the substrate, a coupling rectangular-patch of dimensions W ⫻ L, a rectangular ground plane of dimensions W3 ⫻ Lsub with a rectangular slot of width W4, and a connecting stripline of dimensions (L1 ⫻ W1) were printed. The trace equation of the spiral is r ⫽ r0⌽ ⫹ r1, where r1 is the inner radius, ⌽ the associated angle and r0 the growth rate determined from the spiral width W2 and the spacing s between each turn, r0 ⫽ (s ⫹ W2)/␲. In our case, with the design parameters W2 ⫽ 1 mm, s ⫽ 1 mm, r1 ⫽ D/2 ⫽ 2.5 mm and n ⫽ 6.8 (n is the number of turns), the trace equation is r ⫽ 2/␲ ⌽ ⫹ 2.5 (mm) with ⑀[0, 13.6␲]. Consequently, the spiral length is 379.3 mm, and by adding the microstrip feed line length 20.5 mm, we get the total length of the printed dipole on the top side of the substrate, which is about 400 mm.

Figure 2 return loss

Effect of the capacitive coupling element on the measured

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Figure 3 Measured return loss for the proposed antenna with different rectangular widths W

During the design procedure, the shape (rectangle or disk) and the position of the coupling patch (when it covers the inferior half, the superior half or the total area of the spiral) were studied numerically and experimentally. It is found that the ⫺10-dB impedance bandwidth depends slightly with the shape of the coupling patch and highly with it’s position. In fact, when the rectangular patch covers the superior half of the spiral as proposed in this article, the excited surfacic currents in this region are clearly higher than the two other cases. From the surfacic current analysis, it seems that the resonating wideband results on the overlap of several resonant bands associated each to the arc-dipole situated in the covered region by the coupling patch. The structure was optimized in order to design a compact antenna with a small substrate operating in both C- and X-bands. First, Ansoft-HFSS simulations were performed on the structure

Figure 4 Measured return loss for the proposed antenna with different stripline widths W1

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Figure 5 Prototype of the proposed antenna. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley. com]

by optimizing, especially, the spiral-length, and the radius of the central disk. Next, a coupling element (a rectangular patch (W ⫻ L) connected to the ground plane with a strip line (L1 ⫻ W1)) is added. Finally, the effect of the parameters design on the antenna return loss was studied. Figure 2 shows the effect of the capacitive coupling on the return loss. As it is shown, the ⫺10-dB impedance bandwidth is clearly improved by adding the coupling element. Figures 3 and 4 show the measured return loss for the proposed antenna with various widths W of the rectangular coupling-patch, and various width W1 of the connecting strip line, respectively. The return loss curve corresponding to the optimized structure is plotted with solid thick-line in Figures 3– 4. As it is shown, the

Figure 6

Measurement and simulation results of the antenna return loss

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Figure 7

Measured x-z plane radiation patterns at resonant frequencies 6.68, 8.88, 9.56, and 10.61 GHz, respectively

impedance matching of the printed dipole antenna dependent critically on W and slightly on W1. Figure 3 shows several typical results for the antennas with various values of W when W1 is fixed to be 6 mm. It is noted that, the performance of the wideband 6.27–12.47 GHz is mainly influenced by the width W. The ⫺10-dB impedance bandwidth can be improved by tuning the value of the parameter W. When the

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parameter W is shifted from the optimum value (W ⫽ 12 mm), the bandwidth is remarkably narrowed, or divided into smaller bands. Figure 4 gives the effect of the width W1 on the impedance bandwidth when W is fixed to be 12 mm. As it can be remarked, the width W1 of the connecting strip can be adjusted to slightly improve the antenna in-band impedance matching. The optimum value was found to be W1 ⫽ 6 mm.

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Figure 8

Measured y-z plane radiation patterns at resonant frequencies 6.68, 8.88, 9.56, and 10.61 GHz, respectively

In fact, the variation of the impedance matching, with W and W1, is due to the modification of the coupling region condition between the capacitive coupling rectangular-patch W ⫻ L and the spiral dipole. 3. EXPERIMENTAL RESULTS

Prototype of the optimized printed dipole antenna (see Fig. 5) was fabricated based on the dimensions specified in Figure 1.

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The Return loss of the structure was measured over the frequency range 5–13 GHz by using a Wiltron 360B vector analyzer. Measurements on the radiation patterns were performed in anechoic room. Figure 5 illustrates both measured and simulated results on the return loss. As it is shown, a relatively good agreement between the simulated and measured return loss was observed. The small discrepancies between simulated and measured can be attributed to

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the effect of the SMA connector and fabrication imperfections. The realized antenna presents a wide resonant band that extends from 6.27 GHz to 12.47 GHz, with an impedance matching of ⬃66% (for S11 ⬍ ⫺10 dB). As a result, it can be seen that the impedance bandwidth of this resonant band is suitable for communication systems that operate in the C and X-bands Otherwise, a relatively good agreement between simulated and measured return loss results can observed from Figure 6. The discrepancy between measured and simulated results might be attributed to the effect of the SMA connector and fabrication imperfections. The measured far-field radiation patterns of both E␪ and E␾ components for the (x-z) plane and (y-z) plane, at the resonance frequencies 6.68, 8.88, 9.56, and 10.61 GHz, are shown in Figures 7 and 8, respectively. As it is shown, the radiation patterns for the E␪ and E␾ components present a certain degree of similarity in both x-z and y-z planes with several dips. These dips might be attributed to the possible destructive interaction in radiation between the modifiedspiral dipole printed on the top of the substrate and the coupling element printed on the bottom side. However, the radiation is directed toward both negative and positive y-axis. This effect is explained by the radiated fields, at the same time, of the spiral dipole and the capacitive coupling element printed on both sides of the substrate. The power intensity of the far-field radiation of E␪ and E␾ components are approximately similar in both x-z and y-z planes, except for the frequency 6.68 GHz in the y-z plane. This effect can be explained by the spiral topology of the propagation path and the feeding technique. The maximum measured gains observed in the range 6.27– 12.47 GHz at frequencies 6.68, 8.88, 9.56, and 10.61 GHz are 3, 4.1, 2.9, and 3.5 dBi, respectively. 4. CONCLUSION

A novel compact and wideband printed dipole antenna for multiple wireless services has been realized. The proposed double-sided printed dipole antenna with a spiral dipole exhibits a wideband impedance behavior. The compact size of the antenna was obtained by optimizing the antenna geometry, whereas the broadband behavior was achieved especially by a capacitive coupling. The resonant band is about 6.2 GHz (6.27–12.47 GHz) with a fractional bandwidth of 66%. The measured peak gains over this band at frequencies 6.68, 8.88, 9.56, and 10.61 GHz are 3, 4.1, 2.9, and 3.5 dBi, respectively. The wide impedance bandwidth, the relatively good gain and radiation characteristics, and the simple feeding of the proposed antenna make it a good choice for communication systems that operate in the C and X-bands applications. REFERENCES 1. A.A. Eldek, A.Z. Elsherbeni, and C.E. Smith, Wide-band modified printed bow-tie antenna with single and dual polarization for C- and X-band applications, IEEE Trans Antennas Propag 53 (2005), 3067– 3072. 2. C.W. Su, Y.T. Liu, W.S. Chen, Y.T. Cheng, and K.L. Wong, Broadband circularly polarized printed-spiral-strip antenna for 5-GHz WLAN operation, Microwave Opt Technol Lett 41 (2004), 163–165. 3. T.G. Ma and S.K. Jeng, A printed dipole antenna tapered slot feed for ultrawide-band applications, IEEE Trans Antennas Propag 53 (2005), 3833–3836. 4. A.K. Shackelfford, K.F. Lee, and K.M. Luk, Design of small-size wide-bandwidth microstrip-patch antennas, IEEE Antennas Propag Mag 45 (2003), 75– 83.

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5. A.K. Skrivervik, J.F. Zurcher, O. Staub, and J.R. Mosig, PCS antennas design: the challenge of miniaturization, IEEE Antennas Propag Mag 43 (2001), 12–27. 6. T. Hori, Broadband/multiband printed antennas, IEICE Trans Commun E88-B (2005), 1809 –1817. 7. K.L. Wong, Compact and broadband microstrip antennas, New York, Wiley-Interscience, 2002. 8. C.J. Wang and D.F. Hsu, Studies of the novel CPW-Fed spiral slot antenna, IEEE Antennas Wireless Propag Lett 3 (2005), 186 –188. 9. H. Nakano, M. Ideka, K. Hitosugi, and J. Yamauchi, A spiral antenna sandwiched by dielectric layers antennas and propagation, IEEE Trans Antennas Propag 52 (2004), 1417–1423. © 2008 Wiley Periodicals, Inc.

TUNABLE RING LASER BASED ON A SEMICONDUCTOR OPTICAL AMPLIFIER AT 1300 NM USING A SIMPLE WAVELENGTH SELECTION FILTER Mansik Jeon,1 Jeehyun Kim,1 Jae-Won Song,1 Ho Lee,2 Sanghoon Choi,2 and J. Stuart Nelson3 1 School of Electronic and Electrical Engineering and Computer Science, Kyungpook National University, 1370 Sankyuk-dong, Bukgu, Daegu 702–701, South Korea; Corresponding author: [email protected] 2 School of Mechanical Engineering, Kyungpook National University, 1370 Sankyuk-dong, Buk-gu, Daegu 702–701, South Korea 3 Beckman Laser Institute and Medical Clinic, University of California at Irvine, 1002 Health Sciences Road East, Irvine, California 92612 Received 8 October 2007 ABSTRACT: A simple, compact, and low cost tunable ring laser with a commercial semiconductor optical amplifier (SOA) was demonstrated. The tunable ring laser is based on an external wavelength filter cavity that is analogous with the Littman configuration with a diffraction grating, a mirror, and a simple slit. The unique structural advantage of this new system is that the slit is displaced to select a desired wavelength instead of tilting the mirror as in the Littman configuration. This allows easy control over the selected wavelength by the translating action of the slit. The full width half maximum (FWHM) wavelength turning range is 45 nm, and the wavelength resolution is about 2 pm. The demonstrated tunable ring laser has 2 mW output power. The side mode suppression ratios is 70 –73 dB. © 2008 Wiley Periodicals, Inc. Microwave Opt Technol Lett 50: 1317–1320, 2008; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.23340 Key words: semiconductor optical amplifiers; tunable lasers; tunable wavelength filters 1. INTRODUCTION

In general, a tunable laser source (TLS) is used to generate various wavelength emitted from a single wavelength light source. Commercial and scientific interest in tunable lasers continues to grow rapidly because of their potential application in optical components testing, fiber optic sensors, and wavelength division multiplexing (WDM) transmission systems [1, 2]. Most TLSs are developed at the 1.5 ␮m wavelength region which the optical communication fields use. Recently the applications of TLS have expanded into biomedical imaging fields including spectroscopy and optical coherence tomography [3–5]. These fields demand different wavelength scanning ranges other than the 1.5 ␮m region. For example, Lasers have been actively studied at 800 nm ranges, which are useful for eye imaging. Optical penetration in biological samples is optimized for wavelengths in the vicinity of 1.3 ␮m [6, 7].

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