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IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 23, NO. 8, AUGUST 2013
A 325 GHz Quadrature Voltage Controlled Oscillator With Superharmonic-Coupling Jae-Young Kim, Member, IEEE, Ho-Jin Song, Member, IEEE, Katsuhiro Ajito, Member, IEEE, Makoto Yaita, and Naoya Kukutsu, Member, IEEE
Abstract—We present a 325 GHz quadrature voltage controlled oscillator (QVCO) using 0.25 InP HBT technology and superharmonic coupling of two differential Colpitts VCOs. The indisingle-ended output power vidual VCOs exhibit about and 10 GHz frequency tuning range. The quadrature oscillation is successfully demonstrated with dc power consumption of 92.4 mW and phase imbalance less than at around 325 GHz. Index Terms—InP HBT MMIC, quadrature voltage controlled oscillator (QVCO), 300 GHz, THz wireless communication.
I. INTRODUCTION
T
HE recent emergence of wireless technologies using millimeter waves reflects the huge need for high-speed wireless systems with higher carrier frequencies [1], [2]. In particular, terahertz (THz) waves above 275 GHz have received much attention for the next generation of wireless communications with over 10-Gbit/s data capacity [2]. Although the RF circuit blocks for the transceiver have been challenging in the 300 GHz band, recent semiconductor processes enable high-speed electronic devices operating at up to the 1-THz region with basic functional blocks [3], [4]. For the signal source, fundamental oscillators operating around the 300 GHz band have been demonstrated with a single transistor configuration using InP HEMTs [5]. Differential voltage controlled oscillators (VCO) with InP HBTs have also been reported [6], [7]. The quadrature VCO (QVCO) is another RF circuit block widely used in many applications, such as wireless transceivers and radar equipment. Especially in the THz-wave band, the huge loss of passive structures for quadrature signal generation reveals the advantage of the QVCO. In general, QVCOs in the few-gigahertz band have been realized with LC VCOs by using fundamental injection via coupling transistors [8]. Recently, the push-push superharmonic coupling technique without active devices or a transformer has enabled the development of a highfrequency QVCO operating at around 100 GHz [9], [10]. In this paper, we present a QVCO operating around 325 GHz using InP heterojunction bipolar transistor (HBT) technology. The QVCO is composed of two differential VCOs which are Manuscript received February 25, 2013; revised May 13, 2013; accepted May 17, 2013. Date of publication July 16, 2013; date of current version August 05, 2013. The authors are with the NTT Microsystem Integration Laboratories, NTT Corporation, Kanagawa 243-0198 Japan (e-mail:
[email protected];
[email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/LMWC.2013.2270430
cross-locked for quadrature oscillation with push-push superharmonic-coupling via the common nodes. The details of the individual differential VCO can be found in our previous report [7]. II. CIRCUIT DESIGN The QVCO was designed with the 0.25double heterojunction bipolar transistor (DHBT) process [4]. The extrapolated maximum current gain cutoff frequency ( ) and max) of the transistor are 370 and imum power gain cutoff ( 650 GHz, respectively, at a bias condition of and . The metal interconnects are composed -thick first metal of four layers (M1-M4), where the 0.8 -thick topmost metal (M4) are separated by a (M1) and 3 6 -thick benzocyclobutene (BCB) interlayer. Fig. 1 shows a schematic diagram of the QVCO which is composed of cross-locked two differential VCOs. Each differential VCO follows common-collector Colpitts oscillator topology [7]. The oscillator core is composed of a pair of HBTs (0.25 4 ) commonly biased with current density of 7 . The varactor is realized with a diode-connected ) whose capacitance ranges double-finger HBT (0.25 3 of about 15 at 300 GHz. Because from 10 to 18 fF with ) of the core HBTs is about 12 fF base-emitter capacitance ( in this bias condition, the capacitive feedback is composed of the base-emitter capacitance and varactor capacitance, where the feedback ratio is around one. This configuration enables a simple layout and small parasitics without an MIM capacitor. Then, by loading inductance (25 pH), the oscillation frequency was set at around 325 GHz. To reduce the loading effect, cascode output buffers were added through T-matching circuits, which adjust the input impedance of the buffer stage. The differential operation of each VCO is enabled with common-mode resistors along the center of the symmetric structure, which makes the odd-mode impedance smaller than the even-mode impedance. The differential operation makes these nodes virtual grounds at the fundamental oscillation frequency, thereby inherently generating the superharmonic signals. For quadrature oscillation, the two differential VCOs are in-phase coupled to each other through the common nodes of the tail resistor and the varactor control nodes. The center of the coupler with a quarter-wavelength stub at the superharmonic frequency exhibits high impedance, which enforces the odd-mode operation. Fig. 2 shows the simulated start-up voltage transient waveforms at the VCO output nodes. The two VCOs, which start the oscillation independently, are
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KIM et al.: A 325 GHZ QUADRATURE VOLTAGE CONTROLLED OSCILLATOR WITH SUPERHARMONIC-COUPLING
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Fig. 1. Schematic diagram of the QVCO with super-harmonic-coupling of differential Colpitts VCOs.
Fig. 3. Experimental setup for relative phase measurement of QVCO output waveforms with chip photograph of the integrated QVCO and switches (Area ). The inset is schematic diagram of core, including switches, is 540 420 of the switch. Fig. 2. Simulated voltage transients at the QVCO output nodes of in Fig. 1. and
,
.
.
cross-locked to each other, resulting in steady-state quadrature output after 170 ps. The interconnections were realized with inverted microstrip lines with the topmost layer (M4) as ground. Although the inverted microstrip lines introduce a power loss due to the small metal thickness and width, the advantages of the short connection length and elimination of the inductive and lossy via structure outweigh the power loss. III. RESULTS Fig. 3 shows a photograph of the fabricated chip and the measurement setup. The chip includes the QVCO and two on-chip switches for relative phase measurement. In the QVCO, each core HBT draws bias current of 7 mA with so that the dc power consumption is 92.4 mW. The on-chip switch can select one of the I and Q outputs from the QVCO as shown in inset of Fig. 3. To reduce the loading effect by switching, 6 dB pi-attenuator was used resulting in over 20 dB input return loss and about 13 dB insertion loss. On-wafer measurement was performed with a Cascade WR-3 waveguide probes and a VNA extenders used as the harmonic mixers. By supplying LO signal of around 13 GHz to the harmonic mixer, the oscillator output around 300 GHz is down-converted to the low-frequency for measurement. In the measurement setup, the waveguide probe including a 20 cm-long s-band waveguide section has insertion loss of around 8–10 dB according to the provider’s data. The VNA extender used as the harmonic mixer includes a directional coupler and an attenuator for the output matching so that the conversion loss of the VNA extender is about 25–27 dB. So, the overall power loss by the off-chip signal distribution
Fig. 4. Measured RF spectrum of harmonically frequency down-converted VCO output signal.
path composed of probe, mixer and waveguides was estimated as at around 300 GHz. Because the integrated on-chip switch is highly lossy, the VCO characteristics of oscillation frequency, output power and phase noise were evaluated using separately fabricated VCO chips which do not contain the switches [7]. The single-ended output power as a function of the oscillation frequency can be found in [7, Fig. 3]. The VCOs with design variation for different center frequencies can operate from 310 to 365 GHz, while each VCO is frequency-tuned with the varactor control voltage ( ). The maximum frequency tuning range is about 10 GHz in the 310 GHz band. By excluding the power loss of the setup from the measurement result, the single-ended output power of the VCO was estimated as at 310 GHz. Fig. 4 shows the measured spectrum of frequency down-converted signal. The phase noise is about at 10 MHz offset frequency including the noise contribution of the measurement setup. The quadrature accuracy of the QVCO was evaluated by measuring the phase difference between and outputs. Although the QVCO is free-running, the phase relationship be-
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IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 23, NO. 8, AUGUST 2013
quadrature oscillation was well demonstrated with resulting I/Q phase imbalance less than at around 325 GHz (see Table I) [11], [12]. IV. CONCLUSION The first QVCO that operates at over 300 GHz was demonstrated using the 0.25 InP HBT process. The QVCO exhibits about single-ended output power and about a 10 GHz frequency tuning range with dc power consumption of 92.4 mW. The simultaneous quadrature output of the QVCO can be widely used for 300 GHz-band applications, including wireless transceivers. REFERENCES Fig. 5. (a) Waveforms of the quadrature output of the QVCO, measured with an oscilloscope after harmonic down-conversion and (b) relative phase difference of I/Q output signals.
TABLE I COMPARISON WITH PRIOR ART
tween the four outputs is maintained inside the chip so that the relative phase of the and signals can be measured using the signal as a common reference. To eliminate the phase error due to the path imbalance, one of the and signal selected by the on-chip switch was distributed through the same RF path including the mixer and LO source. As a result, the measurement is not affected by the RF path except the loading effect of the switch. For stable measurement, a VCO analyzer tracked the oscillation frequency of the QVCO by monitoring the signal and interactively tuned the control voltage ( ) to hold the QVCO in a fixed frequency. Fig. 5(a) shows the frequency down-converted waveforms of and signals measured with oscilloscope. The two output waveforms show a nearly 90 phase difference. Fig. 5(b) shows the phase difference of and signals as a function of oscillation frequency. Although the measured frequency range was limited due to the lossy switch and measurement setup, the
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