A Contactless Electrical Energy Transmission System - IEEE Xplore

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 46, NO. 1, FEBRUARY 1999

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A Contactless Electrical Energy Transmission System Don A. G. Pedder, Andrew D. Brown, Senior Member, IEEE, and J. Andrew Skinner

Abstract— Most mains-operated equipment in use today is connected to the supply via plugs and sockets. These are generally acceptable in benign environments, but can be unsafe or have limited life in the presence of moisture. In explosive atmospheres and in undersea applications, special connectors must be used. This paper describes a technique, the contactless energy transfer system (CETS), by which electrical energy may be transmitted, without electrical connection or physical contact, through nonmagnetic media of low conductivity. The CETS, which has been used to transfer up to 5 kW across a 10-mm gap, employs high-frequency magnetic coupling and enables plug-in power connections to be made in wet or hazardous environmental conditions without the risk of electric shock, short circuiting, or sparking. Energy may be transmitted without the necessity for accurately manufactured “plug-and-socket” mechanisms and may be transmitted from source to load, even when there is relative motion. Load-source voltage matching may be made inherent to the system.

(a)

Index Terms—Contactless power transmission, electrical engineering power.

I. INTRODUCTION (b)

A

Fig. 1. The transformer. (a) Transformer with separated core structure. (b) Simple circuit model of the transformer.

Manuscript received December 23, 1997; revised June 1, 1998. Abstract published on the Internet August 25, 1998. This work was supported by ERA Technology Ltd., U.K. D. A. G. Pedder is with the Software and Systems Division, ERA Technology Ltd., Leatherhead, Surrey KT22 7SA, U.K. A. D. Brown is with the Electronic Systems Design Group, Electronics Department, University of Southampton, Southampton, Hampshire SO17 1BJ, U.K. J. A. Skinner is with Coutant Lambda Ltd., Ilfracombe, EX34 8ES, U.K. Publisher Item Identifier S 0278-0046(99)00916-8.

This notwithstanding, the attraction of transformer-based power coupling in certain environments is unquestionably high [2], [3]. Where there is (or may be) relative motion between transmitter and receiver [4]–[6], contact-based techniques (rolling/sliding carbon/metal contacts, trailing leads) have clear disadvantages. Even when there is no relative motion and the environment is relatively benign (for example, the charging of electric vehicles [7]–[9]), the absence of physical galvanic connections is an obvious advantage, certainly from the perspective of reliability and safety. In more hostile environments [10], there seems little realistic alternative to a transformer-based system. However, all the techniques described in the above, while topologically elegant, are simply loosely coupled transformers, and certainly no attention seems to have been paid to matters of EMC compliance [1], aside from [10], where it is not an issue. If the gapped transformer approach is to be applied to power coupling, the effect of the high magnetizing current must be overcome and the efficiency raised, the voltage transfer ratio must be made reasonably constant, and the external magnetic leakage must be reduced to acceptable levels. In the contactless energy transfer system (CETS), the design of the electronics circuitry is such that the output voltage is sensibly constant and efficiency is good. A high magnetizing current is tolerated,

TRANSFORMER may be used to supply electrical energy to a load and, at the same time, provide galvanic isolation. If the primary and secondary windings of the transformer are wound on separate magnetic structures, as shown in Fig. 1(a), then energy coupling is possible without physical connection between the source and load units. A transformer having the form shown in the figure may be represented by the well-known transformer circuit model shown, with source and load, in Fig. 1(b). The large air gap in the magnetic path gives the transformer a low magnetizing and high leakage inductances and . The inductance magnetizing current will be high, resulting in high primary winding loss, and the output voltage will be load sensitive, so that voltage regulation is poor. The transformer will be inefficient and, although the high reluctance of the air gap will result in low flux density, flux leakage will be high. There is also a high probability of noncompliance with electromagnetic compatibility (EMC) and safety regulations [1].

0278–0046/99$10.00  1999 IEEE

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Fig. 3. Basic power circuit.

Fig. 2. Outline CETS arrangement.

since it allows efficiency improvements in other circuit areas, and external magnetic fields are controlled by careful attention to the physical form of the magnetic components. II. SYSTEM CONCEPTS A CETS system is shown in Fig. 2. It operates from an ac supply and provides an isolated dc output, and consists of an input rectifier, a variable-frequency inverter, a magnetic coupling device with a resonant secondary circuit, and an output rectifier and reservoir capacitor. Other circuitry may be provided, for example, a unity power factor input stage or an output regulator, but these are not parts of the CETS. In some applications, the desired additional functionality may be obtained by modifying the CETS rather than by adding extra circuits. In the CETS shown in Fig. 2, the input from the prime electrical power source is rectified to provide a dc supply for a voltage-source variable-frequency square-wave inverter which feeds the primary winding of the magnetic coupling device. The secondary winding of the coupling device feeds the output bridge rectifier via a series capacitor. The dc side of the output rectifier is connected to a reservoir capacitor which feeds the load. Additional filter or regulator circuits may be included here as necessary. The leakage inductance of the magnetic coupling and the in Fig. 2 form a seriessecondary-side series capacitor

tuned circuit. If the inverter operates at the resonant frequency of this circuit, then, at the fundamental frequency, the leakage inductance of the coupling device is nullified and good output voltage regulation is obtained. Should the relative positions of the primary and secondary sections of the coupling device change, then the leakage inductance will vary and the resonant frequency will change. To maintain good voltage regulation, the inverter frequency must accurately track the resonance of the secondary circuit. The external magnetic leakage field is reduced to acceptable levels by attention to the physical form of the two parts of the coupling device, discussed in the next section. III. POWER CIRCUITRY The power circuit of the CETS technology uses a lowloss zero-voltage-switching (ZVS) half- or full-bridge topology using power MOSFET switching devices. The use of ZVS techniques reduces switching losses and reduces highfrequency radiated noise. Fig. 3 shows the basic half-bridge is considered to be conductpower circuit. If MOSFET at the midpoint of and is ing and the voltage has two half the supply voltage, then the current in components, a linear, rising, magnetizing current component plus a near-sinusoidal component coupled from the secondaryside resonant circuit, which drives into a high-frequency short is gated off while current is flowing into circuit. If , then its output capacitance may transformer winding be used as a turn-off voltage snubber, allowing the current in to fall to near zero, with only a small voltage appearing

PEDDER et al.: A CONTACTLESS ELECTRICAL ENERGY TRANSMISSION SYSTEM

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colinear along the length axis , and the windings would be placed at points “ ” and “ .” The distance “ ” and the width of the winding at “ ” would be arranged to minimize fringing effects in that winding. Calculation of the magnetizing flux density is simple, but determination is more difficult, since the of the inductance factor effects of fringing fields are significant. The inductance factor of the core shown in Fig. 4 is dominated by the air given by gap and has an approximate inductance factor (1)

Fig. 4. Typical core structure.

across . Once is in the off state, an oscillation takes place between the nonlinear output capacitance of the two . If sufficient energy is stored MOSFET’s and transformer , the voltage in the magnetizing and leakage inductances of will commutate in a lossless manner across MOSFET is to 0.7 V and its internal diode will conduct. If then gated into the on state, its turn-on transition would be is conducting, the current in essentially lossless. Once will reverse and the lossless switching can be repeated back to . In order to with current commutating from achieve ZVS, the operating frequency, gate drives, and device selection must be optimized for the transformer characteristics. A. Wound Components The gate drive and current sense transformers should have low leakage inductance and, because of the high operating frequencies, the ferrite core flux densities should be low. For Philips 3C80-grade ferrite, operation at approximately 50 mT and 100 kHz is acceptable; the flux density would then fall to about 20 mT as the frequency was increased to 250 kHz. The design of the coupling device (output transformer) involves compromise. The magnetizing current must be relatively high to ensure ZVS of the MOSFET power switches, but, at the same time, the primary magnetizing flux density must be low enough to prevent excessive heating of the primary core section. In experimental work, Siemens N27grade ferrite has been used for the output transformer. An outline of the design path used to derive details of the coupling device is set out in the following. • Select a core structure to meet the physical requirements of the application, preferably with the secondary enclosing the primary, this minimizing external flux leakage. The core overlap should be as large as possible to give good coupling. Fig. 4 shows a possible structure in which the cross-sectional areas of the cores are constant and with a separation . In operation, the cores would normally be

figure achieved is usually about three In practice, the times this calculated value. With core structures of the types considered, fringing effects are extremely high and, in practice, finite-element analysis may be used to give accurate verification of a proposed core structure if desired. • Calculate the number of primary turns so that the magnetizing current is equal to the full-load current referred to the primary circuit. Typically, the magnetizing inductance of the uncoupled primary is found to be about one-half that of the coupled primary. • Calculate the average primary circuit magnetizing flux density using the expression (2) where half the dc supply voltage; frequency of oscillation; number of turns; cross-sectional area of the primary core structure. The expression is derived from the general transformer equation for square-wave excitation • The losses in the primary should then be checked against the core material data. Experimental couplers of N27 material have been found to operate satisfactorily at 150 kHz and flux densities of 100 mT. The number of secondary turns may be calculated by assuming that the coupling coefficient is about 0.6. The secondary turns are then given by the expression (3) is the number of secondary turns and is the where output voltage. • Determine the inductance of the secondary using the figcalculated previously. The leakage inductance, ure for on the secondary side, is typically 50% of the secondary inductance. • Calculate the peak resonant circuit voltage from the secondary leakage inductance. At the secondary-circuit factors used in practice, the secondary current is sensibly times the output sinusoidal, so that the peak current is current (4)

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so that (5) where leakage inductance; dc output current; characteristic impedance of the resonant circuit. If the peak resonant voltage is less or equal to the output voltage, then the secondary flux density is given by (6) is the cross-sectional area of the secondary. where If the peak resonant voltage exceeds the output voltage as ratio a result of high current operation or because the of the resonant circuit is high, the secondary flux density will increase. The increased flux density is a result of the flux associated with the resonant inductor. Fig. 5 shows a series-resonant output stage with typical waveforms for high current operation at the resonant frequency. Now, so that During a half cycle of steady-state resonant operation, the after the start peak flux density is achieved at a time , of the half cycle. For (7) When peak flux density is achieved,

Fig. 5. Secondary voltage waveforms.

so that

so that (8)

, so that does not At low current levels when the exist. Peak flux density then occurs at time , (8) gives driving voltage changes state. When the point in the waveform at which peak flux density is achieved and, by integrating the voltage over one halfcycle from this point, the flux swing can be determined. A representative example is a 150-W CETS providing a 30-V 5-A dc output. For this unit, 22 ; 754 10 ( kHz; s); 5 A; 30 V; 172 V. s and, to find the peak–peak flux From (8), density swing, we must integrate from 1.85 s to (1.85 4.17) s. This gives

With a secondary core assembly of cross-sectional area 176 mm carrying a 15-turn winding, the peak flux density in the secondary core is approximately 88 mT. The flux produced by the driving voltage alone is 24 mT; with high secondary current, the secondary-side core losses of the coupling device increase as a result of energy stored in the leakage inductance. The calculated flux density will only apply to that part of the core where the windings are well coupled to the core. This result is important because it shows that the secondary flux density can be reduced by removing secondary turns and reducing leakage inductance and peak secondary voltage. Reduction in the number of turns also reduces proximity-effect losses in the windings, which should be placed so that they are away from the end poles of the primary core. Experience has shown that flux densities of up to 100 mT at 120 kHz can be used with convection-cooled N27 ferrite. If reduction in turns is necessary, but results in excessive primary flux density, then the core cross-sectional area must be increased. The calculations detailed above enable first-pass design to be carried out, but modification is frequently necessary to optimize performance. The cores used in the experimental work were constructed from commercially available ferrites by cutting with a diamond saw and reassembly using adhesives.


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Fig. 7. Circuit for extraction of secondary-current phase information.

Fig. 6. Secondary-current phase detection.

For volume production, low-cost cores could be manufactured, since the large air gap removes the need for core grinding. IV. CONTROL The CETS technology uses proven resonant-circuit techniques to overcome the characteristics of a transformer with low coupling coefficient and low magnetizing inductance, but the resultant resonant circuit has a high characteristic impedance. Operation close to the resonant frequency is essential to enable good power transfer. The resonant frequency will vary as the secondary winding assembly moves with respect to the primary because of changes in leakage inductance and, also, with change in loading. The control system developed for the CETS uses phase-locked loop techniques to allow the operating frequency to track the resonant frequency and to allow for tolerance variation in the tuning components. As a result, the system output impedance is sensibly constant. If the phase of the secondary current (as referred to the primary winding) is in phase with the primary driving voltage, the resonant condition is effectively met. The operating frequency of the system must be adjusted to meet this condition. The

design of a system having separable primary and secondary units without contact implies that the secondary current cannot be directly monitored, so a signal giving secondary-current phase information must be derived from the primary circuit. The primary current has two components, the magnetizing current and the referred secondary current. Because the driving voltage on the primary is a square wave, the magnetizing current waveform is triangular with the fundamental component lagging the driving voltage by 90 . The low magnetizing inductance on the primary side results in a high magnetizing current. At resonance, the secondary current will be in phase with the driving voltage and will be sensibly sinusoidal in form. The magnitude of the secondary current is load dependent, so that, on the primary side, the referred secondary current may be smaller in magnitude than the magnetizing current. It may, therefore, be necessary to detect the phase of this small component in the presence of a larger, quadrature, triangular magnetizing current, as shown in Fig. 6. The phase of the secondary current ( in Fig. 6) is determined by monitoring the primary current and removing the magnetizing current component. This is achieved using a double-differentiator and limiting amplifier. Double differentiation of the triangular magnetizing current results in the generation of pulses of alternating polarity and synchronous with the points of inflection of the magnetizing current, as shown in Fig. 6(d). Since the secondary current, referred to the primary, is sensibly sinusoidal in form, double differentiation of this component results in inversion and scaling only. The limiting amplifier effectively removes the magnetizingcomponent pulses and produces a square-wave output giving the secondary-current phase information. This is shown in Fig. 6(i). In practice, this signal processing may be carried out by the circuit shown in Fig. 7. The differentiators do not require good dc performance, and low-cost logic inverters may be used. These devices have been proven adequate at frequencies up to about 200 kHz. The integrating capacitors fitted to the two differentiator circuits are designed to limit the highfrequency gain of the amplifiers. It is important that an overdamped characteristic is achieved and that noise does not cause multiple zero crossings of the second differential. The

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(a)

(b)

(c) Fig. 8. CETS output voltage variation.


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first zero crossing of the second differential is used in the phase comparator and false zero-crossing detection will result in incorrect circuit operation. V. PRACTICAL RESULTS A. Low-Power CET System A low-power CETS assembly has been constructed to operate from a 48-V dc source and to supply power to an LED lighting system across a variable air gap. The secondary (load) unit contained constant-current drivers for four strings of eight LED’s per string. The nominal power requirement was 15 mA at 15 V per string. The primary and secondary magnetic structures consisted of identical U cores, minimum separation 6 mm. The variation in dc voltage fed to the constant-current drivers and in string current are shown as a function of core separation and misalignment in Fig. 8. The insert in Fig. 8(a) shows the core structure and defines the misalignment axes. B. High-Power CET System A CETS unit has been constructed to provide an output of 5 kW at 230 V to a heating load when supplied from a 400-V dc source, and the overall efficiency was around 92%. When operating at 2.5 kW and supplied from a 240-V 50-Hz ac source via a conventional unity power-factor input stage, the overall efficiency (including the input stage) was reduced to approximately 83%. Fig. 9 depicts the transfer efficiency and overall power loss as functions of power transfer level for this high-power unit (the final entry in Table I) when fed from the ac supply. Note that the efficiency curve is for the electromagnetic component of the power transfer chain only; losses in the drive and control electronics will take around another 20 W. The air gap between the primary and secondary magnetic structures was 8 mm and the areas of the opposing coupling faces were about 900 mm on the primary and 1600 mm on the secondary. The efficiency and loss characteristics of the other kilowatt application in Table I are similar. A more germane figure of merit is the overall power loss, as this ultimately represents the amount of heat that has to be removed from the system. (For this reason, the power loss is also shown in Fig. 9.)

Fig. 9. Efficiency and power loss for the 5-kW application. TABLE I CETS UNITS BUILT AND TESTED

C. Other CET Systems Other CETS units have been constructed to operate from a wide range of ac and dc supplies and operating at power levels from a few watts upwards; a list of units is given in Table I. VI. FINAL COMMENTS The CETS technology has been tested in a variety of systems operating from various input sources and producing a variety of outputs, both ac and dc. The technology can be used to enable the production of a contactless “plug and socket” which may be employed, for example, in wet conditions at sea or in the open, to enable quick and simple power coupling without the danger of electric shock and without loss of

reliability as a result of contact corrosion. A CETS connector may be coupled or uncoupled, with power applied, underwater. The absence of mating contacts in CETS removes the possibility of sparking, so that, with sealed primary and secondary units, a plug-and-socket assembly suitable for use in hazardous areas may be produced. This offers opportunities in lighting and enables live circuit relamping in dangerous environments. CETS may be used to couple energy into moving loads, such as turbine rotors, brushless exciters in synchronous machines, aircraft propellers and helicopter rotors, and sliding doors. Electric vehicle batteries may be charged using the CETS technique. In a domestic environment, kitchen equipment may

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be powered without connection if standing over a source unit below the worksurface which may remain unbroken. This has advantages from both electrical safety and hygiene points of view. REFERENCES [1] European EMC Directive (89/336/EEC, amended 92/31/EEC), Regulations EN60555, EN60335. [2] K. W. Klontz, D. M. Divan, D. W. Novotny, and R. D. Lorenz, “Contactless power delivery system for mining applications,” IEEE Trans. Ind. Applicat., vol. 31, pp. 27–35, Jan./Feb. 1995. [3] E. Abel and S. M. Third, “Contactless power transfer—An exercise in topology,” IEEE Trans. Magn., vol. MAG-20, pp. 1813–1815, Sept. 1984. [4] A. Esser and H. Skudelny, “A new approach to power supplies for robots,” IEEE Trans. Ind. Applicat., vol. 27, pp. 872–875, Sept./Oct. 1991. [5] K. Lashkari, S. E. Shladover, and E. H. Lechner, “Inductive power transfer to an electric vehicle,” in Proc. 8th Int. Electric Vehicle Symp., 1986, pp. 258–267. [6] A. W. Kelley and W. R. Owens, “Connectorless power supply for an aircraft-passenger entertainment system,” IEEE Trans. Power Electron., vol. 4, pp. 348–354, July 1989. [7] F. Sato, J. Murakami, T. Suzuki, H. Matsuki, S. Kikuchi, K. Harakawa, H. Osada, and K. Seki, “Contactless energy transmission to mobile loads by CLPS—Test driving of an electric vehicle with starter batteries,” IEEE Trans. Magn., vol. 33, pt. 2, pp. 4203–4205, 1997. [8] J. Murakami, F. Sato, T. Watanabe, H. Matsuki, S. Kikuchi, K. Harakawa and T. Satoh, “Considerations of cordless powerstation—Contactless power transmission system,” IEEE Trans. Magn., vol. 32, pt. 2, pp. 5037–5039, 1996. [9] J. W. Piper and J. T. Boys, “Inductive power distribution system,” International Patent WO 92/17929, Oct. 1992. [10] B. J. Heeres, D. W. Novotny, D. M. Divan, and R. D. Lorenz, “Contactless underwater power delivery,” in Proc. IEEE PESC’94, 1994, vol. 1, pp. 418–423.

Don A. G. Pedder received the B.Sc. (Eng.) degree in 1960 from the Royal Military College of Science, Shrivenham, U.K., and the M.Sc. (Eng.), the D.I.C. in 1965, and the Ph.D. degrees from Imperial College, University of London, London, U.K. He was with the Royal Naval Scientific Service from 1953 to 1960. He then assumed a research post at Imperial College until 1964. He became a Lecturer at Kingston Polytechnic, U.K., where he remained until 1986, becoming Head of the Department of Electronic Systems Engineering and building a substantial consultancy practice. In 1986, he left Kingston Polytechnic to accept an appointment as Manager of the Power Electronics Department at ERA Technology Ltd., Leatherhead, U.K., where he is currently a Senior Consultant in the Software and Systems Division. He has worked on high-power electronics since his military service in 1954, long before the term power electronics was in use. His range of experience includes high-power radio transmitters, radar modulators, power supplies, inverters, and high-power active filter systems. He has held Visiting Lectureships in power electronics at Imperial College and Queen Mary College, London, U.K., and was appointed Visiting Professor in Design Automation in the Department of Electronics, University of Southampton, Southampton, U.K., in 1996. Dr. Pedder is a Fellow of the Institution of Electrical Engineers (U.K.).

Andrew D. Brown (M’90–SM’96) received the B.Sc. degree in physical electronics and the Ph.D. degree in microelectronics from the University of Southampton, Southampton, Hampshire, U.K., in 1976 and 1981, respectively. He was appointed a Lecturer at the University of Southampton in 1981, a Senior Lecturer in 1989, and a Reader in 1992. He was a Visiting IBM Scientist in 1983 at Hursley Park, U.K., and a Visiting Professor at Siemens, NeuPerlach Munich, Germany, in 1989. He is currently Head of the Electronic Systems Design Group, Electronics Department, University of Southampton. This group has interests in all aspects of simulation, modeling, synthesis, testing, and design of systems. Dr. Brown is a Fellow of the Institution of Electrical Engineers (U.K.), a Chartered Engineer in the U.K., and a European Engineer.

J. Andrew Skinner received the B.Sc. degree in physics with physical electronics from the University of Bath, Bath, U.K., in 1987. Between 1990–1994, he was a Design Engineer in the Power Electronics Department, ERA Technology Ltd., Leatherhead, U.K., where his main interest was high-frequency power conversion for power supply and lighting applications using resonant and quasi-resonant techniques at frequencies to over 1 MHz. He is currently a Senior Design Engineer with Coutant Lambda Ltd., Ilfracombe, U.K. Mr. Skinner is a Chartered Electrical Engineer in the U.K.