A Low-Profile Wideband Planar Antenna - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 9, SEPTEMBER 2013

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A Low-Profile Wideband Planar Antenna Mingjian Li, Student Member, IEEE, and Kwai-Man Luk, Fellow, IEEE

Abstract—A wideband planar antenna with a low profile of , where is the wavelength referring to the center frequency of the operating band, is proposed. Using the concept of complementary antenna, two shorted quarter-wave microstrip patch antennas, operating as magnetic dipoles, and two planar electric dipoles are connected together and excited simultaneously by a coaxial-to-microstrip impedance transformer. For further size reduction and back radiation suppression, a U-shaped reflector is employed. Experimentally, the proposed antenna exhibited an impedance bandwidth of 51.5% with from 1.96 to 3.32 GHz, a boresight gain of 6.9 1.1 dBi and a unidirectional radiation pattern with low cross-polarization and back radiation levels. Moreover, the dc grounded structure fulfills the requirement for lightning protection in outdoor applications. Index Terms—Low-profile antenna, unidirectional patterns, wideband antenna.

I. INTRODUCTION

W

ITH the advancement of radio frequency technologies, various modern wireless systems have been proposed and developed such as WiFi, WiMax, Zigbee, 3 G and LTE. In these systems, it is desirable to have antennas satisfying stringent requirements including wide impedance bandwidth, unidirectional radiation, stable gain, stable radiation pattern, low profile and so on. Conventional antennas like horn and reflector antennas [1] are incompatible with most of the modern wireless systems due to their bulky structures. In the past decades, tremendous research has been carried out to achieve various design objectives. Several wideband lowprofile antennas were designed by employing slot antenna [2], loop antenna [3] and spiral antenna [4]. By putting a dipole in front of a reflector with a distance of about , the antenna can achieve a unidirectional pattern. For reducing the antenna height, the electromagnetic band-gap (EBG) structure as a kind of artificial electromagnetic materials can be used. By simply placing a dipole antenna [5], a folded dipole antenna [6] or even a spiral antenna [7] on a carefully designed EBG structure, the antenna height could be reduced to less than . Basically, the operating bandwidth of the antenna is limited by the bandwidth of the reflection phase of the EBG surface, thus wideband antennas generally cannot be excited on the EBG surface.

Manuscript received November 15, 2012; revised May 02, 2013; accepted June 04, 2013. Date of publication June 11, 2013; date of current version August 30, 2013. This work was supported by a grant from the Research Grants Council of the Hong Kong SAR, Hong Kong [Project No. CityU 119511]. The authors are with the State Key Laboratory of Millimeter Waves, City University of Hong Kong, Hong Kong (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2013.2267715

Another competitive solution to achieve low profile is microstrip patch antenna which has many advantages, such as planar structure, low cost, and ease of manufacture. However, the conventional patch antenna [8] suffers from a very narrow impedance bandwidth (less than 5%), which cannot fulfill the bandwidth requirement of the modern wireless system. Many bandwidth enhancement techniques were introduced, such as the L-probe feed [9], [10], coplanar coupled feed [11], aperture coupled feed [12]–[14], stacked patches [15]–[18], U-slot patch [19], [20] and E-shaped patch [20]–[23]. By employing these techniques, great enhancement on impedance bandwidth up to 50% has been achieved for . However, the radiation patterns of these patch antennas vary with frequency, and high cross-polarization and strong back radiation usually occur across the operating frequency range. Recently, employing the concept of complementary antenna or quasi-Huygens’ source presented in [24], [25], several wideband antennas designated as the magneto-electric (ME) dipole were proposed by Luk et al. [26]. By combining and exciting a magnetic dipole and an electric dipole together, the resulted ME dipole antenna shows good electrical characteristics, including wide impedance bandwidth, stable gain, low back radiation, low cross-polarization level, and symmetrical E- and H-plane radiation patterns. However, in comparison with microstrip patch antenna, the ME dipoles have an obvious drawback that their heights are generally one quarter wavelength , which may be too large in some practical applications. Although some attempts for reducing the antenna height was carried out [26], [27], the structures of the antennas are complex in construction. In this paper, a low-profile wideband planar antenna is presented. Compared to the original complementary antenna with a height of , the proposed antenna has a low profile of (62% reduction) which is close to the height level of a wideband microstrip patch antenna (normally less than ). Meanwhile, it maintains all good electrical characteristics of complementary antenna (or quasi-Huygens’ source) such as wide band and low back radiation. The antenna design process and the geometry description are elaborated in Section II. The antenna operation principle is analyzed in Section III. Several important parameters are studied in Section IV. The measurements on an antenna prototype are compared to the simulations in Section V. II. ANTENNA DESIGN AND GEOMETRY DESCRIPTION A. The Concept of Complementary Antenna As we all know, electric dipole has a radiation pattern like a figure “8” shape in the E-plane and “O” shape in the H-plane. For an electric dipole placed along y-axis, an equivalent electric

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current is towards -y-axis. The electric-field in the far-field zone can be expressed as

(1) A magnetic dipole has a radiation pattern like a figure “8” shape in the H-plane and “O” shape in the E-plane. For a magnetic dipole placed along x-axis, an equivalent magnetic current is towards x-axis. The electric-field in the far-field zone can be expressed as

(2) By simultaneously exciting the electric dipole and the magnetic dipole with the same magnitude in strength, a Huygens’ Source can be realized. Its far-field electric fields in the E- and H- planes are identical and expressed as

(3) And the normalized radiation pattern can be expressed as

(4) As radiation.

,

, which means there is no back

B. Low-Profile Complementary Antenna A magnetic dipole can be realized by many ways, such as a slot [25], vertically oriented shorted patch antenna [26], triangular loop antenna [26] and rectangular loop antenna [27]. After combining a simple electric dipole, they achieve very good electrical characteristics. However, they have a complex structure or a relatively large antenna height of over . A microstrip patch antenna can be used to realize a magnetic dipole. A quarter-wave shorted patch antenna could produce a generally parallel electric field at the radiating slot which is equivalent to a magnetic current. For ensuring a symmetrical radiation pattern, two mirrored triangular quarter-wave shorted patches are arranged like a bowtie. After they are attached by dipoles and excited by a differential feed by means of a balun, the combination performs as a quasi-Huygens’ source. Although this antenna structure is not a rigorous Huygens’ source, it can achieve a unidirectional radiation pattern over a wide operating band. For achieving a better back radiation cancellation, the phase difference between the electric dipole and magnetic dipole should be carefully considered [28]. The length of the electric dipoles and the length of the ground plane affect the phases of the electric dipole and magnetic dipole respectively. Hence, a rectangular ground plane instead of a square one was used for controlling the phase of the magnetic dipole. For further

Fig. 1. Photos of the antenna prototype. (a) Perspective view of the prototype, (b) side view of the prototype, (c) bottom side of the PCB.

size reduction and back radiation suppression, two sides of the rectangular planar reflector are folded vertically. C. Antenna Geometry Fig. 1 shows perspective and side views of an antenna prototype and the bottom side of the printed circuit board. The proposed antenna is composed of a shorted bowtie, two planar dipoles, a U-shaped reflector and a coaxial-to-microstrip feed. The shorted bowtie comprises two triangular quarter-wave shorted patch antennas. The horizontal portion of shorted bowtie and the planar dipoles are built on the Duroid 5870 substrate with a and a . The vertical shorting wall of the shorted bowtie is realized by using metal columns ( , on one side of the bowtie) which are low-cost nickel spacers. The U-shaped reflector has a dimension of . Thus, the total volume of the antenna is . The top view of the antenna is depicted in Fig. 2(a). The Metal Patch realizes the two electric dipoles and the horizontal portion of the shorted bowtie antenna printed on the bottom side of a dielectric substrate. The Feedline denotes the microstrip section of the feed on the top side of the dielectric substrate. This line includes two portions, i.e. a 50-ohm line of and a quarter-wavelength impedance transformer of . And then a via hole is used for connecting the Feedline and Metal Patch. It is noted that the portion of the microstrip impedance transformer above the 3.2 mm separation gap between the two parts of the Metal Patch leads to a back leakage which affects the radiation pattern slightly. This antenna feeding technique is very similar to the method of a microstrip-line fed slot antenna. Fig. 2(b) shows the side view of the antenna. The metal columns, placed uniformly on both sides of the substrate, connect the ground plane and the bowtie. The coaxial cable

LI AND LUK: A LOW-PROFILE WIDEBAND PLANAR ANTENNA

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Fig. 3. Current distributions. (a) .

, (b)

, (c)

, (d)

Fig. 2. Antenna Geometry. (a) Top view (x-y plane), (b) side view (y-z plane).

TABLE I DIMENSIONS FOR THE PROPOSED ANTENNA

is the wavelength referring to the center frequency of the operating band.

penetrates the ground plane to the printed circuit board. One end of the inner conductor of the cable is connected to the SMA connector which is placed under the ground plane. The other end of the inner conductor is connected to the 50-ohm line portion of the Feedline on the upper side of the printed circuit board. The outer conductor of the cable is connected to the ground plane as well as the horizontal portion of the shorted bowtie on the bottom side of the PCB, which works as a metal column for shorting the bowtie. This feeding structure makes the antenna dc grounded. The total height of the antenna H, including the height of metal columns and the thickness of the substrate, is 10.79 mm. The dimensions for the proposed antenna are summarized in Table I. III. ANTENNA OPERATION For investigating the operation of the proposed antenna, the current distribution on antenna surfaces at different time within a period of the center frequency 2.6 GHz is investigated, as shown in Fig. 3. The PCB substrate is set to be invisible in HFSS Solver for clear illustration. At time , the currents on the planar electric dipole attain maximum whereas the currents on the edges along the middle gap between two shorted patches attain minimum. At

Fig. 4. Effect of the length of the electric dipole ).

(keeping

time , the currents on the edges along the middle gap attain maximum because the radiating slots of the shorted patches are excited, whereas the currents on the planar electric dipoles attain minimum. At time , the currents on the electric dipole are dominated again with opposite direction to the currents at time . At time , the currents on the magnetic dipole are dominated again with opposite direction to the currents at time . The result reveals that two degenerate modes of similar magnitude in strength are excited on the electric planar dipole and the quarter-wave shorted patches. The electric currents of both modes have nearly 90 phase difference in the same direction, which can be explained that equivalent electric and magnetic currents are generally in phase and orthogonal to each other. The crossed electric and magnetic currents are not perfectly in phase and not identical in strength. Hence, this complementary structure can be called as a quasi-Huygens’ source. IV. PARAMETRIC STUDY AND ANTENNA ANALYSIS In order to investigate the effect of varying parameters on the performance of the antenna, a parametric study was performed by HFSS. In this study, the metallic layers were assumed to have

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Fig. 5. Simulated radiation patterns at 2.3 GHz and 2.6 GHz. (a) Proposed antenna, (b) antenna without an electric dipole, (c) antenna without a reflector, (d) antenna with a flat reflector.

zero-thickness and the metal columns were set to be metal cylinders for reducing computation time.

to the shorted bowtie, FBR is enhanced significantly and the cross-polarization radiation level is suppressed, which is due to the nature of the complementary antenna.

A. Electric Dipole For investigating the effect of the electric dipole on the antenna performance, the length of the planar electric dipoles was studied. Fig. 4 shows the front-to-back ratios and the SWRs for different electric dipole length with a constant electric dipole width . As the length of the electric dipoles is reduced, the impedance matching becomes worse and the first resonance is shifted to higher frequencies while a fixed second resonance occurs at about 2.95 GHz. This is probably due to the first resonance mainly influenced by the electric dipole. It is also observed that the front-to-back ratio (FBR) is sensitive to this parameter. By increasing the FBR is decreased significantly. As a result, for a good impedance matching with and a relatively large FBR, was selected. As shown in Fig. 4, the antenna without planar electric dipoles has a large SWR and a very low FBR of less than 15 dB, which is because a complementary antenna cannot be constructed only with the shorted patch antennas. To further demonstrate the effect of the electric dipoles, radiation patterns for the antenna with and without the electric dipoles are depicted in Fig. 5(a) and (b). By attaching the electric dipole

B. Magnetic Dipole The length of the horizontal portion of the shorted bowtie was studied firstly. It can be seen from Fig. 6 that by increasing the SWR is increased over the whole operating band and the second resonance is shifted downwards while maintaining a fixed first resonance occurring at 2.1 GHz. This is because the second resonance is mainly caused by the shorted bowtie. In addition, as decreases, the antenna gain is reduced at high frequencies. Hence, the is chosen to be 87.2 mm. The height of the shorted bowtie H was investigated. As shown in Fig. 7, a larger H degrades the antenna gain at lower frequencies and hence causes a larger gain fluctuation. For achieving , H cannot be too small because smaller H deteriorates the in-band impedance matching. By decreasing H, the impedance bandwidth is reduced from 53.8% to 44.4%, in spite of the electric length of H reduced from to ( denotes the center frequency wavelength for the impedance bandwidth of the antenna with varied H). Thus, for maintaining a good impedance matching, a stable gain and a low profile, was selected.


Fig. 6. Effect of the length of the shorted bowtie

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.

Fig. 8. Effect of the number of metal columns N (on one side of the bowtie).

Fig. 7. Effect of the height of the antenna H.

Fig. 9. Effect of the width of the ground plane

C. Metal Columns

of the planar electric dipoles but also the ground plane of the quarter-wave shorted patches. Basically, the U-shaped reflector affects the phase of the shorted patches. Thus, it has a significant effect on the antenna gain and front-to-back ratio. It can be proved from Fig. 3(b) and (d) that as the currents on the radiating slots of the shorted patch antennas are dominant, strong currents flow on the edge of ground plane. As shown in Fig. 5(c), the antenna has only a planar ground plane with the same size as the dielectric substrate (87.2 mm 88 mm), instead of a U-shaped reflector. It achieves a unidirectional radiation pattern, which is because the combination of microstrip patch and dipole antennas is a quasi-Huygens’ source. As shown in Fig. 9, this antenna (No Reflector line) exhibits a stable gain of 7 dBi. And the front-to-back ratio is generally over 10 dB which is smaller than that from a rigorous Huygen’s Source but much larger than that from a patch antenna with a ground plane of the same size as the upper radiating patch. By adopting the U-shaped reflector, the back radiation can be suppressed to lower than 20 dB. Hence, this quasi-Huygens’ source requires the U-shaped reflector to further reduce back radiation. Firstly, the width of the reflector was studied. It can be seen from Fig. 9 that by reducing , the FBR is increased at high frequencies and decreased at low frequencies, and a larger

The shorting walls of quarter-wave shorted patch antennas are realized by metal columns which are equivalent to inductors in parallel. As shown in Fig. 8, as less metal columns are used, the impedance bandwidth is shifted to the lower frequencies but FBR is degraded significantly. This can be simply explained by the formula of resonant frequency which is given by, (5) Less metal columns used means less inductors in parallel which produces larger inductance, thus the resonance is shifted downwards and the electric length of the antenna height is reduced. However, the number of the metal columns shouldn’t be reduced continuously. As only three metal columns are used on one side of the bowtie, a low FBR is received because the function of shorting wall cannot be accomplished. Hence, the number of the metal columns N was chosen to be 5. D. U-Shaped Reflector Unlike the traditional reflector for electric dipole antenna or slot antenna, the U-shaped reflector is not only the reflector

.

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Fig. 10. Effect of the length of the ground plane


Fig. 12. Simulated and measured SWRs and gains.

.

V. ANTENNA PERFORMANCE An antenna prototype was constructed to verify the proposed design, as shown in Fig. 1. The simulation was implemented by EM simulation software Ansoft HFSS, and the measurements on SWR, gain and radiation patterns were accomplished by Agilent N5230A network analyzer and SATIMO complex antenna measurement system. A. SWR and Gain

Fig. 11. Effect of the height of the reflector vertical walls

.

gain fluctuation is received. For a stable gain and high FBR, was selected. Secondly, the length of the reflector was studied. As depicted in Fig. 10, a larger enhances the antenna gain but decreases the FBR, especially at low frequencies. Thus, was chosen to be 120 mm. Finally, the height of the vertical walls was studied. Fig. 11 shows that the antenna gain is not sensitive to this parameter, whereas the FBR is increased with increasing. For maintaining a low profile, the height of the wall cannot be larger than the antenna height H. So, was selected. To further investigate the function of vertical wall, the antenna with a planar reflector was studied. The reflector with a length of 141.6 mm keeps the phase difference between the electric dipole and magnetic dipole but enlarges the total volume of the antenna. As shown in Fig. 11, although the antenna gain is slightly higher, the FBR is less than 20 dB over the operating frequency range which is not preferable. The radiation patterns of the antenna with a U-shaped reflector and this planar reflector are shown in Fig. 5(a) and (d). The back radiation level is reduced from 19.2 dB to 28 dB and 18.6 dB to 20.2 dB at 2.3 GHz and 2.6 GHz, respectively.

Fig. 12 shows the simulated and measured SWRs and gains. The measured results show that the proposed antenna achieves a wide impedance bandwidth of 51.5% from 1.96 to 3.32 GHz. The simulated impedance bandwidth is 52.9% from 1.96 to 3.37 GHz. Within the operating frequency range, the measured antenna boresight gain increases with frequency and varies in the range of 5.8 to 8 dBi. The simulated boresight gain varies between 5.6 dBi and 8.3 dBi. The agreement between measured and simulated results is good. B. Radiation Patterns The measured and simulated radiation patterns at 2, 2.4, 2.9 and 3.3 GHz are depicted in Fig. 13. Good agreement between measurements and simulations is achieved. It can be seen that the E- and H-plane patterns are generally symmetrical and exhibit good unidirectional radiation characteristic. The measured cross-polarization radiation level is below 24 dB over operating frequency range. And as some of the simulated cross-polarized radiation patterns are lower than 40 dB, they cannot be illustrated on the graphs. The measured front-to-back ratio is larger than 20 dB at lower frequencies and gradually drops down to 18.3 dB at 3.3 GHz. The large FBR is because that the proposed design is a complementary antenna or quasi-Huygens’ source and adopts a U-shaped reflector. VI. CONCLUSION A low-profile wideband planar antenna consisting of two triangular shorted patches, two planar electric dipoles and a U-shaped reflector has been presented in this paper. The antenna operation principle has disclosed its nature of the


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REFERENCES

Fig. 13. Simulated and measured radiation patterns of the proposed antenna. (a) 2 GHz, (b) 2.4 GHz, (c) 2.9 GHz, (d) 3.3 GHz.

complementary antenna. A parametric study has been performed for providing a useful guideline of practical design. An antenna prototype with a low profile of has been fabricated and measured. A wide impedance bandwidth of 51.5% and a boresight gain of 6.9 1.1 dBi have been achieved. A unidirectional radiation pattern with low cross-polarization and low back radiation has been found across the entire operating frequency range. Hence, with a low profile and good electrical characteristics, the antenna will find applications in modern wireless communications.

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Mingjian Li (S’10) was born in Beijing, China, in 1987. He received the B.Sc. (Eng.) degree in electronic and communication engineering from City University of Hong Kong in 2010, where he is currently pursuing Ph.D. degree. His recent research interests include wideband antennas, millimeter-wave antennas and arrays, base station antennas, circularly-polarized antennas and small antennas. Mr. Li received the Honorable Mention at the student contest of 2011 IEEE APS-URSI Conference and Exhibition held in Spokane, US. He was awarded the Best Student Paper Award (Second Prize) in the 2012 IEEE International Workshop on Electromagnetics (IEEE iWEM2012) held in Chengdu, China.

Kwai-Man Luk (M’79–SM’94–F’03) was born and educated in Hong Kong. He received the B.Sc. (Eng.) and Ph.D. degrees in electrical engineering from The University of Hong Kong in 1981 and 1985, respectively. He joined the Department of Electronic Engineering, City University of Hong Kong, in 1985 as a Lecturer. Two years later, he moved to the Department of Electronic Engineering, Chinese University of Hong Kong, where he spent four years. He returned to the City University of Hong Kong in 1992, and is currently Chair Professor of Electronic Engineering and Director of State Key Laboratory in Millimeter waves (Hong Kong). He is the author of three books, nine research book chapters, over 290 journal papers and 220 conference papers. He has received five US and more than 10 PRC patents. His recent research interests include design of patch, planar and dielectric resonator antennas, and microwave measurements. Prof. Luk is a Fellow of the Chinese Institute of Electronics, PRC, a Fellow of the Institution of Engineering and Technology, UK, a Fellow of the Institute of Electrical and Electronics Engineers, USA and a Fellow of the Electromagnetics Academy, USA. He is Deputy Editor-in-Chief of PIERS journals. He was a Chief Guest Editor for a special issue on “Antennas in Wireless Communications” published in the PROCEEDINGS OF THE IEEE in July 2012. He was Technical Program Chairperson of the 1997 Progress in Electromagnetics Research Symposium (PIERS), General Vice-Chairperson of the 1997 and 2008 Asia-Pacific Microwave Conference (APMC), General Chairman of the 2006 IEEE Region Ten Conference (TENCON), Technical Program Co-chairperson of 2008 International Symposium on Antennas and Propagation (ISAP), and General Co-chairperson of 2011 IEEE International Workshop on Antenna Technology (IWAT). He received the Japan Microwave Prize at the 1994 Asia Pacific Microwave Conference held in Chiba in December 1994, and the Best Paper Award at the 2008 International Symposium on Antennas and Propagation held in Taipei in October 2008. He was awarded the very competitive 2000 Croucher Foundation Senior Research Fellow in Hong Kong and the 2011 State Technological Invention Award (2nd Honor) of China.