A Planar Reconfigurable Aperture With Lens and ... - IEEE Xplore

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 58, NO. 12, DECEMBER 2010

3547

A Planar Reconfigurable Aperture With Lens and Reflectarray Modes of Operation Jonathan Yun Lau, Student Member, IEEE, and Sean Victor Hum, Member, IEEE

Abstract—This paper presents the design and experimental characterization of a planar 6 2 6 fully reconfigurable array operating at 5.7 GHz, capable of functioning both as a reconfigurable array lens and a reconfigurable reflectarray. First, the design of the array element, which consists of two varactor diode-loaded patches coupled by a varactor diode-loaded slot, is reviewed. Next, the design and fabrication of a planar array is described, and the varactor biasing scheme is discussed in detail. Experimental results demonstrating 2-D beamforming are presented, where a broadside directivity of 20.8 dBi and a beam-scanning range of 50 by 50 is achieved for the lens mode. The ability of the array to also function as a reflectarray is also demonstrated, achieving a directivity of 19.4 dBi at broadside and a beam-scanning range of 60 by 30 . Not only is this array able to demonstrate full 2-D beamforming, it is also low-cost and easy to fabricate, making it very attractive for applications where high-gain beam-scanning is needed. Index Terms—Antenna arrays, lens antennas, microstrip arrays, reconfigurable antennas, reflectarrays, reflector antennas, varactors.

I. INTRODUCTION

T

HE need for high-gain reconfigurable microwave antennas has recently emerged in many applications such as radar and satellite communications requiring real-time electronic beamforming. Reconfigurable antennas also have significant promise for shaped-beam synthesis, particularly in satellite and other multipoint communication systems. While phased arrays have been presented as elegant solutions for these needs, they have drawbacks that include costly, bulky, and lossy feed networks and transceiver implementations. Optically-inspired antenna designs such as reflectarrays and array lenses have significant potential for addressing these challenges, and have been demonstrated as viable platforms for spatial power combining. Reflectarrays are a prominent example of microwave antennas that inspire this work on array lenses. Techniques for fixed phase control in reflectarrays such as using open-ended waveguides, microstrip stub loading, variable patch size, and a wealth of other techniques are well summarized in [1] and [2]. Manuscript received April 15, 2010; revised July 30, 2010; accepted September 10, 2010. Date of publication November 09, 2010; date of current version December 10, 2010. This work was supported by the Natural Sciences and Engineering Research Council of Canada (NSERC). The authors are with the Edward S. Rogers Sr. Department of Electrical and Computer Engineering, University of Toronto, Toronto, ON M5S 1A1, Canada (e-mail: [email protected]; [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2010.2086373

With advances in micro-electromechanical systems (MEMS) and material technologies, researchers have studied electronic tuning techniques for beamforming in phased arrays [3], [4], and reflectarrays, many of which are summarized in [5], [6]. Common techniques include the use of varactor diodes [7], [8] or MEMS capacitors [9], [10] to reactively load structures in a continuous fashion, or the use of PIN diodes [11] or MEMS switches [12] to switch current paths on the reflectarray element. The concept of spatial power combining to achieve power amplification has also been explored in fixed [13] and reconfigurable reflectarrays [14]. The transmissive counterpart to the reflectarray, the array lens (also known as a transmitarray) has been studied for many years, and typically consist of two coupled antennas, where the coupler or antenna in each element is physically tuned to yield a particular phase shift. Common coupling mechanisms include slots [15] and transmission lines. Array lenses have also been implemented using cascaded layers of resonant antennas such as rings [16]. Fixed array lenses with amplifiers have been demonstrated for use as spatial power combiners [17]–[19] and quasi-optical systems [20]. Active, fixed-pattern array lenses where the coupling structure between two patch antennas includes an amplifier have also been well-studied [21], [22]. Fixed lens arrays have also been proposed for creating spatial band-pass filters [23]. It is only more recently that tunable array lenses have been proposed and demonstrated. A tunable liquid crystal frequency selective surface (FSS) was presented in [24], and a varactor diode-based tunable FSS was presented in [25]. While these surfaces are reconfigurable, it is the operating frequency of the entire lens that was tuned in these designs. That is, phase tunability was not a design objective, and the elements of the arrays could not be individually tuned. Thus, the surfaces are not suitable for beamforming applications where phase control is critical. Moreover, the structures do not have sufficient order to achieve 360 of phase tunability with usable insertion loss. In [26], a reconfigurable array lens was proposed for the purpose of antenna beamforming using MEMS switches to achieve 2-bit phase control. 1-D beamforming was demonstrated experimentally at millimeter wavelengths. In [27], a fully reconfigurable array lens was presented for beamforming. The drawbacks of the design, however, include its physical thickness and fabrication complexity, as microstrip layers need to be soldered together at perpendicular angles. In contrast to the wealth of work that has been presented for reconfigurable reflectarrays, there have not been many investigations into reconfigurable array lenses, especially for beamforming applications.

0018-9480/$26.00 © 2010 IEEE

3548


Fig. 1. Array lens beamforming illustrations. (a) Array lens beamforming phases. (b) Beamforming coordinates.

This underscores the significance of the work that this paper presents, which is the design and experimental demonstration of a reconfigurable array lens at 5.7 GHz that is also capable of functioning as a reflectarray. The design, based upon the array element proposed in [28] is low-cost, easy to fabricate, and can perform full 2-D beamforming. Furthermore, by using varactor diodes that provide a continuous variable capacitance, the phase of individual elements can be accurately tuned without the degradation in directivity and side-lobe levels associated with phase quantization. In this paper, we first introduce definitions and review beamforming theory in Section II. Next, we discuss the element characteristics in Section III, followed by the array design and biasing scheme in Section IV. Section V presents the experimental results of operation as an array lens, and Section VI presents the operation of the array as a reflectarray. Finally, Section VII presents some conclusions of this work. II. COORDINATE SYSTEMS AND BEAMFORMING Consider a lens realized from an planar array with and in the horizontal and element spacing , indexed by be the field produced vertical directions, respectively. Let from the feed illumination at element , and be the phase of that field. In order for the array to create a pencil , a uniform phase front needs to beam in the direction be generated perpendicular to the beamforming direction. of element From array theory, the transmission phase must be Element Phase

(1)

where is the phase shift due to element positioning in the array, is the free-space wave number, and is an integer. From array theory, the predicted far-field pattern (the array factor) with isotropic is elements and element transmission responses

(2) The beamforming phase delays are illustrated in Fig. 1(a), where and denote electrical phase delays. Fig. 1(b) shows the far-field coordinate system used for array factor calculations and beamforming, and is defined in terms of

and . The element orientations and principal polarizations are also shown. This paper focuses on achieving independent element phase control for beamforming, but the investigation is limited to the scanning of pencil beams. For more in-depth discussions on phase shift calculations for arbitrary 2-D beamforming techniques, we refer the reader to [29] and [30]. III. ARRAY ELEMENT DESIGN The array lens/reflectarray, based upon the design presented in [28], consists of two patches placed on either side of a ground plane, coupled by a slot in the ground plane. As shown in Fig. 2, each patch is loaded with two varactor diodes, and another varactor diode is also placed across the slot in the ground plane. Together, the two patches and the loaded slot form three tunable resonators, which can yield 360 of phase tunability with only 3 dB of variation in the transmission magnitude [28]. The top and bottom patches are identical, and are biased with the same voltage. We defer the discussion on the biasing mechanisms to Section IV. This design has the advantage that it only uses two dielectric layers, and does not require any interlayer connections or vias, greatly reducing cost and fabrication complexity. If the array element is placed inside a rectangular waveguide, then from image theory, an infinite array of identical elements is simulated [31]. Thus, we can define two plane wave ports on either side of the element in the waveguide, where the -parameters and describe the offbroadside transmission and reflection characteristics of the element. The element was studied first by full-wave FDTD simulations using SEMCAD-X [32], followed by experimental characterization in waveguide. Fig. 3 shows the measured transmission characteristics of the element for different configurations of the varactor diode bias voltages. For an insertion loss variation of less than 3 dB, the fabricated element achieves 245 of phase tunability. IV. ARRAY DESIGN AND FABRICATION Using this array element, a 6 6 planar array lens was designed and experimentally tested. Preliminary results from the array were presented in [33]. The size of 6 6 was selected so that it could be easily and cost-effectively fabricated, but sufficiently large so that the directivity was high enough to be measured by a planar near-field antenna scanner. The size is selected as a proof-of-concept, but a practical implementation of this

LAU AND HUM: A PLANAR RECONFIGURABLE APERTURE WITH LENS AND REFLECTARRAY MODES OF OPERATION

3549

TABLE I ELEMENT DIMENSIONS AND PARAMETERS

Fig. 2. Array element and bias network design (vertically exaggerated).

observed that magnetic-field directed (along x-axis) bias lines were not excited and did not perturb the fields in the structure if they were sufficiently far away from the patch. On the other hand, resonances were easily excited in the electric-field directed (along y-axis) bias lines. While the optimal solution would be to use resistive bias lines (especially for scaling the design to larger arrays), we have chosen to approximate those using resistive loading at discrete locations along the y-directed lines [34], as shown in Fig. 2. It was determined through simulations that power is best attenuated in the bias lines with resistors. 10 The bias lines in the fabricated array are 0.4 mm thick with 0.3 mm gaps in between them. To accommodate larger array sizes, the width of the bias lines and gaps could be significantly reduced without any problems, since the varactor diodes are reverse biased and do not carry any current. B. Slot Biasing

Fig. 3. Measured S

for optimal configurations from [28].

array design would be significantly larger to capitalize on the advantages of the architecture. To realize larger sizes, the array lens could be assembled in panels, and the bias lines made extremely thin. The element spacing in the array is 30 mm 30 at 5.7 GHz. The dimensions and paramemm, which is ters of the design are shown in Table I. A. Patch Biasing Biasing each element in the array lens is a significant challenge. Unlike reconfigurable reflectarrays, where the biasing network can potentially be placed behind a ground plane, biasing mechanisms are fully exposed to the incident waves from the feeding antenna. Simulations showed that resonances in nonresistive bias lines were easily excited, leading to undesired radiation, coupling, and scattering. As one would expect, it was

The bias network design for the slot varactor was also challenging because for each element, a voltage needed to be developed across a slot in the ground plane, while the ground plane needed to be electrically continuous at high frequencies. Any perturbations in the slot created undesirable results in the transmission response of the element. Furthermore, the use of lumped elements or interdigitated capacitors to create RF continuity but DC isolation was not feasible due to the physical size of the slot and the unit cell. However, since each layer was to be fabricated from substrates with metallization on both sides, two layers of metallization were available to form the ground plane. Thus, as shown in Fig. 2, the 5.0 mm 2.0 mm slot was placed in the lower ground plane, with the two sides of the slot electrically isolated at DC. A large 5.0 mm 5.0 mm rectangular hole was placed in the upper ground plane, which had the same width as the slot, so that the biasing grooves did not electrically extend the slot. The length of the hole was large enough such that a varactor diode could easily be soldered to the lower ground plane. A small cavity milled in the upper substrate (not shown) created room for the slot varactor diode to fit in between the substrates. The two substrate layers were separated by a 0.04 mm Rogers 3001 dielectric bonding film. With the thin film in place, a large capacitance is created between the upper and lower ground planes, resulting in a biased slot that appears electrically continuous at high frequencies, but has sides that are isolated for DC voltages.

3550


analog converter followed by an operational amplifier, and produced a voltage between 0–20 V. B. Characterization

Fig. 4. Fabricated 6

2 6 array.

Fig. 5. Array lens setup and measurement coordinates.

Waveguide tests with elements biased in this way showed that this biasing scheme had minimal effect on the performance [28]. The fabricated 6 6 planar array is shown in Fig. 4. On each substrate, one side had metallization containing the patches, and the other side had metallization containing the ground plane and slots. Since a varactor diode needed to be inserted between the cavity was milled two substrates, a into underside of the upper substrate for each of the cells. All bias lines run to six 40-pin connectors at the edge of the board. V. ARRAY LENS MEASUREMENTS A. Experimental Setup Fig. 5 illustrates the experimental setup along with the coordinate system used for measurements, measured in terms of elevation and azimuth angles. A pyramidal feed horn with a directivity of 17.6 dBi was placed such that the aperture of the horn was 300 mm from the ground plane of the array, which corresponded to an f/D ratio of 1.67. The feed horn was positioned such that the array was prime-focus fed. The feed horn was 180 mm long, with an aperture size of 140 mm 130 mm. Nearfield SysThe entire setup was placed on a planar tems Inc. (NSI) near-field antenna scanner. The patches and slot biases of each element were connected to a series of custom-designed USB voltage controllers. As there were 36 elements, 72 independent voltage channels were required. Three 32-channel voltage controller boards were used to control the elements. Each channel had an 8-bit digital-to-

As it was known from repeated waveguide tests that the elements were somewhat sensitive to fabrication error, and the voltage-capacitance characteristics of the varactor diodes vary from diode to diode, the behavior of each element in the fabricated array needed to be measured. To characterize the array, for each element, the open-ended waveguide (OEWG) probe of the scanner was moved to the broadside position of that element. Measurements were conducted with the waveguide positioned 24 mm away from the element-under-test in the longitudinal direction. A 2-D sweep of the two element control voltages was performed, with the patch voltage varying between 1–20 V, and the slot voltage varying between 4–20 V. As each element was characterized, the bias voltages of all the other elements were kept at zero volts, since it is known from waveguide tests that that zero volt biases make the elements completely reflective. The magnitude and phase received by the probe at that single position is taken to be the transmission response of the element. In this method, the near-field scanner was used only to position the probe, and no near-field scanning is actually performed. We note that while the gain of the OEWG probe is included in the measurements, the absolute magnitude and phase of the characterization is not important. By placing the probe at broadside for each element, the OEWG probe pattern consistently added a constant insertion loss to each measurement. magnitudes and The 2-D voltage sweep produced a set of phases for each configuration. For each element, a number of configurations may produce the same phase shift [28]. Thus, for each phase shift , the configuration that yields the maxis called the optimal configuration for phase shift imum . Ideally, all elements have the same optimal configuration curves, but in reality they differ due to fabrication errors and varactor diode variances. An advantage of having two degrees of freedom of control for each element is that these variances can be accommodated, to a certain degree. The average 3 dB-variation phase range of the fabricated elements is 195 , ranging between 149 and 244 . If the magnitude variation is relaxed to 10 dB, then the average phase range of the elements is 284 , ranging between 205 and 360 . Fig. 6 shows histograms of the phase ranges for the fabricated elements. We note that the variations did not depend upon element position, so the variations cannot be attributed to the feed illumination. Nevertheless, despite the variation in the tuning range between the elements, the characterization shows that all of the elements exhibit a useful tunable phase range for beamforming purposes. C. Measurements The planar near-field antenna scanner was used to measure the near-field response at 5.7 GHz, from which the far-field response was computed. The broadside configuration was tested first, and produced a well-defined pencil beam with an isolation of 18 dB between the co-polarization and cross-polarization, as shown in Fig. 7. Fig. 8 shows the far-field calibrated gain of -plane pencil beams directed from to 25 in elevation


3551

Fig. 6. Phase range histograms for fabricated elements. (a) 3-dB magnitude variation. (b) 10-dB magnitude variation.

Fig. 7. Co- and cross-polarizations in the two principal planes.

Fig. 9. Measured far-field pattern as the azimuth angle is scanned (H -plane). TABLE II PENCIL BEAM DIRECTIVITY, PEAK GAIN, SIDE-LOBE LEVEL, AND BEAMWIDTH

Fig. 8. Measured far-field pattern as the elevation angle is scanned (E -plane).

at zero azimuth angle, which corresponds to , , 270 . Similarly, Fig. 9 shows the far-field gain of -plane pencil beams directed from to 25 in azimuth at zero elevation angle, which corresponds to , , 180 . We note that there is a slight asymmetry beand , which is due to a combitween the beams at nation of element variations and fabrication error. The direc-

tivities (DIR), peak gain (GAIN), side-lobe levels (SLL), and half-power beamwidths (BW) of the pencil beams are summarized in Table II. The 1-dB gain bandwidth of the broadside beam was 280 MHz, or a fractional bandwidth of 4.9%. From the plots, we see that the magnitude response decreases as the beam scans from broadside. A small amount of the reduction can be attributed to the variation in the patch antenna pattern, since at 25 offbroadside, the patch’s antenna pattern has losses of 0.5 dB and 0.9 dB in the -plane and -plane, respectively. However, this behavior is primarily due to the fact that as the scanning angle is increased, the range of required

3552


Fig. 10. Measured aperture fields (array bounds denoted by rectangle). (a) Magnitude (broadside beam). (b) Phase (broadside beam). (c) Magnitude (15 beam). (d) Phase (15 beam).

phase responses in the elements is also increased. Since the elements produce large insertion losses for a range of phase shifts, some elements in the array can effectively become disabled. Thus, as scanning angle is increased, there is a reduction in the overall amount of power transmitted through the array, and the gain is reduced. We see the effect of the nonconstant insertion losses for different desired element phases in Fig. 10, where the near-field measurements are holographically back-projected to yield the fields on the array aperture. For the broadside pencil beam, the phase in Fig. 10(b) is very constant over the entire aperture, and the magnitude in Fig. 10(a) shows the effect of the expected feed in the -plane is taper. When a pencil beam with formed, we see from Fig. 10(d), and from (1), that a full 360 of phase tunability is required. The array manages to create the required phase gradient, but the magnitudes of different elements deteriorate. From Fig. 10(c), we see that certain elements effectively cease to transmit, resulting in reduced gain and directivity. An alternative approach would have been to configure such elements to transmit with a certain amount of phase error but with larger insertion losses. While the overall gain may be improved, phase errors would result in degradation of the directivity and side-lobe levels. D. Discussion The directivity of a hypothetical uniform aperture with 180 mm) is the same dimensions as the array (180 mm . So, the array achieves a directivity at broadside of 20.8 dBi, which is only 0.9 dB less than that of an

ideal aperture. This difference is caused by the feed taper efficiency of 0.8 dB. There is a significant difference between the array’s measured directivity and gain, which is due primarily to low efficiency of the feeding system and element losses. Losses will be discussed in greater detail later. Nevertheless, the measured half-power beamwidths in both principal planes . are very close to the theoretical value of The theoretical first side-lobe level of a uniform aperture, 13.3 dB, is also very close to the side-lobe levels achieved by this array. In fact, the magnitude tapering due to the feed resulted in smaller -plane side-lobes, but at the cost of slightly wider beamwidths. At broadside, the 6 6 array lens demonstrates beamforming performance that is very close to optimal. The excitation of each element can be calculated using the responses. By feed illumination and the measured element combining the excitations with the standard array factor of a planar array (2), and the far-field element pattern of a patch antenna, an expected far-field gain pattern can be calculated. Figs. 11 and 12 show that the expected and measured far-field gain patterns correspond very well, confirming that mutual coupling is not a significant problem, as suggested from experiments in [28]. The ripples were likely a result of diffraction effects at the edge of the array. We can divide the losses into feed system losses and array losses. Feed system losses consist of feed tapering loss, feed horn loss, and spillover loss, and are summarized in Table III. The array losses, namely the resistive loss, specular reflection, and back-scatter, can be estimated using the unit cell experimental data from [28] and near-field feed horn measurements. To form a broadside beam, the bias voltages of element


3553

Fig. 13. Reflectarray setup.

from broadside, the reduction in gain is primarily due to mismatch and loss from the varactor resistance.

Fig. 11. Measured far-field pattern (E -plane).

VI. REFLECTARRAY-MODE OPERATION

Fig. 12. Measured far-field pattern (H -plane). TABLE III ARRAY LENS LOSS BUDGET FOR BROADSIDE BEAM

are tuned to a configuration that gives , where is an integer. The power incident on the array can be es, and the power transmitted through each eletimated by ment by . Then, the total expected loss due to specular reflection, back-scatter, and dissipation through the array can be calculated to be (3)

The loss budget is summarized in Table III and shows that the expected gain is in good agreement with the measured gain. In the other beamforming cases as the beam-scanning angle scans

In addition to array lens operation, this array element is also able to operate as a reflectarray cell, which is a unique feature of this design. When the slot varactor diode is biased with a low voltage ( 1 V), the slot ceases to be resonant, and as a result no power flows through the slot. In this case, the array element operates as a single-pole resonator, and the patch varactor diodes can be used to tune the reflection phase of the element as in [7]. With the capability to operate in both array lens and reflectarray modes, the beam-scanning range is increased to include both the forward and backward directions. The array in a reflectarray setup is shown in Fig. 13. A smaller pyramidal horn with a directivity of 14.1 dBi, and an aperture of 91 mm 64 mm and a length of 163 mm was placed such that the center of the horn aperture was 200 mm from the array aperture plane, and 150 mm below the center of the array. The feed horn was tilted at an angle of 30 downward from the prime-focus position in the -plane. A smaller feed horn was used for reflectarray-mode testing to reduce the effects of feed blockage. Desired element phases for a pencil beam are computed in the same way as the array lens case, using (1). The measured far-field gain patterns for the -plane and -plane are shown in Figs. 14 and 15. In the -plane, beams for positive elevation angles have much larger magnitudes than those for negative elevation angles. This is due to the effect of feed blockage. In the -plane, the array achieves high-directivity beams in azto 30 , with a directivity of 19.4 dBi at imuth angles from increases, broadside. However, as the beam-scanning angle the side-lobe levels increase from 16.3 dB at broadside to 5.0 . We refer to [7] for details on how the loss budget dB at is calculated for a reconfigurable reflectarray of this type. In comparison to array lens beamforming patterns shown in Figs. 8 and 9, scan angle has a much smaller effect on gain. This is because as the scan angle increases, the required phase range of the elements increases. Since the reflection phase range of the array elements is larger, demonstrated to be over 300 in [28], the number of elements with large phase errors is reduced. Given that the peak gains of both the array lens and reflectarray are about the same, we conclude that the transmission and reflection efficiencies of the array are similar.

3554


Fig. 14. Reflectarray measured far-field pattern (E -plane).

same size and feeding configuration, there is 4.8 dB of loss, which is attributed to resistive losses in the varactor diodes, specular reflection, and back-scatter. The far-field beam patterns were shown to correspond very well to predicted results. In addition to array lens operation, the array was also able to function as a reflectarray, achieving a directivity of 19.4 dBi at . The versatility and broadside and a scanning range of ease of fabrication make this array lens design very attractive for applications where high-gain beam-scanning is needed. We note that due to distortion, varactor diodes are only suitable for low-power operation, namely receive-mode operation. Furthermore, the resistive loss in the varactor diodes presents a drawback to practical deployment, but is primarily a technological issue. In order to support high-power or transmit applications with lower losses, MEMS capacitors could be used in place of the varactor diodes [9]. Future work will focus on improving the phase tuning range of the unit elements, which is limited by the loss in the varactor diodes in the current design. While it is unlikely that the resistive loss of the diodes can be eliminated, different positioning of the diodes may improve performance. We will also aim to reduce the two control voltages to one voltage per element, since biasing complexity increases rapidly as array size increases. Finally, the possibility of increasing the phase agility of the element will be explored. REFERENCES

Fig. 15. Reflectarray measured far-field pattern (H -plane).

An interesting observation is that in Fig. 14 a substantial lobe at an elevation angle of 30 is observed at all scanning angles. These lobes can be attributed to specular reflection from the feed, indicating that the array elements have reduced ability to couple power well at higher angles of incidence. With the ability to function as both as a lens and a reflector, this design can be used in more sophisticated configurations, such as a conformal array where some parts of the array are operating as reflectors and other parts as transmitters. VII. CONCLUSION A fully reconfigurable 6 6 array lens has been experimentally demonstrated at 5.7 GHz. Requiring only two microstrip layers with no interlayer connections, this design is a low-cost and easy-to-fabricate candidate for high-gain reconfigurable beamforming. Furthermore, this design achieves a tunable three-pole response with an array thickness of about one-tenth of a wavelength. The array lens achieved a directivity of 20.8 dBi at broadside and was able to electronically beam-scan over range. When compared to an ideal aperture of the a

[1] D. M. Pozar, S. D. Targonski, and H. D. Syrigos, “Design of millimeter wave microstrip reflectarrays,” IEEE Trans. Antennas Propag., vol. 45, no. 2, pp. 287–296, Feb. 1997. [2] J. Huang and J. A. Encinar, Reflectarray Antennas. Piscataway, NJ: IEEE Press, 2007. [3] A. Tombak and A. Mortazawi, “A novel low-cost beam-steering technique based on the extended-resonance power-dividing method,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 2, pp. 664–670, Feb. 2004. [4] L. Schulwitz and A. Mortazawi, “A compact dual-polarized multibeam phased-array architecture for millimeter-wave radar,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 11, pp. 3588–3594, Nov. 2005. [5] R. Sorrentino, R. Gatti, L. Marcaccioli, and B. Mencagli, “Electronic steerable MEMS antennas,” in Proc. Eur. Conf. Antennas Propag., Nov. 2006. [6] R. Sorrentino, R. V. Gatti, and L. Marcaccioli, “Recent advances on millimetre wave reconfigurable reflectarrays,” in Proc. Eur. Conf. Antennas Propag., Mar. 2009. [7] S. V. Hum, M. Okoniewski, and R. J. Davies, “Modeling and design of electronically tunable reflectarrays,” IEEE Trans. Antennas Propag., vol. 55, no. 8, pp. 2200–2210, Aug. 2007. [8] M. Riel and J.-J. Laurin, “Design of an electronically beam scanning reflectarray using aperture-coupled elements,” IEEE Trans. Antennas Propag., vol. 55, no. 5, pp. 1260–1266, May 2007. [9] S. V. Hum, G. McFeetors, and M. Okoniewski, “Integrated MEMS reflectarray elements,” in Proc. Eur. Conf. Antennas Propag., Nov. 2006. [10] J. Perruisseau-Carrier and A. K. Skrivervik, “Monolithic MEMS-based reflectarray cell digitally reconfigurable over a 360 phase range,” IEEE Antennas Wireless Propag. Lett., vol. 7, pp. 138–141, 2008. [11] C. Apert, T. Koleck, P. Dumon, T. Dousset, and C. Renard, “ERASP: A new reflectarray antenna for space applications,” in Proc. Eur. Conf. Antennas Propag., Nov. 2006. [12] H. Rajagopalan, Y. Rahmat-Samii, and W. A. Imbriale, “RF MEMS actuated reconfigurable reflectarray patch-slot element,” IEEE Trans. Antennas Propag., vol. 56, no. 12, pp. 3689–3699, Dec. 2008. [13] R. W. Clark, G. H. Huff, and J. T. Bernhard, “An integrated active microstrip reflectarray element with an internal amplifier,” IEEE Trans. Antennas Propag., vol. 51, no. 5, pp. 993–999, May 2003. [14] K. Kishor and S. Hum, “Modeling of active reconfigurable reflectarray elements,” in Proc. Int. Symp. Antenna Technol. Appl. Electromagn. Can. Radio Sci. Meeting, Feb. 2009.


[15] D. M. Pozar, “Flat lens antenna concept using aperture coupled microstrip patches,” Electron. Lett., vol. 32, no. 23, pp. 2109–2111, Nov. 1996. [16] C. G. M. Ryan, J. R. Bray, Y. M. M. Antar, M. R. Chaharmir, J. Shaker, and A. Ittipiboon, “A broadband transmitarray using double square ring elements,” in Proc. Int. Symp. Antenna Technol. Appl. Electromagn. Can. Radio Sci. Meeting, Feb. 2009. [17] Z. Popovic and A. Mortazawi, “Quasi-optical transmit/receive front ends,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 11, pp. 1964–1975, Nov. 1998. [18] A. B. Yakovlev, S. Ortiz, M. Ozkar, A. Mortazawi, and M. B. Steer, “A waveguide-based aperture-coupled patch amplifier array-full-wave system analysis and experimental validation,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 12, pp. 2692–2699, Dec. 2000. [19] S. C. Ortiz, J. Hubert, L. Mirth, E. Schlecht, and A. Mortazawi, “A high-power Ka-band quasi-optical amplifier array,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 2, pp. 487–494, Feb. 2002. [20] M. P. DeLisio and R. A. York, “Quasi-optical and spatial power combining,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 3, pp. 929–936, Mar. 2002. [21] M. E. Bialkowski and H. J. Song, “A Ku-band active transmit-array module with a horn or patch array as a signal launching/receiving device,” IEEE Trans. Antennas Propag., vol. 49, no. 4, pp. 535–541, Apr. 2001. [22] P. P. de la Torre and M. Sierra-Castaner, “Design of a 12 GHz transmit-array,” in Proc. IEEE Int. Symp. Antennas Propag., Jun. 2007, pp. 2152–2155. [23] A. Abbaspour-Tamijani, K. Sarabandi, and G. M. Rebeiz, “A millimetre-wave bandpass filter-lens array,” IET Microw. Antennas Propag., vol. 1, no. 2, pp. 388–395, Apr. 2007. [24] W. Hu, R. Dickie, R. Cahill, H. Gamble, Y. Ismail, V. Fusco, D. Linton, N. Grant, and S. Rea, “Liquid crystal tunable mm wave frequency selective surface,” IEEE Microw. Wireless Compon. Lett., vol. 17, no. 9, pp. 667–669, Sep. 2007. [25] F. Bayatpur and K. Sarabandi, “Tuning performance of metamaterialbased frequency selective surfaces,” IEEE Trans. Antennas Propag., vol. 57, no. 2, pp. 590–592, Feb. 2009. [26] C.-C. Cheng, B. Lakshminarayanan, and A. Abbaspour-Tamijani, “A programmable lens-array antenna with monolithically integrated MEMS switches,” IEEE Trans. Microw. Theory Tech., vol. 57, no. 8, pp. 1874–1884, Aug. 2009. [27] P. Padilla, A. Munoz-Acevedo, M. Sierra-Castaner, and M. Sierra-Perez, “Electronically reconfigurable transmitarray at Ku band for microwave applications,” IEEE Trans. Antennas Propag., vol. 58, no. 8, pp. 2571–2579, Aug. 2010. [28] J. Y. Lau and S. V. Hum, “Analysis and characterization of a multipole reconfigurable transmitarray element,” IEEE Trans. Antennas Propag., vol. 59, no. 1, Jan. 2011. [29] O. M. Bucci, G. Mazzarella, and G. Panariello, “Reconfigurable arrays by phase-only control,” IEEE Trans. Antennas Propag., vol. 39, no. 7, pp. 919–925, Jul. 1991. [30] A. K. Bhattacharyya, “Projection matrix method for shaped beam synthesis in phased arrays and reflectors,” IEEE Trans. Antennas Propag., vol. 55, no. 3, pp. 675–683, Mar. 2007.

3555

[31] P. Hannan and M. Balfour, “Simulation of a phased-array antenna in waveguide,” IEEE Trans. Antennas Propag., vol. 13, no. 3, pp. 342–353, May 1965. [32] SPEAG, SEMCAD X 2010 [Online]. Available: http://www. speag.com [33] J. Y. Lau and S. V. Hum, “Design and characterization of a 6 6 planar reconfigurable transmitarray,” in Proc. Eur. Conf. Antennas Propag., Apr. 2010. [34] C. Mias and J. H. Yap, “A varactor-tunable high impedance surface with a resistive-lumped-element biasing grid,” IEEE Trans. Antennas Propag., vol. 55, no. 7, pp. 1955–1962, Jul. 2007.

2

Jonathan Yun Lau (S’08) was born in Calgary, Alberta, Canada. He received three B.Sc. degrees in computer engineering, psychology, and applied mathematics from the University of Calgary, Alberta, Canada, in 2005, the M.A.Sc. degree in electrical and computer engineering from the University of Toronto, Toronto, ON, Canada, in 2007, and is currently pursuing the Ph.D. degree in the Electromagnetics Group, University of Toronto. Dr. Lau received the APEGGA gold medal award for his undergraduate achievements in 2005. He has received a number of scholarships for his graduate work including the NSERC Canada Graduate Scholarship, Bell University Labs Scholarship, and Ontario Graduate Scholarship in Science and Technology. His research interests are in the areas of transmitarrays and reconfigurable antenna systems.

Sean Victor Hum (S’95–M’03) was born in Calgary, Alberta, Canada. He received the B.Sc., M.Sc., and Ph.D. degrees from the University of Calgary in 1999, 2001, and 2006, respectively. In 2006, he joined the Edward S. Rogers Sr. Department of Electrical and Computer Engineering, University of Toronto, Toronto, ON, Canada. where he currently serves as an Assistant Professor. His present research interests lie in the area of reconfigurable RF antennas and systems, antenna arrays, and ultrawideband communications. Dr. Hum received the Governor General’s Gold Medal for his master’s degree work on radio-on-fiber systems in 2001. In 2004, he received a IEEE Antennas and Propagation Society Student Paper award for his work on electronically tunable reflectarrays. In 2006, he received an ASTech Leaders of Tomorrow award for his work in this area. He is also the recipient of three teaching awards. He has served on the steering committee and technical program committee for the 2010 IEEE AP-S International Symposium on Antennas and Propagation. In August 2010, he was appointed as an Associate Editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION.