IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 24, NO. 5, MAY 2014
297
A Reconfigurable Bandpass Filter Based on a Varactor-Perturbed, T-Shaped Dual-Mode Resonator Hsuan-Ju Tsai, Bo-Chih Huang, Nan-Wei Chen, Member, IEEE, and Shyh-Kang Jeng
Abstract—A microstrip bandpass filter (BPF) with high centerfrequency tunability, bandwidth reconfigurability, and a one-side band edge transmission zero (TZ) is presented. The reconfigurable BPF is developed with a simple T-shaped dual-mode resonator, which is externally coupled and perturbed with varactor diodes. Also, the design of adjustable admittance inverters for tunable resonators external coupling is proposed to mitigate the in-band performance degradation that hurdles the ones with a fixed coupling. It is demonstrated that the 1 dB bandwidth is able to be varied from 55 to 175 MHz for center frequency ranging from 750 to 1240 MHz with a low insertion loss ( 2.9 dB). Index Terms—Dual mode, reconfigurable filter, resonator, tunable filter, varactor.
I. INTRODUCTION
R
ECONFIGURABLE/tunable bandpass filters (BPFs) have attracted great attentions owning to their applications to reduce the complexity in modern communication systems [1]. According to frequency agility, BPFs of this kind can be categorized into three types: Type-I has a passband with a fixed center frequency and a tunable bandwidth [2]; Type-II features tunable center frequency with a constant passband bandwidth [3]–[5]; Type-III has a passband with a tunable center frequency as well as a controllable bandwidth [1], [6]–[9]. The adopted structures and approaches for the above-referred filter designs are outlined as follows. In [2], Type-I BPFs are synthesized with the varactor-tuned transmission-line (TL) structures of two separately relocatable band-edge transmission zeros (TZs) for bandwidth adjustment. As for Type-II filter design, the dual-mode open-loop resonator with varactor-controlled mode coupling is adopted for center frequency tuning [3], and the selectivity can be further enhanced via an introduction of a TZ on the lower/higher band edge. However, the in-band response, particularly the return loss, is affected while the center frequency varies. To obtain tunability on center frequency as well as bandwidth, the dual-mode circular ring resonator [6], the dual-mode triangular patch Manuscript received September 26, 2013; accepted November 20, 2013. Date of publication March 04, 2014; date of current version May 06, 2014. This work was supported in part by the National Science Council of Taiwan, under contract NSC 102-2221-E155-016. H.-J. Tsai and S.-K. Jeng are with the Department of Electrical Engineering and the Graduate Institute of Communication Engineering, National Taiwan University, Taipei 10617, Taiwan (e-mail:
[email protected];
[email protected];
[email protected]). B.-C. Huang and N.-W. Chen are with the Department of Communications Engineering, Yuan Ze University, Jhongli 32003, Taiwan (e-mail:
[email protected];
[email protected]). Color versions of one or more of the figures in this letter are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/LMWC.2014.2306893
Fig. 1. (a) Conventional 2nd-order capacitively fed BPF in a symmetric circuit topology. (b) An equivalent circuit of the one in Fig. 1(a) with the absorpinto . (c) The proposed reconfigurable BPF in a TL circuit tion of representation.
[7], and the dual-mode loop-shaped resonator [8] perturbed with varactors were employed for Type-III BPF realization. It is shown that the even/odd mode resonance varies with the varactors for center frequency and bandwidth tuning because they are under the control of the mode coupling. Apart from the dual-mode resonators, the microstrip combline embedded with varactor-tuned coupling has been presented in [9]. In this work, a varactor-perturbed dual-mode T-shaped resonator is proposed for synthesis of a Type-III filter with a higher frequency agility. The dual-mode resonator is perturbed with varactors and externally coupled with adjustable admittance inverters. With a specified passband response, the center frequency together with bandwidth of the BPF can be altered with the incorporated varactors via a perturbation of the even/odd mode resonance. Moreover, a TZ can be introduced at the lower/higher passband edge for one-side selectivity enhancement without in-band response degradation observed in [3]. In what follows, the proposed design approach described in a circuit perspective is presented in Section II. The experimental verification is outlined in Section III. II. FILTER DESIGN AND CHARACTERIZATION Fig. 1(a) shows the equivalent circuit of a capacitively-fed, 2nd-order BPF prototype in a symmetric network representais the system characteristic admittance, tion. In Fig. 1(a), and are the admittance inverters, and denotes the is realized with a series capacitor shunt resonator. Here, and a shunt capacitor of negative capaciof capacitance . Assuming that resonates at the filter center tance
1531-1309 © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
298
IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 24, NO. 5, MAY 2014
Fig. 2. Dual-mode circuits for solving the resonant frequencies: (a) The evenmode circuit and (b) the odd-mode circuit.
frequency, , the corresponding spectively, expressed as [8]
In (1), the admittance inverter
and
are, re-
(1)
Fig. 3. Detailed layout and dimensions of the proposed BPF with bias circuitry (unit: mm).
(2)
as follows. First, and with a given inequality ( or ) are simply determined with and . Next, the admittance inverter is obtained using (3) where
is parameterized as
(10) (3)
is the susceptance slope parameter of and the where where and are external quality factor respectively the first and second element values of a 2nd-order is the BPF fractional bandlowpass filter prototype, and into width. By absorbing the negative capacitance , Fig. 1(b) shows the resulting network employed as the proposed BPF prototype. Fig. 1(c) presents the corresponding TL circuit representation of the proposed BPF. In Fig. 1(c), all and (subscript capacitors have variable capacitance, and would be or ) are the characteristic admittance and the electrical length of the TL sections, respectively. By taking advantage of the circuit symmetry with respect to the plane denoted in Fig. 1(c), the entire BPF can be characterized using the even- and odd-mode analysis. Specifically, Fig. 2(a) and (b) show the corresponding even- and odd-mode equivalent circuits. With these circuits, it is straightforward to find the even- and odd-mode resonant frequencies, i.e., and , via imposing zero input admittance condition. Here, is written as the even-mode input admittance (4) In (4),
, and (5)
In (5) and (6), and Similarly, the odd-mode input admittance
(6) . is
(7) . As for the filter parameters, where the center frequency and the coupling coefficient can be respectively estimated using (8) and (9) [6] (8) (9) To synthesize the proposed BPF with the specified , , and passband response, the corresponding parameters are obtained
However, the evaluation of calls for since and are functions of . To tackle this difficulty, an iterative , scheme is proposed. Specifically, with an initial guess of in (10) is calculated, and in (3) is then obtained. Next, in is checked for satis(1) is evaluated, and the convergence of obtained via (2) faction of three conditions: the capacitance is closed enough to that calculated in the previous iteration, the constraints on with (8), and the constraint on with (9). Afterand are determined with the converged parameters. wards, . SpecifOn the other hand, the TZ occurs when ically, without affecting the passband response, the TZ can be specified at the lower or higher passband edge by letting the ) input impedance ( ) looking into the TL section ( , equals to zero. Also, the lower/higher band-edge TZ occurs is less/greater than [4]. when III. FILTER SIMULATION AND MEASUREMENT The filter is designed on a Rogers 3006 substrate with , , and a thickness of 1.28 mm. The filter is characterized with the commercial full-wave solver, HFSS. In simulation, the device models of Skyworks high- varactor capacitors SMV1413 (1.8–9.2 pF) and SMV1235 (2.4–18 pF) are respectively adopted to mimic the variable capacitance (subscript would be or ) and . Note that the larger tuning is exploited to achieve capacitance ratio associated with filter compactness and the introduction of TZ at the lower/higher pass-band edge. On the other hand, all varactors are employed in an anti-series configuration with a concern on varactor biasing and linearity [7]. To demonstrate the capability of the filter, the design of the BPF with Chebyshev passband response of a ripple of 0.14 dB (i.e. return loss of 15 dB) is synthesized. , The corresponding TL parameters are determined as , , and , both at 750 MHz. The filter layout together with bias circuitry is shown in Fig. 3. For experimental verification, the photograph of the fabricated filter along with the bias copper wires is displayed in Fig. 4. Note that the filter input and output ports are connected with bias tees for dc blocking and proper varactor biasing at the pad . With a demonstration of center frequency and bandwidth parameters of the tuning, the measured and simulated BPF with TZ at the lower/higher band edge are shown in Figs. 5(a) and (b), respectively. More specifically, the first four cases together show the tuning of the center frequency and bandwidth of the BPF with a lower band-edge TZ, while the
TSAI et al.: RECONFIGURABLE BPF BASED ON A VARACTOR-PERTURBED, T-SHAPED DUAL-MODE RESONATOR
299
TABLE I , IL, , AND CORRESPONDING MEASURED VARACTOR BIAS, , 1 dB BANDWIDTH FOR DIFFERENT CASES SHOWN IN FIG. 5
Fig. 4. Photograph of the fabricated filter. TABLE II COMPARISON OF THIS WORK WITH OTHERS
Fig. 5. Demonstration of the instantaneous and tuning when TZ is located at (a) the lower passband edge or (b) the higher passband edge (solid: measurements; dashed: simulations).
Fig. 6. Simulated (dashed lines) and measured (solid lines with outlined circles) maximum 1 dB bandwidth and its corresponding IL as a function of (the lines in gray represent the responses with a lower band-edge TZ illustrated in the inset at upper left corner; the lines in black represent the responses with a higher band-edge TZ illustrated in the inset at upper right corner).
other four cases exhibit the ones with a higher band-edge TZ. As a result, the demonstrated BPF shows no in-band response less than 14 dB. Table I tabdegradation with in-band ulates the corresponding bias voltages, the insertion loss (IL), the input third-order intercept points (IIP3s), and its measured 1 dB bandwidth. On the other hand, the simulated and measured mid-band IL and the maximum 1 dB bandwidth with respect are presented in Fig. 6 to demonstrate the proposed to filter performance. In Fig. 6, the results associated with the lower/higher band-edge TZ response are respectively presented in gray/black colored lines. As a result, the 1 dB bandwidth is able to be varied from 55 to 175 MHz for ranging from 750 to 1240 MHz. Finally, the comparison of this filter performance against related past works is summarized in Table II. Note that the tuning ratio of corresponds to the percentage of the ratio of its tuning range to its averaged value. Also, the additional parameter descriptions are footnoted at the bottom of Table II. From this table, we can see that our work almost outperforms the other designs, and it is more balanced in all requirements.
#: 3 dB bandwidth, ##: 1 dB bandwidth, : Filter with a finite TZ switchable capability at the lower/higher passband edge, : The overlapped regions with the lower/higher TZ.
IV. CONCLUSION In conclusion, a reconfigurable dual-mode microstrip BPF with high center frequency and bandwidth tenability has been proposed and demonstrated. With this design, the lower/higher band-edge TZ can be introduced for one-side selectivity enhancement. It is demonstrated that the passband 1 dB bandwidth of the BPF can be varied from 55 to 175 MHz with the center frequency ranging from 750 to 1240 MHz. Also, the demonstrated BPF shows no in-band response degradation with in-band less than 14 dB and IL less than 2.9 dB. REFERENCES [1] C. Lugo and J. Papapolymerou, “Six-State Reconfigurable Filter Structure for Antenna Based System,” IEEE Trans. Antennas Propag., vol. 54, no. 2, pp. 479–483, Feb. 2006. [2] H.-J. Tsai, N.-W. Chen, and S.-K. Jeng, “Reconfigurable bandpass filter with separately relocatable passband edge,” IEEE Microw. Wireless Compon. Lett., vol. 22, no. 11, pp. 559–561, Nov. 2012. [3] Y.-H. Chun and J.-S. Hong, “Electronically reconfigurable dual-mode microstrip open-loop resonator filter,” IEEE Microw. Wireless Compon. Lett., vol. 18, no. 7, pp. 449–451, July 2008. [4] W. Tang and J.-S. Hong, “Varactor-tuned dual-mode bandpass filters,” IEEE Trans. Microw. Theory Tech., vol. 58, no. 8, pp. 2213–2219, Aug. 2010. [5] L. Athukorala and D. Budimir, “Compact second-order highly linear varactor-tuned dual-mode filters with constant bandwidth,” IEEE Trans. Microw. Theory Tech., vol. 59, no. 9, pp. 2214–2220, Sep. 2011. [6] M.-F. Lei and H. Wang, “An analysis of miniaturized dual-mode bandpass filter structure using shunt-capacitance perturbation,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 6, pp. 861–867, Mar. 2005. [7] A. L. C. Serrano, F. S. Correra, T.-P. Vuong, and P. Ferrari, “Synthesis methodology applied to a tunable patch filter with independent frequency and bandwidth control,” IEEE Trans. Microw. Theory Tech., vol. 60, no. 3, pp. 484–493, Mar. 2012. [8] H.-J. Tsai, N.-W. Chen, and S.-K. Jeng, “Center frequency and bandwidth controllable microstrip bandpass filter design using loop-shaped dual-mode resonator,” IEEE Trans. Microw. Theory Tech., vol. 61, no. 10, pp. 3590–3600, Oct. 2013. [9] Y.-C. Chiou and G. M. Rebeiz, “A quasi elliptic function 1.75–2.25 GHz 3-pole bandpass filter with bandwidth control,” IEEE Trans. Microw. Theory Tech., vol. 60, no. 2, pp. 244–249, Feb. 2012.
