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IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, VOL. 11, 2012
A Small and Lightweight Antenna for Handheld RFID Reader Applications Jae-Hoon Bang, Chinzorig Bat-Ochir, Hyung-Suk Koh, Eun-Jong Cha, and Bierng-Chearl Ahn
Abstract—This letter presents a circularly polarized antenna suitable for handheld radio frequency identification (RFID) reader applications. The proposed antenna consists of four meandered monopole elements fed by a series feed network, each segment having an equal magnitude and a successive 90 phase difference. The monopole elements and the feed network are created on two separate printed circuit boards. Four coaxial lines are used to connect the feed network to the monopoles. Short-circuited tuning stubs are employed for impedance matching. A prototype antenna mm and weighing 13 g is designed, measuring fabricated, and tested. Measurements show that the fabricated antenna has at 919 MHz a gain of 2.2 dBic, a 3-dB gain beamwidth of 150 , and a 3-dB axial-ratio beamwidth of 146 . Index Terms—Circular polarization, lightweight antenna, radio frequency identification (RFID) reader, small antenna.
I. INTRODUCTION
R
ADIO frequency identification (RFID) systems in the UHF band are widely used for such diverse applications as logistics, manufacturing, sales, and tracking [1]. An RFID system consists of a transponder or a tag and an either portable or fixed reader. An RFID reader antenna is required to radiate a circularly polarized (CP) wave because the linearly polarized tag can be oriented arbitrarily [2]. In addition to the CP requirement, an antenna for handheld RFID applications needs to be small and lightweight. There has been a pressing need to replace the heavier 130–140-g ceramic patches widely used in commercial handheld RFID readers, which is reflected in the intense research efforts that have been carried out in recent years [3]–[7]. Basically, there are two major approaches used to create small and lightweight CP antennas. The first is to use two crossed dipoles of reduced size fed by a coaxial cable [3], [4]. The 90 phase difference is obtained by the so-called “self-phasing” method where the lengths of the two dipoles are slightly different so that the dipole currents have a 90 phase difference. Manuscript received June 08, 2012; revised July 23, 2012; accepted August 28, 2012. Date of publication September 06, 2012; date of current version October 01, 2012. This work was supported by the Basic Science Research Program through the National Research Foundation of Korea (NRF) funded by the Ministry of Education, Science and Technology (2012-0000479) and the Korea Healthcare Technology R&D Project, Ministry of Health & Welfare, Republic of Korea, under Grant A100054. J.-H. Bang and B.-C. Ahn are with the Department of Radio and Communications Engineering, Chungbuk National University, Cheongju City 361-763, Korea (e-mail:
[email protected]). C. Bat-Ochir and H.-S. Koh are with MAC Technologies, Inc., Ochang 363883, Korea. E.-J. Cha is with the Department of Medicine, Chungbuk National University, Cheongju City 361-763, Korea. Color versions of one or more of the figures in this letter are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/LAWP.2012.2217311
To obtain symmetric currents on the two halves of the dipole, a balun may need to be incorporated. The drawbacks regarding this configuration include the bidirectional radiation and the difficulty in further miniaturization. A unidirectional radiation can be obtained by placing a ground plane on one side of the antenna. This, however, makes the impedance matching difficult and causes the axial-ratio performance to degrade significantly. The second approach is to use four inverted-F miniaturized monopoles in a meandered or spiral configuration [5], [6]. The monopoles are placed at 90 intervals around a circular or square substrate and excited in equal amplitude and successive 90 phase delays using a suitably designed feed network, such as a parallel or Wilkins on power divider, a series power divider, or a transmission-line or lumped-element quadrature hybrid. This structure achieves a better degree of miniaturization and a unidirectional radiation at the cost of a more complicated structure. Other types of antennas that have been studied for handheld RFID readers include a patch perturbed by slits and circular slots [7], a three-element printed Yagi antenna [8], and a helical antenna [9]. In this letter, we present a small and lightweight CP antenna suitable for handheld UHF reader applications. The proposed structure is based on the scheme proposed by Huchard et al. [5], where four nonmeandered printed inverted-F antennas are fed by a combination of three quadrature hybrids. The feed network presented in this letter is based on the meandered series feed network described in [10]. Four inverted-F monopoles with a high degree of meandering are employed for size reduction. Short coaxial lines are used to connect the feed network to the monopole elements as well as to support the upper substrate containing the monopoles. Tuning stubs are placed on the upper substrate together with the monopole elements and short-circuited to the outer conductor of the coaxial line. This approach enables a further size reduction and a mechanically strong structure that sufficiently endures the shock force experienced when the reader is dropped onto hard surfaces. The design of the proposed antenna is presented, followed by the fabrication, measurement, and analysis of the test results. II. ANTENNA DESIGN Fig. 1 shows the structure of the proposed antenna. The antenna consists of four radiating elements on the upper substrate, a series-type feed network on the lower substrate, and four coaxial lines placed between the two substrates. The upper and lower substrates have the same dimensions. The monopole radiating elements are excited in equal magnitude and have successive 90 phase delays to obtain a right-hand circularly polarized (RHCP) radiation.
1536-1225/$31.00 © 2012 IEEE
BANG et al.: SMALL AND LIGHTWEIGHT ANTENNA FOR HANDHELD RFID READER APPLICATIONS
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Fig. 1. Structure of the proposed antenna. Fig. 3. Series feed network.
Fig. 2. Meandered monopole element.
The monopole lengths are greatly reduced by meandering the strip in the manner shown in Fig. 1. The inner boundary of the meandering is a circle with diameter . One may use a square boundary as well. The degree of meandering is dictated by the size and performance requirements. In reported results from studies on small RFID antennas, the antenna footprint can range from to mm , and the height from 1.6 to 17 mm. In the proposed structure, we were able to realize a mm antenna with a tag-read performance better by a few tens of centimeters than that of a ceramic patch with the same footprint. The maximum achieved distance is about 3 m, which has been verified by repeated tests in realistic environments. The proposed antenna was designed using the following steps. First, with the overall antenna size given, a single monopole element with a short-circuited tuning stub was designed. Next, a microstrip feed network was designed that would properly excite the radiating elements. Then, four identical radiating elements were connected to the feed network and lengths of radiating elements simultaneously adjusted to compensate for mutual coupling. Finally, the line width of each tuning stub was individually adjusted to improve the impedance matching. The widely used Microwave Studio ver. 2011 by CST was used in the design of the proposed antenna. The first step in the design involves the meandered monopole element shown in Fig. 2. The monopole is implemented on a 0.92-mm-thick substrate without a backside metallization. In Fig. 2, the dielectric material of the upper substrate is removed for clarity. The monopole is connected to the feed network via a 100- coaxial line. A short-circuited tuning stub is connected
Fig. 4. Designed feed network performance.
between the inner and outer conductors of the coaxial line. The widths of the meandered line and the tuning stub are 1 mm. The lengths of the meandered line and the tuning stub were simultaneously adjusted to have the lowest reflection at 919 MHz, which turns out to have been and , respectively. The second step involves the design of the feed network shown in Fig. 3. The series power-divider topology from [10] is employed, where the input is equally divided into the four outputs with successive 90 shifts. Meandering is applied for size reduction to the quarter-wavelength lines between the ports, whose characteristic impedances are denoted in Fig. 3. The right-angle bends are rounded in order to reduce reflections. The feed network is created on a mm substrate with a dielectric constant of 4.30, a loss tangent of 0.002, and a thickness of 0.92 mm. Fig. 4 shows the performance of the designed feed network. The magnitude of the transmission coefficient was dB, and the phase difference was over the 900–940 MHz range. In the third step of the antenna design, the feed network on the lower substrate is connected to the four radiating elements on the upper substrate using short coaxial lines. The mutual coupling effect is compensated for by adjusting the element
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IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, VOL. 11, 2012
Fig. 8. Fabricated antenna. Fig. 5. Antenna reflection coefficient versus the monopole length controlled . by
Fig. 9. Reflection coefficient of the fabricated antenna. Fig. 6. Antenna performance versus the antenna height.
0.70, 1.00, 0.70, and 1.00 mm, respectively. This completes the theoretical design of the antenna, the results of which were compared to the measurements discussed in Section III. III. ANTENNA FABRICATION AND MEASUREMENT
Fig. 7. Improvement in the antenna reflection coefficient by an optimized design for the tuning stubs (Solid line: tuning stub widths at 0.7, 1.0, 0.7, and 1.0 mm, respectively. Dashed line: all tuning stub widths at 0.7 mm. Dotted line: all tuning stub widths at 1.0 mm).
lengths, which is done by modifying from Fig. 1. Fig. 5 shows the antenna reflection coefficients versus the . With a of 34 mm, the radiating elements resonate at 919 MHz. The ground-to-monopole distance is initially set at 7 mm. The efon the antenna performance was studied; the results fect of are shown in Fig. 6. The best overall performance was obtained when was 9.0 mm. In the final step, the antenna impedance matching is further improved by individually adjusting the line width of the tuning stub. Fig. 7 shows the results. The best performance was obtained when line widths of the tuning stubs for ports 2–5 were
The designed antenna was fabricated using standard printedcircuit processes. Coaxial lines were soldered onto the lower and upper substrates. Fig. 8 shows the fabricated antenna with a coaxial cable connected to the backside of the antenna for measurements. The weight of the fabricated antenna was 13 g excluding the test cable. The performance of the fabricated antenna was measured using a microwave network analyzer (HP 8720C) and a spherical far-field measurement facility. Fig. 9 shows the reflection coefficient of the fabricated antenna. The measured reflection coefficient was less than 10 dB over 912–927 MHz. The measured and simulated performances agreed well. Figs. 10 and 11 show the measured gain and axial ratio on the -axis versus frequency, respectively. The fabricated antenna had a gain of 0.0–2.4 dBic over 909–927 MHz and an axial ratio of less than 3.0 dB at 906–921 MHz. Fig. 12 shows the axial ratio pattern on the -plane at 919 MHz. The axial ratio was smaller than 3.0 dB over 146 in the upper hemisphere. A similar axial ratio performance was observed on the -plane. Finally, Fig. 13 shows the RHCP gain patterns of the fabricated antenna. The gain pattern on the azimuth plane ( -plane) was omnidirectional within 5.5 dB, as shown in Fig. 13(a). Fig. 13(b) shows the elevation pattern on the -plane. The elevation gain pattern on the -plane was similar to Fig. 13(b).
BANG et al.: SMALL AND LIGHTWEIGHT ANTENNA FOR HANDHELD RFID READER APPLICATIONS
Fig. 10. Fabricated antenna’s gain on the -axis versus frequency.
Fig. 11. Fabricated antenna’s axial ratio on the -axis versus frequency.
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Fig. 13. RHCP gain patterns of the fabricated antenna at 919 MHz on (a) the -plane and (b) the -plane (Solid line: simulation. Dashed line with o: measurement).
made up of a two-layer structure where the monopole elements and the feed network have been realized on separate substrates and connected by coaxial lines. The antenna size has been reduced by employing meandered lines in the monopoles and feed network; the weight has been minimized by using thin printed-circuit boards. The fabricated antenna measuring mm and weighing 13 g has exhibited the gain of 2.2 dBic and the gain and axial-ratio beamwidths of 150 and 146 , respectively, at 919 MHz. The proposed antenna features small size, light weight, and good performances in gain, axial ratio, and front-to-back ratio and shows that it can replace the heavier ceramic patch currently used in handheld UHF RFID readers. REFERENCES
Fig. 12. Axial ratio pattern on the 919 MHz.
-plane of the fabricated antenna at
The antenna’s maximum gain was 2.2 dBic at 919 MHz. The 3-dB beamwidth was 150 . The front-to-back ratio was about 10 dB. IV. CONCLUSION A small and lightweight antenna has been proposed for use in handheld RFID reader applications. Four printed meandered monopole elements and a series feed network have been employed to generate a circular polarization. The antenna has been
[1] K. Finkenzeller, RFID Handbook, 2nd ed. New York: Wiley, 2004. [2] J. Uddin, M. B. I. Reaz, M. A. Hasan, A. N. Nordin, M. I. Ibrahimy, and M. A. M. Ali, “UHF RFID antenna architectures and applications,” Sci. Res. Essays, vol. 5, pp. 1033–1051, May 2010. [3] H.-M. Chen, Y.-K. Wang, Y.-F. Lin, and Z.-Z. Yang, “Single-layer crossed dipole antenna with circular polarization for handheld RFID reader,” Microw. Opt. Technol. Lett., vol. 50, no. 5, pp. 1172–1176, May 2011. [4] Y.-F. Lin, Y.-K. Wang, H.-M. Chen, and Z.-Z. Yang, “Circularly polarized crossed dipole antenna with phase delay lines for RFID handheld reader,” IEEE Trans. Antennas Propag., vol. 60, no. 3, pp. 1221–1227, Mar. 2012. [5] M. Huchard, C. Delaveaud, and S. Tedjini, “Miniature antenna for circularly polarized quasi isotropic coverage,” in Proc. 2nd EuCAP, Nov. 11–16, 2007, pp. 1–5. [6] W.-I. Son, H.-L. Lee, M.-Q. Lee, S.-B. Min, and J.-W. Yu, “Compact square quadrifilar spiral antenna with circular polarization for UHF mobile RFID reader,” in Proc. Asia–Pacific Microw. Conf., Dec. 7–10, 2010, pp. 2271–2274. [7] Nasimuddian, Z. Chen, and X. Qing, “Asymmetric-circular shaped slotted microstrip antennas for circular polarization and RFID applications,” IEEE Trans. Antennas Propag., vol. 58, no. 12, pp. 3821–3828, Dec. 2010. [8] P. V. Nikitin and K. V. S. Rao, “Compact Yagi antenna for handheld UHF RFID reader,” in Proc. IEEE Antennas Propag. Soc. Int. Symp., Jul. 11–17, 2010, pp. 1–4. [9] P. V. Nikitin and K. V. S. Rao, “Helical antenna for handheld UHF RFID reader,” in Proc. IEEE Int. Conf. RFID, Apr. 14–16, 2010, pp. 166–173. [10] J.-H. Bang, B. Enkhbayar, D.-H. Min, and B.-C. Ahn, “A compact GPS antenna for artillery projectile applications,” IEEE Antennas Wireless Propag. Lett., vol. 10, pp. 266–269, 2011.
