Active MMIC-Based Wireless Communication at 220 GHz - IEEE Xplore

IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY, VOL. 1, NO. 2, NOVEMBER 2011

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All Active MMIC-Based Wireless Communication at 220 GHz Ingmar Kallfass, Jochen Antes, Thomas Schneider, Fabian Kurz, Daniel Lopez-Diaz, Sebastian Diebold, Hermann Massler, Arnulf Leuther, and Axel Tessmann

Abstract—A wireless data link operating at a carrier frequency of 220 GHz is supporting a data rate of up to 25 Gbit/s in on-off-keyed PRBS as well as complex 256-QAM (quadrature amplitude modulation) transmission. The millimeter-wave transmit and receive frontends consist of active multi-functional millimeter-wave microwave integrated circuits (MMICs), realized in 50 nm mHEMT technology and packaged into split-block waveguide modules. The paper presents system considerations for wireless links in the 200–300-GHz range, discusses the design and performance of dedicated broadband transmit and receive MMICs, and presents link experiments. With an RF transmit power of 3.4–1.4 dBm in the IF frequency range from 0 to 20 GHz, a receiver conversion gain of better than 4.8 dB up to 270 GHz and an estimated noise figure of less than 7.5 dB at 1 PRBS with a data rate of up to 25 Gbit/s is 220 GHz, a 231 transmitted over 50 cm and received with an eye diagram quality factor 3. At 10 Gbit/s, an uncorrected bit-error rate (BER) of 1.6 10 9 is measured over a distance of 2 m. A 256-QAM signal with approx. 14 Mbit/s is received with an uncorrected BER of 9.1 10 4 . Index Terms—Millimeter-wave amplification, millimeter-wave FET integrated circuits, millimeter-wave frequency conversion, millimeter-wave monolithic integrated circuits (MMICs), wireless communication, 220 GHz.

I. INTRODUCTION

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XPLOITING the high millimeter-wave (mmW) frequency range from roughly 200 to 300 GHz for high data rate wireless communication gives rise to a number of unique and important benefits. The primary motivation being the availability of high absolute bandwidth, additional advantages include the small aperture and system size, the Manuscript received March 01, 2011; revised May 17, 2011; accepted May 25, 2011. Date of publication August 04, 2011; date of current version October 28, 2011. This work was supported by the German Federal Ministry of Research and Education (BMBF) in the frame of the MILLILINK project under Grant 01BP1023, and by the German Federal Ministry of Defence (BMVg) and the Bundeswehr Technical Center for Information Technology and Electronics (WTD81). I. Kallfass is with the Fraunhofer Institute for Applied Solid-State Physics (IAF), D-79108 Freiburg, Germany, and also with the Karlsruhe Institute of Technology, Karslruhe D-76131, Germany (e-mail: (e-mail: [email protected]); [email protected]). T. Schneider is with the Institut für Hochfrequenztechnik, Hochschule für Telekommunikation, D-04277 Leipzig, Germany. F. Kurz is with Siemens CT, D-80333 München, Germany. D. Lopez-Diaz, H. Massler, A. Leuther, and A. Tessmann are with the Fraunhofer Institute for Applied Solid-State Physics, D-79108 Freiburg, Germany. J. Antes and S. Diebold are with the Karlsruhe Institute of Technology, Institut für Hochfrequenztechnik und Elektronik, D-76131 Karlsruhe, Germany. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TTHZ.2011.2160021

strong degree of freedom to tailor the data links’ range and directivity to the needs of a specific application. Furthermore, the large frequency range offers a large potential to implement novel concepts for secure communication, as well as to build multi-functional frontends, serving both, as communication and sensing devices. Promising communication applications include the manifold short range scenarios of wireless personal area networks (WPANs), medium to long range directional links for telecommunication backhaul networks and last mile access, as well as intra-machine communication such as wireless sensor readout and board-to-board communication. Frequency regulation in this range is ongoing, but already the ISM band around 245 GHz provides 2 GHz of bandwidth. Wireless communication in the high mmW and sub-mmW frequency range, sometimes also daringly referred to as Terahertz communication, is today mainly addressed by commercially available passive electronic (Schottky diodes) [1], nonlinear optical [2]–[4] or InP-based electro-optical integrated technologies [5]. A comprehensive development of InP high electron mobility transistor (HEMT) based transmitters and receivers for successful directional links with up to 10 Gbit/s data rates at 120 GHz was reported in [6], [7]. By using a forward-error correction (FEC) the same signal could be transmitted over a distance of 5.8 km [8]. In the range 200–300 GHz, the theoretically exploitable bandwidth is enough to equal the data rates in today’s optical communication networks of 40 Gbit/s and even 100 Gbit/s. This paper presents data link experiments with up to 25 Gbit/s data rate, carried out at a carrier frequency of 220 GHz. The analog receive and transmit frontends are using high performance active millimeter-wave monolithic integrated circuits (MMICs), packaged into compact waveguide modules. In chapter II, we discuss the link budget and antenna gain requirements for 200–300 GHz links, considering realistic values of MMIC-based transmitter and receiver performance. Prospective transmit and receive frontend architectures are briefly addressed. After giving a detailed description in chapter III of the underlying high-speed transistor and MMIC technology used in this work, dedicated amplifying and frequency-translating MMICs for this application are covered in chapter IV. Finally, in chapter V, we present link experiments using, both, highly complex modulation formats and simple on-off-keying (OOK). While the link budget calculations focus on the advantageous center frequency of 240 GHz, the link experiments are carried out at 220 GHz center frequency, where the components for the generation and distribution of the local oscillator signal to the subharmonically driven receive and transmit modules were readily available to us.

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TABLE I PARAMETERS FOR THE LINK BUDGET CALCULATION

Fig. 1. Atmospheric attenuation in the millimeter- and submillimeter-wave frequency range under clear sky, foggy and rainy weather conditions.

II. SYSTEM CONSIDERATIONS FOR 200–300 GHZ WIRELESS COMMUNICATION A. Link Budget In the frequency range from 200 to 300 GHz, the standard earth atmosphere features a large transmission window without absorbing molecules (Fig. 1). The “am” model [9] provided by the Harvard-Smithsonian Center for Astrophysics predicts an atmospheric attenuation of 2.0 to 4.2 dB/km under clear sky conditions at sea level altitude. Recommendations of the International Telecommunication Union (ITU) allow to factor in the effects of fog and rain attenuation [10], [11]. According to these models, heavy fog with a liquid water density of 0.5 g/m , which would be a prohibitive condition to any optical signal, adds a moderate attenuation of 5.2 to 7.8 dB. A much more severe limitation to outdoor medium and high range communication applications is the combined absorption and refraction effect of rain on the mmW signal [12]. Heavy rain of 50 mm/h will attenuate the signal by 19–20 dB. It is interesting to note, however, that the attenuation due to rain imposes an almost equally strong impairment to any mmW and sub-mmW signal. Even at the important wireless frequency bands at 57–65 GHz and 71–87 GHz, rain drops will cause a 17 dB attenuation. Hence, in terms of the communication link budget, the move to high mmW frequencies does not entail an increasing impact of the rain attenuation. The following link budget calculation is based on the condiaccording to the tion for the minimum received power logarithmic formula

(1) is the noise power comwhere prising the thermal noise in the used bandwidth and the receiver noise figure NF, SNR is the required signal-to-noise ratio, is the transmit power, and are the transmit and reis the ceive antenna gain, respectively, is a safety margin due to e.g., antenna free-space path loss,

Fig. 2. Required antenna gain and diameter for a 80-GHz bandwidth communication link centered at 240 GHz.

misalignment or frontend loss, and is the atmospheric attenuation. The linear antenna gain is calculated based on

(2) where is the antenna efficiency, and is the diameter of a parabolic antenna. We assume a case of transmission of simple OOK data with a data rate of 40 Gbit/s in a total bandwidth of 80 GHz, centered about 240 GHz. The spectral efficiency, hence, is 0.5 bit/s/Hz. Furthermore, we assume a required energy per bit to noise of 10 dB. The goal of the power spectral density ratio link budget calculation is to estimate the required antenna gain for a desired link range. The transmit power of 7 dBm [13] and the receiver noise figure of 6 dB [14] are based on actual achieved MMIC performance. On top of the FSPL we add the atmospheric attenuation for conditions of clear sky and heavy rain, as well as a safety margin of 5 dB. Table I summarizes the assumed parameters to carry out the link budget calculation. Fig. 2 shows the resulting required Rx and Tx antenna gain as a function of transmission range. The corresponding diameter of a parabolic antenna is taking into account a realistic 50% antenna efficiency. For directional links, the impact of heavy rain is clearly visible above a range of 100 m. While an Rx and Tx antenna gain cm, would be of 44 dB, with a corresponding size of sufficient to transmit over 1 km under clear sky conditions, the

KALLFASS et al.: ALL ACTIVE MMIC-BASED WIRELESS COMMUNICATION AT 220 GHz

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antennas have to make up for the additional 20 dB loss in heavy rain with 10 dB more gain per antenna, resulting in a required antenna size of 28 cm. Distances of up to 1 m, however, as required in typical intramachine communication and WPAN scenarios, can be covered by only 13 dB antenna gain. Such values can be achieved by on-wafer integrated planar antennas in combination with dielectric lenses [15]. B. Frontend Architecture Trying to adopt a coherent communication architecture would involve the recovery of the carrier signal in the 200–300-GHz range. In order to detect the phase of the carrier signal by a phase detector, it has to be down-converted by either a frequency-divider chain or a mixer, or a combination of both. Although mixers are readily available in this frequency range, and frequency dividers operating up to 331 GHz have recently been demonstrated [16], [17], the adoption of complex modulation schemes such as binary phase-shift keying (BPSK) or quadrature phase-shift keying (QPSK) adds to the complexity of the system and limits the degrees of freedom in the phase-locked loop (PLL) design. With an assumed data rate of 40 Gbit/s, a carrier recovery for BPSK by means of a Costas-Loop would require a phase detector operating at 40 GHz. By using higher order MPSK modes, the required phase detector frequency could be reduced, but it imposes higher requirements on the allowable phase jitter of both the transmitted signal and the local oscillator on the receiving side. An incoherent approach with a significantly less complex receiver architecture can be implemented by either direct detection, or by using a down-conversion mixer producing a zero intermediate frequency (IF) or nonzero intermediate frequency (IF) , corresponding to a homodyne and heterodyne detection, respectively. Expressed in terms of the transmit and receive local oscillator frequencies, this means the case of and , respectively. With direct detection, no phase-modulated signals can be received, but, due to its low-complexity receiver architecture, it is the preferred approach if amplitude-shift keyed (ASK) signals only have to be transmitted, such as with standard on-off keying (OOK) modulation in optical fiber systems [6], [7]. The same limitation to amplitude-shift keying modulation applies in the case of incoherent homodyne detection, where one has to rectify and add two down-converted signals with a quadrature phase relation, in order to recover the transmitted signal regardless of the phase relation between the receiver and transmitter LO. This mandates the use of an IQ-mixer in the receiver and the implementation of ). a broadband magnitude recovery (i.e., producing Heterodyne, i.e., nonzero-IF, detection allows for the use of single-ended mixers and the adoption of complex phase and amplitude modulated signals [1]. However, the heterodyne approach imposes a strong limitation to the useable bandwidth as well as the need for selecting the wanted band of the down-converted signal by filtering. Prospective frontend architectures make use of a suitable sub-harmonic LO frequency generation. One possibility is to use the newly allocated frequency range from 55 to 65 GHz, where high quality tunable frequency sources have already been

Fig. 3. Evolution of cutoff frequencies and breakdown voltages when scaling the IAF mHEMT transistor from 100 to 50 and 35 nm gate length.

demonstrated (e.g., [18]–[20]) and will become commercially available in the wake of the expected consumer applications of 60 GHz wireless communication. The signal of such a source can then be multiplied by two, to the frequency range of 110–130 GHz, and be applied to either a sub-harmonic mixer or a combination of another frequency doubler and fundamental 200–300 GHz mixer. Such a frontend would be suitable e.g., for addressing the ISM band at 245 GHz. Today, mainly due to imaging applications in the atmospheric window around 94 GHz, the W-band (WR-10, 75–110 GHz) is well addressed by frequency sources, themselves often consisting of a combination of a highly stable low-frequency oscillator and frequency multiplier chain (e.g., commercially available source modules). Components in F (WR-8, 90–140 GHz) and D-Band (WR-6, 110–170 GHz) are much less frequent and considerably more expensive. In this work, we therefore choose the upper limit of the W-band for sub-harmonic LO signal generation to drive the transmit and receive frontend components. III. MMIC TECHNOLOGY A. Technology Candidates Active MMICs based on InP or GaAs metamorphic HEMT (mHEMT) technologies are today entering sub-mmW operating frequencies up to 550 GHz [21]–[23]. Also, the InP hetero-structure bipolar transistor (HBT) is a viable technology for the implementation of sub-mmW MMICs [24]. Their ability to integrate multiple circuit functionalities on a single chip makes MMICs highly attractive for compact, cost-efficient and easy-to-deploy frontend solutions. While Si-based high speed transistors have achieved competitive cutoff frequencies, their associated MMIC frequency of operation generally lags behind that of compound semiconductor based MMICs. They are, however, unrivalled in their capability of highly dense on-chip integration of RF and baseband functional blocks as well as their cost efficiency in high volume markets. Compound semiconductor based MMICs, on the other hand, excel in their RF performance with respect to noise figure, bandwidth and output power, and are able to address low volume markets.

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Fig. 4. Block diagram of the transmit and receive MMICs, consisting of a post- and pre-amplifier stage, respectively, and identical mixer and LO frequency doubler stages.

B. Metamorphic HEMT To enable submillimeter-wave frequencies of operation in amplifying MMICs, we are adopting a continuous scaling approach of mHEMTs and the accompanying MMIC process. Over the past 10 years, the transistors’ gate length has been successfully scaled from 100 down to presently 35 nm. The development of 20 nm devices is ongoing. Fig. 3 shows the (transit freevolution of the achieved cutoff frequencies (maximum frequency of oscillation), along quency) and with the inevitable reduction of the on- and off-state breakdown voltages. The MMICs presented in this paper are employing the 50-nm mHEMT variant, which is based on a composite In Ga As/In Ga As channel and a single-side -doping, grown by molecular beam epitaxy (MBE), which is accommodated on top of a metamorphic buffer with linear As compositional grading In Al Ga on 4-inch semi-insulating GaAs wafers. With the optimized MBE grown layer sequences, channel mobilities and channel cm V s and electron densities as high as cm have been measured [25]. The obtained of cutoff frequencies in a 2 10 m wide device are an 380 GHz, extrapolated from S-parameter measurement up to of more than 600 GHz, extrapolated 110 GHz, and an from the small-signal gain of submillimeter-wave amplifiers. The transistors show a maximum drain current density and transconductance of 1200 mA/mm and 1800 mS/mm, respectively. The on- and off-state breakdown voltages have been measured to 2.2 and 1.6 V, respectively. The fast transistor alone is not sufficient to up-scale the operational frequency of MMICs. The wavelength-related size reduction of all passive circuit elements mandates such measures as the reduction of ground-to-ground spacing in coplanar transmission lines, and the increase of capacitance density in MIM capacitors. Moreover, unwanted substrate propagation modes have to be suppressed for wave confinement to the transmission lines. The latter is achieved by the introduction of a backside process involving the thinning of the GaAs wafer to currently 50 m and the dry-etching of substrate through-vias to form frequent shortcuts between the ground planes on the wafer’s front- and backside. The vias with a diameter of 20 m can also be placed directly underneath an MIM capacitor; a measure which allows us to dramatically increase the compactness of submillimeter-wave MMICs. IV. MMIC-BASED TRANSMIT AND RECEIVE FRONTEND A. Circuit Topology The analog frontend used in the transmission experiments of this work consists of RF amplifying, frequency converting and

frequency multiplying stages. Both, the single-chip receiver and transmitter are driven by a subharmonic local oscillator signal, which is generated by a frequency multiplier chain. Although the presented transmission setups employ commercially available source modules for LO generation, the mHEMT technology allows to implement highly compact and broadband frequency multiplier MMICs, which are suitable for the wireless communication scenario of this work [26], [27]. Fig. 4 shows the functional blocks of the receive and transmit MMICs, dedicated to wideband communication in the frequency range from 200 to 300 GHz. A 4-stage small-signal amplifier is used as an RF low noise pre-amplifier (LNA) or post-amplifier stage in the receiver and transmitter MMIC, respectively. An identical resistive mixer stage converts from or to a wide IF frequency range. In order to use a subharmonic LO drive, the mixers in both MMICs are preceded by a frequency doubler stage. An LO frequency of 110 GHz, resulting in an RF carrier frequency of 220 GHz, is chosen since we use our in-house power amplifiers at this frequency in order to generate sufficient LO drive power to the receive and transmit MMICs. B. Circuit Design Fig. 5 shows the circuit schematic of the transmit MMIC (only one out of the four identical amplifier stages is included). The frequency doubler uses a conventional single-ended class-B transistor in common-source configuration. The optimum transistor for the targeted output frequency of 220 GHz is found by harmonic balance simulation to have a 4 15 m gate width. The reactive input matching network performs a conjugate complex power match to the fundamental frequency, while fully reflecting higher order harmonics. At the doubler output, the stub shorts the fundamental frecharacteristic open-ended quency component at the transistor drain, while the remaining matching network achieves power-matching to the second harV), the tranmonic. Biased under class-B conditions ( sistor generates a strong second-order harmonic component. Without any intermediate amplification stage, the output power from the doubler stage is matched to the gate terminal of a second 4 15 m transistor, which is operated as voltage-conV. Its gate bias is set close trolled resistor, i.e., at to threshold for minimum LO power requirements. In order to maximize the IF frequency range, an intentional low-pass filter, typically a shunt capacitor, is omitted at the mixer’s IF port. Thus, only the non-negligible pad capacitance and slightly reduced transmission line impedance of the tapered IF line act as bandwidth limiting factors. The amplifier was originally developed in 50-nm mHEMT technology as a broadband small-signal amplifier covering the WR-3 waveguide band from 220 to 325 GHz [22]. As a stand-alone amplifier, it achieves a measured small-signal


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Fig. 5. Simplified circuit schematic of the MMIC in the transmit case. In the receiver, the RF amplifier stages are mirrored. Only one of the four RF amplification stages is shown.

Fig. 7. Waveguide-packaged transmit or receive MMIC. Fig. 6. Chip photos of the single-chip transmit and receive MMICs, realized in 50 nm mHEMT technology. Chip size is 0.75 2 mm for each MMIC.

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gain of more than 16.5 dB between 220 and 320 GHz, when V, a second gate applying a drain voltage of V, a gate voltage of V voltage of mA. Each of the four and a total drain current of identical amplifier stages uses two 2 10 m transistors in cascode configuration. High frequency stability is achieved by an intentional transmission line between the common-source and the common-gate device, while low frequency stability is enforced by small series resistors in the gate and drain bias lines. The chip photographs of the Tx and Rx MMICs, realized in 50-nm mHEMT technology, are shown in Fig. 6. The transmission line environment is grounded coplanar waveguide (GCPW), with a small ground-to-ground spacing of 14 m for wave confinement and compact design at the frequency of operation. Both MMICs have identical interfaces to the outside world with respect to the RF/LO/IF ports and the location of the DC supply pads. C. Waveguide Packaging Due to their identical chip dimensions and interconnects, the transmit and receive MMICs can be mounted into identical splitblock waveguide packages using WR-10 (75–110 GHz) waveguide dimensions for the LO port, and a WR-3 (220–325 GHz) waveguide at the RF port (Fig. 7). The transitions from and to the waveguides are realized by 50 microstrip lines on 50 m thick

quartz substrates for height compatibility with the MMIC. More detailed information on the packaging technique can be found in [28]. Moreover, the frequency characteristics of the transmitter and receiver are perfectly matched to each other. The transmit output power, however, is limited to the saturated output power of the LNA stage, which has been optimized and dimensioned for maximum small-signal gain and lowest noise figure. The modules are measured by applying the LO frequency of 110 GHz from a W-band source module, post-amplified by an in-house 110 GHz power amplifier module with a maximum output power of 17 dBm. The receiver module is measured for its conversion gain by applying an RF signal in the frequency range from 220 to 270 GHz from an Oleson WR-3 T/R frequency extension module. In order to calculate the conversion gain from the measured IF power, the WR-3 source module’s output power is first carefully measured using an Erickson calorimeter. For the receiver to remain in linear regime, the RF power is attenuated 30 dBm by a waveguide attenuator. Fig. 8 shows the reto ceiver conversion gain in a sweep of the RF frequency and with 12 dBm of LO power applied at 110 GHz. The peak conversion gain of 1.5 dB is observed at 226 GHz, corresponding to an IF frequency of 6 GHz. Up to 270 GHz, the conversion gain is better than 4.8 dB. In a second measurement, we apply an RF signal below the 2xLO frequency from an Oleson WR-5 (140–220 GHz) T/R module, yielding the conversion characteristics of the lower sideband in the following transmission experiments. The performance in the sideband is limited, both,

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Fig. 8. Conversion gain of the receiver module versus RF frequency. Straight lines are measured, dashed lines are simulated results.

by the lower cutoff frequency of the LNA stage at approx. 208 GHz, and by the WR-3 waveguide cutoff. Also included in Fig. 8 is the simulated frequency response of the receiver module. The simulation is carried out on the MMIC level, but includes the additional loss of 1.3 dB incurred during waveguide packaging. For the doubler and resistive mixer, we employ custom nonlinear models. The cascode stages of the LNA can only be simulated with small-signal models. Due to the non-availability of large-signal models for cascode configurations, we can only simulate the receiver, but not the compression behaviour of the transmitter. The conversion gain level is well predicted in simulation, but the measured result is shifted up in frequency. Although, due to the lack of a suitable noise diode in this frequency range, the noise performance of the receiver has not yet been measured directly, we are able to confidently estimate a receiver noise figure between 6.5 and 7.5 dB at 220 GHz. This is based on two methods. 1) The noise measurement at 210 GHz of a similar LNA based on the 50-nm mHEMT technology shows a noise figure of 4.8 dB [14]. The Pospieszalski-type noise models included in the small-signal models predicted this level very well, and we therefore can extrapolate to a noise figure of 5.0 dB at 220 GHz. Accounting for the measured loss of 1.3 dB in the waveguide-to-quartz transition, and the mixer noise contribution based on the Friis formula for cascaded stages, the noise figure of the packaged receiver is expected to lie at 7.5 dB at 220 GHz. 2) The standalone LNA used in the transmitter and receiver has been used to investigate the sensitivity enhancement of a spectroscopic measurement of the water line at 321 GHz [29]. The improvement of the signal-to-noise ratio allowed to derive a measured noise figure for the packaged LNA of 7.9 dB. An extrapolation to 220 GHz results in an estimated receiver noise figure of 6.5 dB. The transmit module’s output power (Fig. 9) is evaluated by applying an IF signal from an Agilent E8257D PSG generator and measuring the total output power with an Erickson calorimeter. This measurement yields the sum total power of the frequency components at RF and 2 LO, the latter due to the finite LO-to-RF isolation of the up-converter. In order to obtain the RF transmit power only, we first measure the

Fig. 9. RF output power of the transmitter module versus IF.

power without any IF signal applied, and subtract this from the measurements including an IF signal, which are carried out at identical LO power and frequency. To confirm the validity of this approach, we verify that the measurement with a VDI subharmonic H-band mixer and a spectrum analyzer shows no component of the applied IF power. dependence of the We find that the RF output power varies from 3.4 to 1.4 dBm in the IF frequency range from 0 to 20 GHz, when an LO power of 10 dBm is applied at 109 GHz, and an IF power of 5 dBm. This measurement yields the sum of the power contributions from, both, the upper (USB) and the lower (LSB) RF sideband, the LSB being partially suppressed by two mechanisms: the lower waveguide cutoff, and the frequency response of the LNA stage in the transmitter. V. 220 GHZ WIRELESS LINK A. Link Setup The transmitter and receiver modules are used in various transmission experiments with different configurations. The general measurement setup is shown in Fig. 10. The transmit signal is up-converted to the RF range around 220 GHz by a 110 GHz LO signal. This sub-harmonic LO signal is generated in an HP W-band source module, split and fed to two in-house power amplifier modules, which deliver the required power level of 11 dBm to the receiver and transmitter modules’ LO ports. In the receive path, we include a W-band phase shifter to adjust the LO phase. This is required because the employed up- and down-conversion mixers are single-ended, and an incoherent LO at the receiver could result in IF signal cancellation depending on the phase relation of the receiver LO with respect to the transmit LO. At the RF ports of the receiver and transmitter we use commercial WR-3 circular horn antennas, each with an antenna gain of approx. 15 dB. For better beam alignment, we use high density polyethylene (HD-PE) plane-convex lenses to produce a collimated beam. The lenses are optimized for a center frequency of 325 GHz with ZEMAX1, a commercial ray-tracing program, normally used in the optical domain. The signal is fed to the lenses via the horn antenna in the focal point of the lense. The diameter of the lenses is approx. 70 mm. The down-converted signal is 1ZEMAX

is a registered trademark of Radiant ZEMAX LLC.


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Fig. 10. Coherent measurement setup for wireless transmission experiments at 220 GHz.

enough quality to fulfill the frequently adopted threshold condition of 1 10 for uncorrected BER required for reliable data transmission. C. OOK Transmission and BER Measurement up to 10 Gbit/s Over a 2 m Distance

Fig. 11. Setup for the wireless transmission of DVB-C signals.

post-amplified by suitable amplifiers to adjust the signal level to the required range of the respective analyzing equipment. B. DVB-C Transmission In a first experiment, standard DVB-C television signals are transmitted and their quality at the receiver is evaluated [30]. The two horn antennas are separated by 1 m and no lenses are used. The setup for this case is sketched in Fig. 11. The transmit signal is generated in a PC by a commercially available quadrature amplitude modulation (QAM) modulator card, which converts the stored HDTV transport stream file into a DVB-C signal by using a 16- up to 256-QAM modulation format. We choose a modulation carrier frequency of 643 MHz. The channel bandwidth is 8 MHz. Thus, accounting for a 0.15 safety roll-off we use a symbol rate of 6.9 MBd. Using 64 QAM or 256 QAM modulation, this results in a bit rate of 41.4 or 55.2 Mbit/s, respectively. Since we use only one out of the four possible channels, our data is transmitted with a bit rate of approx. 10.4 to 13.8 Mbit/s. The down-converted signal is post-amplified with dBm and fed without further mod50 dB to a power level of ification to either a commercially available HDTV televisor, or a Kathrein MSK 33/M Sat/TV/FM signal meter. Using the signal meter, we can evaluate the constellation diagrams of the received signals and derive bit error rates (BER). Fig. 12 shows the constellation diagrams for 16- up to 256 QAM transmission. The signal meter calculates BER prior to error correction. At 16- and 64-QAM, the BER is better than 2 10 , while for 128- and 256-QAM, the BER rises to 1.7 10 and 9.1 10 , respectively. Even the 256-QAM signal has a high

For eye-diagram measurements an Anritsu MP1758A bit pattern generator (BPG) is used to provide a pseudo random binary word length, that is used sequence (PRBS) with up to as an input signal (Fig. 13). The BPG can generate data rates up to 12.5 Gbit/s at four independent channels. The output amplitude and offset for the output signal can be adjusted separately for each channel. The output amplitude of the BPG is adjusted in a way that the corresponding input power at the transmitter module is 4 dBm. Due to the DC blocking characteristic of the bias-T in front of the transmit module, the baseband signal is converted to a polar bit stream, which results in a BPSK signal after up-conversion. The output signal of the receiver module is fed to one of the input channels of a HP 83480A Digital Communication Analyzer combined with a HP54752A two channel 50 GHz module. The analyzer is triggered by the BPG’s trigger output to the 1/32 clock. A second output from the BPG is directly fed to the analyzer for reference. The modules are aligned with a metallic mirror in such a way that the transmitted signal is in line with the receiver. The distance between the modules and the mirror is approx. 1 m, so that the total transmission distance is 2 meters. The input signal at the transmitter is fed to the IF port via a SHF bias-tee with a lower cut-off frequency of 50 kHz. The receiver’s downconverted IF signal is amplified by two cascaded broadband amplifiers of type SHF 804 TL to meet the minimum input requirements of the measurement equipment, i.e., the eye opening, which was approx. 25 mV up to 25 Gbit/s. This post-amplifier chain provides a gain of approximately 42 dB. Fig. 14 shows the eye diagram measurements of a PRBS with bits at 7.5 and 10 Gbit/s rates. The eye opening a length of is quantified by the quality factor, defined here as (3)

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Fig. 12. Constellation diagrams for 16- to 256-QAM transmission and corresponding uncorrected bit error rates.

Fig. 13. Setup for the wireless transmission of broadband OOK signals.

Fig. 14. Eye diagram measurements at 7.5 (left) and 10 Gbit/s (right) over a distance of 2 m yield quality factors of 3.55 and 3.58, respectively.

where and are the mean values and standard deviations of the measured logic high and low values, respectively. The measured -factors of the 7.5 and 10 Gbit/s data transmission are 3.55 and 3.58, respectively. To evaluate the bit errors introduced by the data link we use an Anritsu MP1764A bit error detector, which is triggered by the BGP’s trigger output. The BER is measured for data rates of 7.5 the bit error rates and 10 Gbit/s. For a pattern length of for a data rate of 7.5 Gbit/s and for are pattern the bit error rates a data rate of 10 Gbit/s. For a and , respectively. Each bit error rate are

measurement was performed for a time period of 30 minutes to get meaningful results. To assure that the measured bit error rates are correct, additional bit errors can be introduced at the BPG. If the error detector detects those manually introduced errors one can assume that the measurement is correct. D. OOK Transmission up to 25 Gbit/s Over a 50 cm Distance In order to determine the maximum data rate achievable with the analog Rx and Tx frontend components of this 220 GHz wireless link, the setup is changed (Fig. 15) to feed the output from a Sympuls BMG4x30G bit pattern generator to the IF port


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Fig. 15. Setup for the wireless transmission of OOK signals up to 25 Gbit/s.

Fig. 16. Eye diagram measurements at: (left) 12 and (right) 15 Gbits) over a distance of 50 cm yield good eye opening with quality factors of 6.61 and 4.18, respectively.

Fig. 17. Eye diagram measurements at: (left) 20 5 Gbit/s and (right) 25 Gbit/s over a distance of 50 cm show significantly reduced eye opening, albeit with quality factors >3.

of the transmit module. The BPG takes its clock signal from an Agilent E8257D signal generator, and outputs a complemenV signal with up to 30 Gbit/s into a 50 load. The tary transmitter and receiver are placed opposite of each other and the link distance is reduced to approximately 50 cm. To measure the input power at the receiver in this setup, we place an Erickson calorimeter after the receiving horn antenna and find a power level of 21 dBm, which is low enough for the receiver to remain in linear operation. At 12 and 15 Gbit/s, the measured eye diagrams of a 2 PRBS show a clear opening and associated quality factors of 6.61 and 4.18, respectively (Fig. 16). When increasing the bit rate to 20 and 25 Gbit/s and adjusting the transmitter’s LO phase for optimum eye opening, the obtained quality factors are 3.14 and 3.09 (Fig. 17). Although this degree of eye opening must be considered inadequate for reliable data transmission, the result nevertheless demonstrates the potential of wireless communication based on broadband MMIC technology in the frequency range between 200 and 300 GHz. Table II summarizes all results of the data transmission experiments. E. Discussion All of the presented transmission experiments are doublesideband transmissions, where the up-conversion leads to an

TABLE II SUMMARY OF THE 220 GHZ TRANSMISSION EXPERIMENTS

LSB and USB, both of which are down-converted in the receiver. In both, the receiver and transmitter, the LSB is partially suppressed by the frequency characteristic of the LNA and by the lower waveguide cutoff. The measured frequency characteristic of the receiver suggests that a relatively sharp suppression of the LSB occurs below 208 GHz, 12 GHz away from the carrier, which leaves a useable IF bandwidth of 12 GHz for double sideband experiments. A signal using more bandwidth will be distorted due to the unsymmetrical sideband contributions. In our experiments, the DVB-C transmission remains unaffected from this limitation, since it uses a bandwidth of only

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8 MHz around a carrier of 643 MHz. The successful transmission of 256-QAM demonstrates the high linearity performance of the presented link. The OOK transmission experiments present clear eye diagrams up to 15 Gbit/s, but start to degrade beyond this rate. The degradation is due to the signal requiring more bandwidth in both sidebands than is supported by the present link, which is limited in LSB bandwidth. Future generations of receive and transmit modules, designed for a carrier frequency of 240 GHz, are expected to support double sideband transmission of data rates up to 40 Gbit/s. VI. CONCLUSION Active single-chip receive and transmit MMICs have been employed to demonstrate wireless data transmission at 220 GHz. In various setups, both, highly complex data with 256-QAM modulation and PRBS with up to 25 Gbit/s data rate have been transmitted, and the received signal quality analyzed. To the best of our knowledge, this is the first time wireless data transmission beyond 10 Gbit/s has been demonstrated. The achieved results are proof of the high potential of active electronics for broadband wireless communication systems at high millimeter to submillimeter-wave frequencies. ACKNOWLEDGMENT The authors wish to express their gratitude to the partners of the Millilink project for valuable discussions. J. Rosenzweig and R. Makon at IAF for advice on BER measurements. Finally, the authors express their gratitude to the colleagues at IAF for their excellent contributions during epitaxial growth and wafer processing, and to M. Schlechtweg and O. Ambacher for their continuous support. REFERENCES [1] C. Jastrow, S. Priebe, B. Spitschan, J. Hartmann, M. Jacob, T. Kurner, T. Schrader, and T. Kleine-Ostmann, “Wireless digital data transmission at 300 GHz,” Electron. Lett., vol. 46, no. 9, pp. 661–663, 29, 2010. [2] A. Hirata, M. Harada, and T. Nagatsuma, “120-ghz wireless link using photonic techniques for generation, modulation, and emission of millimeter-wave signals,” J. Lightw. Technol., vol. 21, no. 10, pp. 2145–2153, Oct. 2003. [3] H.-J. Song, K. Ajito, A. Hirata, A. Wakatsuki, T. Furuta, N. Kukutsu, and T. Nagatsuma, “Multi-gigabit wireless data transmission at over 200-ghz,” in 34th Int. Conf. on Infrared, Millim. , and Terahertz Waves (IRMMW-THz 2009), 2009, pp. 1–2. [4] P.-L. Chen, S.-T. Chang, S.-T. Ji, S.-C. Lin, H.-H. Lin, H.-L. Tsay, P.-H. Huang, W.-C. Chiang, W.-C. Lin, S.-L. Lee, H.-W. Tsao, J.-P. Wu, and J. Wu, “Demonstration of 16 channels 10 Gb/s WDM free space transmission over 2.16 km,” in 2008 Dig. IEEE/LEOS Summer Topical Meetings, 2008, pp. 235–236. [5] T. Nagatsuma, A. Hirata, M. Harada, H. Ishii, K. Machida, T. Minotani, H. Ito, T. Kosugi, and T. Shibata, “Millimeter-wave photonic integrated circuit technologies for high-speed wireless communications applications,” in 2004 IEEE Int Solid-State Circuits Conf. Dig. Tech. Papers. (ISSCC. 2004), 2004, vol. 1, pp. 448–449. [6] T. Kosugi, M. Tokumitsu, T. Enoki, M. Muraguchi, A. Hirata, and T. Nagatsuma, “120-GHz Tx/Rx chipset for 10-Gbit/s wireless applications using 0.1 mu;m-gate InP HEMTs,” in 2004 IEEE Compound Semicond. Integr. Circuit Symp., 2004, pp. 171–174. [7] R. Yamaguchi, A. Hirata, T. Kosugi, H. Takahashi, N. Kukutsu, T. Nagatsuma, Y. Kado, H. Ikegawa, H. Nishikawa, and T. Nakayama, “10-gbit/s mmic wireless link exceeding 800 meters,” in 2008 IEEE Radio and Wireless Symp., 2008, pp. 695–698. [8] A. Hirata, T. Kosugi, H. Takahashi, J. Takeuchi, K. Murata, N. Kukutsu, Y. Kado, S. Okabe, T. Ikeda, F. Suginosita, K. Shogen, H. Nishikawa, A. Irino, T. Nakayama, and N. Sudo, “5.8-km 10-gbps data transmission over a 120-ghz-band wireless link,” in IEEE Int. Conf. on Wireless Inf. Technol. Syst. (ICWITS), Sept. 3, 2010, pp. 1–4.

[9] S. Paine, Harvard Univ., “AM atmospheric model,” 2011 [Online]. Available: http://www.cfa.harvard.edu/~spaine/am/ [10] Specific Attenuation Model for Rain for Use in Prediction Methods, ITU Recommend. ITU-R P.838-3, ITU, 2005, . [11] Attenuation Due to Clouds and Fog, ITU Recommend. ITU-R P.840-4, ITU, Oct. 2009. [12] J. Hansryd, Y. Li, J. Chen, and P. Ligander, “Long term path attenuation measurement of the 71–76 GHz band in a 70/80 GHz microwave link,” in 2010 Proc. 4th Eur. Conf. on Antennas and Propagation (EuCAP), 2010, pp. 1–4. [13] I. Kallfass, P. Pahl, H. Massler, A. Leuther, A. Tessmann, S. Koch, and T. Zwick, “A 200 GHz monolithic integrated power amplifier in metamorphic HEMT technology,” IEEE Microw. Wireless Compon. Lett., vol. 19, no. 6, pp. 410–412, Jun. 2009. [14] A. Tessmann, I. Kallfass, A. Leuther, H. Massler, M. Kuri, M. Riessle, M. Zink, R. Sommer, A. Wahlen, H. Essen, V. Hurm, M. Schlechtweg, and O. Ambacher, “Metamorphic HEMT MMICs and modules for use in a high-bandwidth 210 GHz radar,” IEEE J. Solid-State Circuits, vol. 43, no. 10, pp. 2194–2205, Oct. 2008. [15] Y. Yan, Y. Karandikar, S. E. Gunnarsson, B. M. Motlagh, S. Cherednichenko, I. Kallfass, A. Leuther, and H. Zirath, “Monolithically integrated 200-GHz double-slot-antenna and resistive-mixers in a GaAsmHEMT MMIC process,” IEEE Trans. Microw. Theory Techn., to be published. [16] M. Seo, M. Urteaga, A. Young, and M. Rodwell, “A 305–330 GHz 2:1 dynamic frequency divider using InP HBTs,” IEEE Microwave and Wireless Components Letters, IEEE, vol. 20, no. 8, pp. 468–470, 2010. [17] M. D’Amore, C. Monier, S. Lin, B. Oyama, D. Scott, E. Kaneshiro, A. Gutierrez-Aitken, and A. Oki, “A 0.25 m InP DHBT 200 GHz static frequency divider,” in IEEE Annu. Compound Semicond. Integr. Circuit Symp. (CISC 2009) , 2009, pp. 1–4. [18] T. Nakamura, T. Masuda, K. Washio, and H. Kondoh, “A 59 GHz pushpush VCO with 13.9 GHz tuning range using loop-ground transmission line for a full-band 60 GHz transceiver,” in IEEE Int. Solid-State Circuits Conf., 2009, pp. 496–497, 497a. [19] L. Li, P. Reynaert, and M. S. J. Steyaert, “A 60-GHz CMOS VCO using capacitance-splitting and gate-drain impedance-balancing techniques,” IEEE Trans. Microw. Theory Techn., vol. 59, no. 2, pp. 406–413, Feb. 2010. [20] A. Barghouthi, A. Krause, C. Carta, F. Ellinger, and C. Scheytt, “Design and Characterization of a V-Band Quadrature VCO Based on a Common-Collector Sige Colpitts VCO,” NAME OF PUBLICATION?, pp. 1–3, 2010. [21] W. R. Deal, K. Leong, X. B. Mei, S. Sarkozy, V. Radisic, J. Lee, P. H. Liu, W. Yoshida, J. Zhou, and M. Lange, “Scaling of InP HEMT cascode integrated circuits to THz frequencies,” in Proc. IEEE Compound Semicond. IC Symp., Monterey, CA, Oct. 2010, pp. 195–198. [22] A. Tessmann, A. Leuther, V. Hurm, H. Massler, M. Zink, M. Riessle, and R. Loesch, “High-gain submillimeter-wave mHEMT amplifier MMICs,” in Proc. IEEE MTT-S Int. Microwave Symp., Anaheim, CA, May 2010, pp. 33–36. [23] W. Deal, “Solid-state amplifiers for terahertz electronics,” in IEEE MTT-S Int. Microw. Symp. Dig., May 2010, pp. 1122–1125. [24] J. Hacker, M. Seo, A. C. Young, Z. Griffith, M. Urteaga, T. Reed, and M. Rodwell, “Thz MMICs based on InP HBT technology,” in IEEE MTT-S Int. Microw. Symp. Dig., May 2010, pp. 1126–1129. [25] A. Leuther, A. Tessmann, I. Kallfass, H. Massler, R. Loesch, M. Schlechtweg, M. Mikulla, and O. Ambacher, “Metamorphic HEMT technology for submillimter-wave MMIC applications,” in Proc. 22nd Int. Conf. on Indium Phosphide and Related Materials, Piscataway, NJ, May 2010, pp. 425–429. [26] I. Kallfass, A. Tessmann, H. Massler, P. Pahl, and A. Leuther, “An allactive MMIC-based chip set for a wideband 260–304 GHz receiver,” in Proc. 5th Eur. Microw. Integr. Circuit Conf., Paris, France, Sep. 2010. [27] I. Kallfass, H. Massler, A. Tessmann, A. Leuther, M. Schlechtweg, and G. Weimann, “A broadband frequency sixtupler MIMIC for the W-band with >7 dBm output power and >6 dB conversion gain,” in Proc. IEEE MTT-S Int. Microwave Symp., Honolulu, HI, June 2007, pp. 2169–2172. [28] A. Tessmann, A. Leuther, V. Hurm, H. Massler, M. Zink, M. Kuri, M. Riessle, R. Lösch, M. Schlechtweg, and O. Ambacher, “A 300 GHz mHEMT amplifier module,” in Proc. Indium Phosphide and Related Materials Conf., Newport Beach, CA, May 2009, pp. 196–199. [29] I. Kallfass, A. Tessmann, A. Huelsmann, A. Leuther, J. S. Cetnar, J. M. Noble, and D. T. Petkie, “MMIC technology for spectroscopy applications,” in SPIE Symp. on Opt./Photon. in Security and Defence, Millimetre Wave and Terahertz Sensors and Technology, Toulouse, France, Sep. 2010.

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[30] I. Kallfass, A. Tessmann, J. Antes, D. Lopez-Diaz, M. Kuri, H. Massler, and A. Leuther, “Wireless DVB-C transmission at 220 GHz using active single-chip receive and transmit MMICs,” in IEEE Int. Microw. Symp., Baltimore, MD, 2011.

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Daniel Lopez-Diaz was born in 1982. He received the Dipl.-Ing. degree from the University of Karlsruhe in 2008., and is currently working toward the Ph.D. degree from the Fraunhofer Institute for Applied Solid State Physics, Freiburg, Germany. His main research interests are the design of frequency conversion circuits above 200 GHz and millimeter-wave imaging.

Ingmar Kallfass received the Dipl.-Ing. degree in electrical engineering from University of Stuttgart in 2000, and the Dr.-Ing. degree from University of Ulm in 2005. In 2001, he worked as a visiting researcher at the National University of Ireland, Dublin. In 2002, he joined the department of Electron Devices and Circuits of University of Ulm as a teaching and research assistant. In 2005, he joined the Fraunhofer Institute for Applied Solid State Physics with a focus on nonlinear millimeter-wave integrated circuit design. Since June 2009, he is a professor at the Karlsruhe Institute of Technology. Karlsruhe, Germany, in the field of high speed integrated circuits, in a shared professorship within the framework of the German Excellence Initiative.

Sebastian Diebold was born in Karlsruhe, Germany, in 1982. He studied electrical engineering at the Karlsruhe Institute of Technology (KIT), Germany, and graduated with the Dipl.-Ing. degree in 2009. While working on his diploma degree at the Fraunhofer Institute for Applied Solid State Physics, he investigated mHEMT behavior at high millimeter-wave (mmW) frequencies by comprehensive measurements. He continues these studies as research associate and teaching assistant at the KIT, where he is involved in mmW circuit design and characterization.

Jochen Antes was born in 1983. He received the Dipl.-Ing. degree in electronic engineering from the Karlsruhe Institute of Technology, Karlsruhe, Germany, in 2010. He joined the Department of High Frequency Techniques and Electronics, Karlsruhe Institute of Technology Karlsruhe, Germany, as a research and teaching assistant, where he is currently pursuing a doctorate degree. His main interest is on high speed mixed-signal circuit design and millimeter-wave wireless links.

Thomas Schneider received the diploma degree in electrical engineering from the Humboldt Universität zu Berlin, Germany, in 1995, and the Ph.D. degree in physics from the Brandenburgische Technische Universität Cottbus, Germany in 2000. Since 2000, he has been with the HfT in Leipzig, Germany. Since 2006, he has been the head of the Institut für Hochfrequenztechnik, at the HfT. In his diploma thesis he investigated the nonlinear properties of optical fibers and during his doctoral studies he was engaged in the investigation of linear and nonlinear phenomena induced by an ultrafast refractive index grating. His current research interests include nonlinear optical effects in telecommunications, light storage, high resolution spectroscopy and the generation of millimetre and Terahertz waves.

Fabian Kurz was born in 1983 in Münster, Germany. He studied electrical engineering at the Dresden University of Technology (TUD), and received the Dipl.Ing. degree in 2010. In 2008, he joined Siemens Corporate Technology (SCT), Munich, Germany, as an intern and working student. He is currently doing postgraduate research in the field of wireless high-speed communication systems.

Hermann Massler was born in Radolfzell, Germany, in 1965. He studied electrical engineering at the Technical University Karlsruhe where he received the Diploma degree in 1993. While working on his diploma degree at the Kernforschungszentrum Karlsruhe (KfK) he did quasi-optical measurements at 140 GHz. He continued these studies as Research Assistant in the KfK for one more year. Since 1994 he has been with the Fraunhofer Institute for Applied Solid State Physics working on transistor and IC characterization up to 500 GHz.

Arnulf Leuther received the Dipl. Phys. degree and the Ph.D. degree in physics from the Technical University of Aachen, A He has been with the Fraunhofer Institute for Applied Solid State Physics since 1996, working primarily in the development of HEMT technologies for sensor and communication systems up to 600 GHz.

Axel Tessmann received the Dipl.-Ing. and Dr.-Ing. degrees in electrical engineering from the University of Karlsruhe, Germany, in 1997 and 2006, respectively. In 1997, he joined the Fraunhofer Institute for Applied Solid State Physics, where he is involved in the development of monolithic integrated circuits and subsystems for high-resolution imaging systems. His main research areas are the design and packaging of millimeter-wave ICs using high electron mobility transistors on GaAs, GaN and InP as well as circuit simulation and linear and nonlinear device modeling. He is currently head of the millimeter-wave packaging and subsystem group at the Fraunhofer IAF.