and Three-Phase Control for Grid Connected Electric Vehicles

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IEEE TRANSACTIONS ON SMART GRID, VOL. 4, NO. 4, DECEMBER 2013

A Unified Single- and Three-Phase Control for Grid Connected Electric Vehicles Arnaldo Arancibia, Student Member, IEEE, Kai Strunz, and Fernando Mancilla-David, Member, IEEE

Abstract—In order to reduce cost, size, weight, and volume of on board chargers in electric vehicles, it has been proposed to pursue integration for the dual usage of converters for both charging and propulsion. The solution here goes further in that it also supports the control of slow single-phase and fast three-phase charging through one and the same power electronic converter. A unified control methodology for the on-board chargers in electric vehicles is proposed. The control is distinguished in that it can perform four-quadrant operation while connected in single-phase or three-phase mode. The operation and network synchronization are automatically adjusted based on given terminal voltage and current measurements without the need for supplementary status signals. An analysis illustrates how the various measurements are processed for obtaining single- and three-phase flexibility while retaining compatibility with well-established methods of converter current mode control. The proposed control is validated via detailed simulation and demonstrated in a smart grid laboratory. The hardware implementation in the laboratory substantiates the claims of flexible single- and three-phase control. Index Terms—Electric vehicles, power electronic control, single-phase charging, three-phase charging, vehicle to grid, voltage sourced inverter.

I. INTRODUCTION

A

S THE penetration of plug-in hybrid electric (PHEVs) and plug-in electric vehicles (EVs) continues to increase, uncoordinated and uncontrolled charging of these vehicles could significantly impact the distribution grid [1], [2]. Within the Smart Grid Initiative, the need to accommodate for the impact of a large number of EVs performing charging and vehicle-to-grid (V2G) calls for intelligent control methodologies [3]–[5]. Also, both the size of current EV batteries as well as the driving needs of EV owners call for faster charging times. However, the aforementioned features have to be supported by the deployment of adequate infrastructure in the utility grid [6], including power converters, smart meters, and smart load management strategies [7]–[10]. Several EV modeling and control strategies have recently been reported in the literature [11]–[16], including also reactive power compensation [17]–[19]. But the lack of standards

Manuscript received May 18, 2012; revised September 16, 2012, December 26, 2012, and March 29, 2013; accepted May 04, 2013. Date of publication September 12, 2013; date of current version November 25, 2013. Paper no. TSG-00282-2012. A. Arancibia and K. Strunz are with the School of Electrical Engineering and Computer Science, Technical University of Berlin, 10623 Berlin, Germany (e-mail: ). F. Mancilla-David is with the University of Colorado Denver, Denver, CO 80204 USA. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TSG.2013.2271096

Fig. 1. Generalized scheme of a grid-connected electric vehicle system: (a) conventional system with three set of converters for each functionality, (b) integrated converter for multiple functionalities.

in the automotive industry has led to different circuit topologies and control structures. Nowadays, according to the SAE J1772 standard [20], the IEC62196-2 standard [21], and CHAdeMO [22], grid-connected EV charging modes include single-, threephase and dc charging. SAE J1772 defines Level 1, Level 2, and Level 3 charging, for both dc and ac charging modes. While dc charging mode considers off-board EV supply equipment (EVSE), ac charging mode is performed mainly with on-board chargers [20]. Fig. 1(a) shows a scheme of an EV’s power stage, where there are three distinct functionalities: 1) propulsion, 2) Level 3 fast charging station which directly feeds the battery with dc power, and 3) single-phase Level 1 and Level 2 charging performed by an on-board charger and an EVSE control box. The figure suggests there exists an extensive installation of converters which results in a non-optimal usage of components, also leading to additional weight and cost. For example, the Nissan Leaf includes an on-board charger, a plug for ac Level 2 and dc Level 3, and the EVSE control box for conventional domestic plugs. However, for Level 2 charging an EVSE system must be installed at home. According to [23], the cost of a Nissan Leaf’s EVSE system is around $2400 USD.

1949-3053 © 2013 IEEE

ARANCIBIA et al.: A UNIFIED SINGLE- AND THREE-PHASE CONTROL FOR GRID CONNECTED ELECTRIC VEHICLES

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Fig. 2. Circuit schematic of the proposed unified grid-connected electric vehicle system.

In order to reduce cost, size, weight, and volume of on-board chargers, it has been proposed to integrate the charging converters with the converters of the propulsion system [24]–[27]. For example, the Nissan Leaf propulsion inverter feeds a 80 kW three-phase motor. Such an inverter could support fast charging thus allowing a single converter to be utilized for both propulsion and fast charging. However, a key point is that the on-board single-phase charger still is an inevitable component of EVs, because it is needed for connecting to domestic single-phase 110V/220V, 15A/30A outlets [24]. Motivated by the above discussion, this paper proposes a novel unified control scheme for EV charging able to perform four-quadrant operation in both single- and three-phase systems, and applicable to a single power converter architecture as suggested in Fig. 1(b). The main contributions of the control are: 1) a novel unified phase locked loop (PLL) which can autonomously track single- and three-phase signals, 2) a current signal preprocessing stage which allows for the use of the same control strategy indifferent to the single- or three-phase nature of the network that is connected to, and 3) a unified pulse width modulation (PWM) strategy for a three-leg two-level voltage source inverter (VSI) which can operate as a single-phase H-bridge or a three-phase VSI. The paper is organized as follows. Section II provides the EV circuital model, followed by the description of the unified control strategy in Section III. Detailed Matlab/Simulink computer simulations in Section IV and experimental prototyping in Section V are presented in order to validate the control scheme. Finally, the conclusions of Section VI close the paper. II. ELECTRIC VEHICLE MODEL The architecture of the EV’s power conversion system considered in this paper is shown in Fig. 2. It includes a battery bank connected to a dc bus via a bi-directional dc-dc converter and a VSI which serves as the interface to the ac system. The battery is modeled through a Thévenin equivalent including its transient behavior. The battery parameters are adopted from [28]. The bidirectional dc-dc converter regulates the battery charge and discharge processes. The ac grid is modeled by an ideal voltage source behind an impedance and may correspond to a single- or three-phase network. The EV

may connect to the ac grid at a domestic outlet or at a charging station, leading to single- or three-phase charging operating modes, respectively. It is assumed that in either case there will be a metering device which will provide the grid voltage signal for the EV control. Fig. 2 also illustrates the VSI connects to the grid through a standard LCL filter to comply with power quality requirements [29]. Design guidelines for properly sizing this filter are readily available in the literature [30]. Details of the unified control, which is the focus of this paper, are provided in the next section. III. PROPOSED UNIFIED CONTROL This section provides details on the unified control proposed in this paper. The following subsections explain the overall control strategy as well as the key components which make this control indifferent to the single- or three-phase nature of the ac grid network the EV is connected to. A. Flexible Control System Structure A conceptual block diagram of the proposed control is presented in Fig. 3. As customary in inverter control, it is assumed that dc and ac currents and voltages are available for measurement. According to active and reactive power references, the inverter allows for power exchange with the ac grid in either direction. Hence, the controller proposed in this paper is able to regulate the active and reactive power exchange in the following operating modes: i) single-phase ac charging, ii) three-phase ac charging, iii) single-phase V2G, and iv) three-phase V2G. In all four modes, the system can provide reactive power compensation as well. A key observation for the operation of the single- and threephase flexible controller is that the controller is always fed with a set of three voltages. When connected to a three-phase grid, the set of three voltages and currents corresponds to a standard set of three-phase quantities (1)

(2)

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Fig. 5. Proposed unified PLL block diagram.

Fig. 3. Overall block diagram of the proposed flexible unified control.

Fig. 6. Block diagram of a second order filter.

C. Unified PLL Fig. 4. Block diagram illustrating the battery charger control.

However, when connected to a single-phase grid, the set of three voltages and current takes the form of

(3)

(4) are the magnitude, angular freIn (1) and (3) , and quency and phase of the control voltage signals, respectively. In (2) and (4) and are the magnitude and phase of the control current signals, respectively. Fig. 3 shows that the overall control features two main blocks: the battery charger control and the proposed VSI unified control. The latter contains three main stages: i) a unified PLL to track the magnitude and phase angle of the ac voltage at the point of common coupling (PCC), ii) a current control loop whose function is the synthesis of modulating functions according to the active and reactive power references, and iii) a unified PWM scheme which generates the VSI switch pulses according to the modulating functions. These three stages are conceived to provide the basis for the flexibility of the VSI control by supporting both single- and three-phase operation without the need for supplementary status signals or circuitry. The battery charger control works independently of the VSI control and are interfaced through keeping a constant dc-link voltage level. B. Battery Charger Control As illustrated in Fig. 4, the battery charger control consists of a dc-link voltage loop cascaded with a battery current loop. The control keeps the dc-link voltage between the VSI and the charger at and gives the duty ratio .

A PLL block is utilized to track the grid voltage magnitude and phase [31]. In both single- and three-phase networks, orthogonal sinusoidal signals are required to generate the phase comparator [32]. In three-phase systems, instantaneous orthogonal signals arise naturally from Clarke’s transformation [33]. For single-phase networks orthogonal signals do not occur naturally and are artificially created using filters leading to unwanted delays [34]. Thus, tracking speed and frequency transient response are reduced when compared to three-phase PLLs. This paper proposes a PLL able to track single- and three-phase voltage signals with adequate tracking speed in both cases. To achieve this, the proposed PLL features five blocks as depicted in Fig. 5. The “Filter” block synthesizes three pairs of orthogonal signals which are fed into a novel “ Matrix” block in order to generate a pair of orthogonal signals in the coordinates. These signals are transformed into the reference frame via the rotation matrix , where the quadrature component carries the information about the phase difference and thus acts as a phase comparator. Finally, the block controls the frequency, which is fed into a standard voltage controlled oscillator. 1) Filter: As suggested in Fig. 6, the selected filter is based on a generalized second order integrator [35], [36]. The filter generates two output signals from each sinusoidal input: one in-phase and one in-quadrature . It is to be noted that as defined in (1) and (3) comprises three signals, and hence the filter block outputs a total of six filtered signals. Additionally, it is worth mentioning that the filter block could alternatively generate as the derivative of the original input signal. This approach features a faster performance during transients as reported in [37], but unfortunately would have a large noise level. As a result, the integral component is chosen in this paper. 2) Matrix: The Matrix processes the input voltage signals and the signals coming from the “Filter” block to generate a unique pair of orthogonal signals as defined in (5) and (6). The novelty of the Matrix stems from its ability to “delete”


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Fig. 7. Block diagram for the inverter’s current control.

the unneeded filtered signal in the three-phase case. The processing is autonomous and is not dependent upon external status signals.

where is a proportional gain, is the input signal magnitude, is the nominal or expected frequency of the system, and is the transfer function with a non-zero value at . Further details on and the selection of and may be found in [38]. D. VSI Current Control

(5)

(6) The structure of (6) suggests an interpretation in terms of two partitions for the resulting voltages. The partition to the left of the dashed line in (6) may be interpreted as a “conventional” transformation. The partition to the right corresponds to an “additional” term which in the single-phase case provides the means to generate an orthogonal signal, while in the three-phase case it cancels out due to the inherent symmetries encountered in such systems. 3) Matrix: The Matrix block rotates the coordinates into the plane. This transformation, defined in (7) and in (8), makes the voltage signals and behave as dc quantities. (7)

The function of the current control loop is the generation of modulating signals to feed the pulse width modulator according to the active and reactive power references. As in the PLL block, the current control also receives single- or three-phase signals. The processing of input current signals and the modulation of the switches must be performed accordingly to account for the number of phases available at the ac network. As suggested in Fig. 7, this paper proposes a novel pre-processing stage to autonomously generate a pair of orthogonal current signals. Details of this implementation are described in the following subsections. 1) Calculation of Current References: As explained above, the VSI is current-controlled according to the active and reactive power references. The active power reference takes a negative or positive value under charging or discharging mode, respectively. In either mode the reactive power reference can take a positive or negative value depending on the needs of the power system. Therefore, the control allows for four-quadrant operation—a quite desirable feature for the smart grid [18], [39]. The translation of power into current references is performed considering the definition of active and reactive power in the reference frame [38],

(8) (10)

: The input signal contains the 4) PLL Controller information about and [32]. The PLL controller brings this signal to zero resulting in . The output of is the frequency of the input voltage signal. features a polynomial structure which can be a simple first order PI controller [33] or a higher degree polynomial as suggested in [37]. Herein the structure of is selected as

and provides a meaIn steady state the PLL forces surement for , which allows to compute the current references from (10),

(9)

(11)

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2) Proposed Preprocessing Stage: This control stage preprocesses the input currents and autonomously generates the currents which feed a conventional current control block. Fig. 7 suggests the preprocessing is performed in two steps. A signal generator synthesizes a set of three artificial currents which are then combined with the set of the three measured currents through the “Matrix .” This yields the currents independent of number of phases in the input currents. The artificial currents signal are generated through (12) (12) and the

currents are obtained via

subsection, a conventional current control scheme in the reference frame is implemented in order to generate the inverter modulation signals [38]. The errors from the measured currents and their references are added to feed-forward loops in order to decouple the dynamics of the components so that a better dynamic performance can be obtained [31], [38]. Procedures to tune the PI controllers are readily available in the literature [40]. E. PWM The PWM of the unified VSI control of Fig. 3 modulates the pulses of the VSI switches according to the number of signals from the current control. In Fig. 7 it can be seen that, independently of the number of phases present in the ac grid, the current control always outputs a set of three signals. In order to quantify the control action in either case, a conventional modulation for the two-level three-phase VSI [41] depicted in Fig. 1 is considered:

(13) (17) where in the

with

and readily follow from the modulation indices reference frame,

(14) Basic algebra yields the following expressions for the currents:

(15) In the case of a three-phase grid, the term cancels out—this can be easily verified given the definition of and considering (1) and (2)—and the conventional currents are obtained. In the case of a single-phase grid, the voltages and currents are defined by (3) and (4) leading to the following expressions,

(18) On the other hand, when two of the three legs of the VSI—e. g. phases a and b—connect to a single-phase ac network, the three-phase VSI takes the form of a single-phase H-bridge inverter. The unified PWM modulator is exactly the same as in the three-phase case, hence the VSI output voltage takes the form of a line-to-line voltage: (19) In the single phase case, a conventional sine-triangle modulation strategy for the VSI ought to be as [41]: (20) where

and

are defined as,

(16) In (16), and feature the same fundamental component and differ only by the noise— contains switching noise while is a purely sinusoidal signal. The resulting noise will add to . Hence, becomes a phase-shifted signal with respect to with a similar noise level to . As in the case of the PLL, the matrix may also be interpreted through two partitions defined by the dashed line in (14). The block to the left of the dashed line in (14) represents a conventional transformation while the block to the right side corresponds to an additional block which provides the means to generate the proper components in the single-phase case while it cancels out under three-phase operation. 3) Interface With Conventional Current Control: Once the currents have been generated according to the previous

(21) Accordingly, when connected to a single-phase ac grid, the current control is forced by the active and reactive power references and the PLL phase to compute the modulation indices in such a way that (19) and (20) become equivalent, that is, (22)

IV. SIMULATION AND PERFORMANCE ANALYSIS The performance of the proposed control scheme is assessed through three case studies. The EV is considered to be connected to either a single- or three-phase grid. A start-up case study is


TABLE I PARAMETERS FOR SINGLE- AND THREE-PHASE SYSTEMS

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voltages and the PLL process is finalized as suggested in Fig. 5. 2) Current Control Preprocessing: The unifying action of the current preprocessing stage from Fig. 7 is illustrated by the set of side by side single- and three-phase waveforms of Fig. 9. From top to bottom the plot depicts signals for the input currents [Fig. 9(a) and (b)], artificial currents [Fig. 9(c) and (d)], the conventional currents [Fig. 9(e) and (f)], the additional currents [Fig. 9(g) and (h)], and the unified currents [Fig. 9(i) and (j)]. As in the case of the PLL, the goal is the synthesis of identical currents independent of the number of phases. It may be readily observed from Fig. 9(g) and (h) that the additional signals provide the means to achieve the same unified currents, as depicted in Fig. 9(i) and (j). It is noteworthy that in the single-phase case the additional signals do not carry any harmonics. However, the noise on the conventional signal contributes to have a similar harmonic content on the unified signals as compared to their three-phase counterparts. Furthermore, the additional signals as obtained through the matrix of (15) are obtained through an algebraic operation leading to comparable control speeds under both single- and three-phase operating modes. B. Case of Unified PWM Performance

used to show the unifying action of the PLL and current control blocks. A second case study is performed in order to validate the unified PWM modulator. Finally, a third case study is presented to illustrate the V2G capability under both singleand three-phase operating scenarios. For this purpose, the circuit schematic of Fig. 2 along with the control scheme described in Section III were simulated using the Matlab-Simulink’s SimPowerSystems blockset. A complete list of the various parameters used in the simulation studies is presented in Table I. A. Case of Start-Up The start-up case study simulates the transient when plugging the EV into the ac grid. The synchronization process takes place, and the charging power is settled at the pre-established power references. In this example, the charging power is set at 3.5 kW and 10.5 kW for the single- and three-phase charging scenario, respectively. In both cases, the charging process is set at unity power factor. As the single-phase power reference is exactly 1/3 of the three-phase reference, it is expected to observe identical voltages and currents. 1) Unified PLL: Side by side single- and three-phase simulated waveforms are presented in Fig. 8. From top to bottom the figure depicts the input voltage signals [Fig. 8(a) and (b)], in-phase filtered voltages [Fig. 8(c) and (d)], in-quadrature filtered voltages [Fig. 8(e) and (f)], the conventional voltage signals [Fig. 8(g) and (h)], the additional voltage signals [Fig. 8(i) and (j)], and the unified voltage signals [Fig. 8(k) and (l)]. The unifying effect of the matrix can be readily appreciated from the figure. After the filtered signals are synthesized, the matrix generates proper additional signals that yield identical unified voltages. As seen from the figure, in the three-phase case the additional signals cancel out, as predicted by the theory of Section III. Once the voltages are synthesized, the rotational matrix generates the

This section illustrates the performance of the PWM modulator under a single-phase operating scenario. As explained in Section III-E, the hardware features a three-phase VSI, and for single-phase operation the modulation index and phase angle of the modulator must mimic the performance of a single-phase H-bridge according to (22). For the purpose of confirmation, a separate simulation of a single-phase H-bridge with conventional sine-triangle modulation [41] was implemented and compared against the architecture discussed in this paper. The results for the modulation index and phase in the conventional single-phase case are presented in Fig. 10(a) and (c), respectively. For the proposed unified PWM, Fig. 10(b) and (d) show the modulation index and the phase, respectively. It may be seen from the figure that the prediction made by (22) is accurate, on an average basis, for and . A different noise level is observed as the actual switching pattern is different in both modulation strategies. C. Case of V2G Operation Different operating modes for both single- and three-phase operating scenarios were simulated over the course of one second. This time window was selected for illustrative purposes to show the functions of the unified control. Fig. 11(a) shows the results for single-phase active power, while Fig. 11(b) shows the results for three-phase active power. In the same way, Fig. 11(c) shows the results for single-phase reactive power, while Fig. 11(d) shows the results for three-phase reactive power. At , the system is charging and the reactive power is set to zero. At , the reactive power is set to a negative value. At , discharging is requested and the system starts to inject power to the grid. Finally, at , the reactive power reference is changed to a positive value. The comparison between Fig. 11(a) and 11(b) shows that the three-phase active and reactive power are exactly three times

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Fig. 8. PLL voltage signals in simulation (a) single-phase input voltages, (b) three-phase input voltages, (c) single-phase in-phase filtered voltages, (d) three-phase in-phase filtered voltages, (e) single-phase in-quadrature filtered voltages, (f) three-phase in-quadrature filtered voltages, (g) single-phase conventional voltages, (i) single-phase additional voltages, (j) three-phase additional voltages, (k) single-phase voltages, (h) three-phase conventional voltages. unified voltages, and (l) three-phase unified

the single-phase active and reactive power, as expected. It may be observed that the controller can operate at an arbitrary power factor with seamless transition between different operation modes. V. EXPERIMENTAL VERIFICATION The main components of the architecture illustrated in Fig. 2 along with the unified control described in Section III were prototyped in the E-MERGE (Emerging, Mobile, and Electric Resources in Grids of Energy) Smart Grid Laboratory of the Technical University of Berlin [42]. The purpose of the experiment is to demonstrate in-hardware the ability of the proposed unified control to perform four-quadrant operation while connected to a single- and three-phase network. A. Hardware Description The overall diagram of the hardware setup is illustrated in Fig. 12 where two configurations were implemented: the

prototype connected to a 220 V single-phase grid as shown in Fig. 12(a), and the prototype connected to a 400 V three-phase grid as shown in Fig. 12(b). In both cases the ac frequency is 50 Hz. As the figures suggest, in single-phase operation, the first and second legs of the inverter are connected to the two terminals of the ac grid while the third leg is left unconnected. For three-phase operation, the inverter’s legs are connected to phases a, b, and c. The battery was prototyped through an electric storage of 0.9 Ah operated at 50% state of charge (SOC). The dc-dc converter is a bidirectional converter, and the inverter utilized corresponds to a 5.5 kVA two-level three-phase converter equipped with a built-in LCL filter. A dedicated plug was constructed for connecting the same inverter to either the a 220 V single- or 400 V the three-phase grid. The control system of Fig. 3 was implemented in a commercial real time target (RTT) DSP unit which is able to interpret control systems from Simulink. The RTT unit drives the dc-dc converter and inverter’s IGBTs through an EtherCAT Fieldbus. The switching


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Fig. 9. Preprocessor current signals in simulation (a) single-phase input currents, (b) three-phase input currents, (c) single-phase artificial currents, (d) three-phase currents, (f) three-phase conventional currents, (g) single-phase additional currents, (h) artificial currents, (e) single-phase conventional currents, (i) single-phase unified currents, and (j) three-phase unified currents. three-phase additional

Fig. 10. Unified PWM in simulation (a) modulation index for conventional single-phase case, (b) modulation index for proposed unified PWM, (c) phase for conventional single-phase case, (d) phase for proposed unified PWM.

frequencies for the dc-dc converter and inverter were selected at 16 and 8 kHz, respectively. B. Experiment Description The experiment consists of allocating various values of active and reactive power references to illustrate four-quadrant opera-

Fig. 11. V2G operation in simulation (a) single-phase active power, (b) threephase active power, (c) single-phase reactive power, and (d) three-phase reactive power.

tion. The VSI is first connected to the single-phase network, followed by the connection to the three-phase network. A window of 80 s is considered where the power commands are sequentially changed as described in Table II. A larger time window, compared to the simulations of Section IV, was selected in order

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Fig. 13. Four-quadrant operation in-hardware experimental verification (a) single-phase active power, (b) three-phase active power, (c) single-phase reactive power, and (d) three-phase reactive power. Fig. 12. Experimental setup circuit schematic: (a) connection to the single- and (b) three-phase grid. TABLE II POWER COMMANDS SEQUENCE

to respect the rate of change of power for the various devices involved in the experiments. Selecting a larger time window also allows for a more realistic timeframe to transition among the operating quadrants. Here, the results of the physical quantities are presented in order to compare them with the results of Section IV. C. Experiment Results Fig. 13(a) and (c) show the experimental results for singlephase active and reactive power, respectively. The power setpoints are changed sequentially according to Table II. It can be appreciated that four-quadrant operation is obtained, which validates charging and V2G operation in the single-phase grid. In the same way, Fig. 13(b) and (d) show the experimental results for three-phase active and reactive power, respectively. Again, according to Table II, the power setpoint is changed to obtain four-quadrant operation, which validates charging and V2G operation in a three-phase network. Fig. 14 illustrates, over three 50 Hz cycles, experimental waveforms for the VSI voltages and currents in both singleand three-phase networks. Fig. 14(a) shows the VSI voltages when connecting to the single-phase network. For phase a, the voltage is 220 V RMS which corresponds to the grid nominal voltage. For phase b, which is the inverter leg connected to the other wire of the single-phase network, the voltage is zero. For

Fig. 14. Voltage and currents waveform in hardware experimental verification (a) single-phase grid voltage, (b) three-phase grid voltage, (c) single-phase grid current, and (d) three-phase grid current.

phase c, which is the unconnected VSI leg, the resulting voltage is also zero. Fig. 14(c) shows the VSI currents. It can be appreciated the currents through phases a and b are opposite to one another while the current in the open VSI leg is zero. Fig. 14(b) shows the VSI voltages when connecting to the three-phase network. Fig. 14(d) shows the VSI currents, where balanced three-phase currents are recognized. The qualitative agreement of Fig. 14(a) and (b) with the simulations of Fig. 8(a) and (b) for voltages, and Fig. 14(c) and (d) with the simulations of Fig. 9(a) and (b) for currents, is observed. VI. CONCLUSION The integration of on-board charging converters with converters for propulsion into a single unit represents an attractive alternative for the realization of EVs’ power conversion system. The integration is attractive because it leads to savings in cost and weight. For such an integrated converter solution, a novel


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Arnaldo Arancibia received the B.S. and Ing. Civil degrees in electrical engineering from the Universidad de Chile, Santiago, Chile, in 2004 and 2006, respectively. After that he worked for Fundación Chile as a researcher in renewable energy and energy efficiency. Since October 2008 he has been working towards his Ph.D. degree at the Technische Universität Berlin, Germany, where he also works as Research Assistant. He was a Visiting Scholar at L’Ecole supérieure d’électricité (Supélec) in Gif-sur-Yvette, France, and at the University of Colorado, Denver, CO, USA. His research interests are electric vehicles, control systems, distributed generation, and smart grids.


Kai Strunz graduated with the Dipl.-Ing. degree from the University of Saarland, Saarbrücken, Germany, in 1996, and the Dr.-Ing. degree summa cum laude from the same university in 2001. From 1995 to 1997, he pursued research at Brunel University, London, U.K. From 1997 to 2002, he worked at the Division Recherche et Développement of Electricité de France (EDF) in the Paris area. From 2002 to 2007, he was Assistant Professor of Electrical Engineering at the University of Washington, Seattle, WA, USA. Since September 2007, he has been Professor for Sustainable Electric Networks and Sources of Energy at Technische Universität Berlin, Germany. He was the Chair of the IEEE PES Innovative Smart Grid Technologies (ISGT) Europe 2012 in Berlin. Dr. Strunz received the Dr.-Eduard-Martin Award from the University of Saarland in 2002, the National Science Foundation (NSF) CAREER Award in 2003, and the Outstanding Teaching Award from the Department of Electrical Engineering of the University of Washington in 2004.

Fernando Mancilla-David (S’05–M’07) received the B.S. degree in electrical engineering from the Universidad Técnica Federico Santa María, Valparaíso, Chile, in 1999, and the M.S. and Ph.D. degrees in electrical engineering from the University of Wisconsin-Madison, Madison, WI, USA, in 2002 and 2007, respectively. Currently, he is an Assistant Professor at the University of Colorado Denver. He was a visiting scientist at ABB Corporate Research in Västerås, Sweden, in the summer of 2008, and has held visiting professor positions at L’Ecole supérieure d’électricité (Supélec) in Gif-sur-Yvette, France, in the summers of 2009 and 2010, at the Technische Universität Berlin, Berlin, Germany, in the summer of 2011, and at the Università Degli Studi Roma Tre, Rome, Italy in the summer of 2012. His research interests are power system analysis, energy systems, utility applications of power electronics, control systems, and optimization problems.