Application of insulated gate bipolar transistor to zero ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 4, NO. I , JANUARY 1989

Application of Insulated Gate Bipolar Transistor to Zero-Current Switching Converters R. RANGAN, DAN Y. CHEN,

SENIOR MEMBER, IEEE,

JIAN YANG,

Abstruct-The problems associated with insulated gate bipolar transistor (IGBT) devices in PWM converters, such as turn-off current tailing and turn-off latching, are largely avoided in a zero-current switching resonant converter. Phenomena induced by d v l d t , such as the power losses and latching, are identified as the predominant problems in using IGBT devices for very high frequency resonant operations.

I. INTRODUCTION ESONANT converters have recently received considerable attention for high-frequency switching power supply applications. Electromagnetic interference (EMI) reductions, fast response, and high power density are among the advantages in using such converters. However, because of the circuit's operation, the power switch (or switches) in the circuit conducts a much higher current pulse, and consequently, a larger conduction loss exists than in its pulsewidth modulated (PWM) counterpart. Power MOSFET's are capable of very fast switching but are relatively high in conduction resistance. Insulated gate bipolar transistors (IGBT's), on the other hand, are low in conduction resistance but relatively slow in switching speed. The objective of the research leading to this paper is to investigate the suitability of IGBT devices for very high-frequency zero-turrent switching converters. It has been reported that a family of converter configurations operate under the same principle of zero-current switching [1]-[3]. However, the electrical stress on the power semiconductor switch in the family of converters are identical. The discussion and the verification of the results presented here are usually focused on buck-type converters in the zero-current switching family.

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JOHN LEE

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11. RESONANT OPERATION OF IGBT Fig. 1 shows the transistors current and voltage waveform of a full-wave quasi-resonant current converter [11, r2]. The magnitude of the resonant current I, is, theoretManuscript received July 1, 1987; revised March 9, 1988. This work was supported in part by the General Electric Foundation through a grant to the Virginia Polytechnic Institute and State University. This paper was presented at the 1987 IEEE Power Electronics Specialists Conference, Blacksburg, VA, June 21-26. R. Rangan is with Converter Concept, 1 Industrial Park Way, Pardeeville, WI 53954. D. Y. Chen and J. Lee are with the Department of Electrical Engineering, Virginia Polytechnic Institute and State University, Blacksburg, VA 2406 1. J. Yang was with the Virginia Polytechnic Institute and State University, Blacksburg, VA. He i s now with Chan-Sha University, Hunan, China. IEEE Log Number 8823456.

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Fig. 1. Waveforms of full-wave switch.

ically, at least the same as Io, the PWM switch current, to ensure zero-current switching. In reality, however, I, is several times Io. Therefore, the conduction loss of the switch becomes a major portion of the total switch losses in a zero-current switching resonant converter. The symbols used in the figures are defined as follows: VGs gate-source voltage,

,Z drain current, VDs drain source voltage, Zdiode antiparallel diode current. IGBT's are chosen for investigations for resonant converter applications for two reasons. First, the characteristics of low conduction resistance fits well in a resonant application in which a large resonant-current pulse flows through the transistor. Second, the problems associated with IGBT's in conventional PWM converters applications, such as turn-off current tailing and the possibility of latching, are alleviated in zero-current switching quasiresonant converters. In a PWM forced turn-off application, IGBT devices may latch for several reasons such as overcurrent, fast interruption of current, and large du/dt at turn-off. High junction temperature aggravates the situation. In a zero-current switching operation, however, none of the factors mentioned above matter. Overcurrent may cause the device to latch, but because of resonant operation, the device unlatches before zero-current crossing. Because of zero current turn-off, and the diode conductions at the end of device turn off, latching due to turnoff never occurs.

0885-8993/89/0100-0002$01.OO 0 1989 IEEE

RANGAN et al. : APPLICATION OF INSULATED GATE BIPOLAR TRANSISTOR

111. SOURCES OF DEVICE POWERLoss IN RESONANT OPERATION Tables I-IV show the sources and magnitude of transistors power loss under different circuit conditions. Information about MOSFET power losses is included for comparison purposes. In the comparison, 400-V 10-A devices are chosen. A GE IGBT and an RCA COMFET were used. Both are fast switching devices with a turn-off time of approximately 0.8 ps. Note that the MOSFET chip is three to four times as large as the IGBT chip. The operating conditions such as conduction duty cycle, frequence of operation, and voltage and current levels are indicated below each table. These losses were obtained by calculation using experimental waveforms. Each loss component shown in the table will be described in the following. The symbols used in are defined as follows: conduction duty cycle of IGBT and diode, drain-source maximum voltage, fr resonant frequency, fs switching frequency, I,, Io (see Fig. 1).

TABLE I SUMMARY OF DEVICE POWER LOSSES IN WATTS AT 50 kHz (SWITCHING)

Conduction loss Tum-on loss Schottky loss Antiparallel diode conduction loss Antiparallel diode reverse recovery loss dv/dr loss Capacitive loss Total

Turn-On Loss Turn-on loss of a zero-current switching quasi-resonant converter is much lower in comparison to a PWM converter. In a PWM converter, the commutation diode reverse recovery characteristics play an important role in determining turn-on loss. In a resonant converter, however, the problem is much reduced because the diode reverse recovery process is smoothed by the resonant inductor used in the circuit. From the waveform shown in Fig. 3, a significant current-voltage overlap during turn-on observed. The drain source voltage is initially very high, from 3 to 12 V, depending on circuit operating conditions during the last several hundred nanoseconds. It was first believed that this is caused by the dynamic saturation of an IGBT and large power loss would occur. It turned out that the current-voltage waveforms overlapping was due to the parasitic inductance of the IGBT package (approximately 20 nH). A similar phenomenon occurs in a MOSFET resonant converter operating under the same circuit condi-

MOSFET (IRF 450)

6.0 negligible

9 negligible 1.8 0.4 0.2

-

0.4 0.2 negligible negligible 6.6

Operation conditions: D , = 0.45, jr = 60 kHz, & 200 V, I,,, = 7 . 5 A, Io = 2 . 2 A .

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TABLE I1 SUMMARY OF DEVICE POWER LOSSES IN WATTSAT 150 kHz (SWITCHING)

Do,, V,,

Conduction Loss Fig. 2 shows the I-V characteristics of an IGBT and a MOSFET. Both are 400-V 10-A devices. Note that there is a forward threshold voltage, VDsof approximately 0.7 V to keep the IGBT in conduction, which is not needed in a MOSFET. However, at high current levels, the conduction drop of an IGBT is much smaller than that of a comparable MOSFET. The relative magnitude of the conduction loss and total losses depend on switching frequency, conduction duty cycle and the current level involved. As shown in Table 1, 90 percent of the total losses is in conduction loss at 50 kHz, the percentage drops to 37 percent and, at 500 kHz, the percentage drops much further to about 20 percent.

GE IGBT (4D 10)

Conduction loss Tum-on loss Schottky loss Antiparallel diode conduction loss Antiparallel diode reverse recovery loss dv/dr loss Capacitive loss Total

GE IGBT (4D 10)

MOSFET (IRF 450)

6 0.3

9 negligible 1.5 0.4 0.5

-

0.4 0.5 1

0.5 8.7

-

1.4 12.8

Operation conditions: Don = 0 . 4 5 , f , = 180 kHz,& = 150 kHz, VD, = 200 V, I,,, = 7.5 A, Io = 2 . 2 A .

TABLE 111 SUMMARY OF DEVICE POWER LOSSES IN WATTS AT 250 kHz (SWITCHING)

Conduction loss Tum-on loss Schottky conduction loss Antiparallel diode conduction loss Antiparallel diode reverse recovery loss dv/dr loss Capacitive loss Total Operation conditions: Do, = 0.30, f, = = 200 V, I,,, = 1.5 A, Io = 2 . 2 A.

RCA (FAST IGBT)

(IRF 450)

4 0.8

6 negligible

-

0.2 0.6 4.5 0.9 11

MOSFET

1

0.2 0.6

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2.2 10.1

500 MHz, & = 250 kHz, VDs

TABLE IV SUMMARY OF DEVICE POWER LOSSES IN WATTS AT 500 kHz (SWITCHING)

Conduction loss Tum-on loss Schottky conduction loss Antiparallel diode conduction loss Antiparallel diode reverse recovery loss dv/dr loss Capacitive loss Total

RCA (FAST IGBT)

MOSFET (IRF 450)

4 2.3

6 0.3 1 0.2 2.5

-

0.2 2.4 10 1.I 20.7

-

4.4 14.4

Operation conditions: Do, = 0.30, f r = 1 MHz, & = 500 kHz, VDS = 200 V, I,,, = 6.6 A, lo = 3 . 3 A.

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 4. NO. I . JANUARY 1989

(b)

Fig. 2. I-Vcharacteristics. (a) IGBT (GE 4D10). (b) MOSFET (IRF 450). Both devices are rated 400 V, 10 A . IGBT chip area is about 1 / 3 MOSFET chip. Vertical 1, = 1 A/div. Horizontal VDs = 0.5 V/div.

tions, but the effect is not as pronounced as in an IGBT converter. This is because the high conduction resistance of the MOSFET dilutes the effect of the parasitic inductance. In any event such waveforms overlapping do not contribute to power loss. Tum-on loss is therefore determined by the rise time of the device which is closely related to gate drive circuit capability. In the calculation shown in Tables I-IV, 50ns rise time was used for IGBT and 15 ns was used for MOSFET. From the device point of view, there is a limit on tum-on rise time of IGBT because of the transit time of bipolar action. The time for such action in the IGBT used is approximately 50 ns. It can be seen from Tables I-IV that the turn-on loss in IGBT becomes noticeable only when the frequency exceeds 250 kHz.

Schottky Diode Loss A series-connected Schottky diode is used to bypass the integral diode of the MOSFET. This is necessary to avoid a large diode reverse-recovery loss due to the slow recovery of the integral diode. Shottky diode conduction loss should be included in a comparison in a MOSFET converter. Since an integral diode does not exist in an IGBT, there is no need to add a Schottky diode in the IGBT circuit. Antiparallel Diode Conduction Loss Diode conduction loss depends on diode current, diode conduction voltage drop, and the diode conduction cycle.

(c) Fig. 3. Parasitic inductance effect of device. Arrows indicate large voltage drop immediately afer turn-on. (a) Y: 5 V/div, I: 1 A/div, r: 2 ns/div. (b) V: 2 V/div, I : 1 A/div, t : 500 +s/div. (c) V: 5 V/div, I: 1 A/div, f: 200 ns/div.

The diode duty cycle increases with a decreasing load. Even under open-load condition diode conduction loss is still considerably less than transistor conduction loss because of small conduction voltage drop.

IGBT Reverse Conduction Loss When the resonant current flows reversibly through the switch, the current is shared by the antiparallels diode and IGBT reverse conduction current. Depending upon the device stored charge at the instant of reverse current flow, the current shared by the IGBT could be significant, as shown in Fig. 4. In general, the slower the IGBT, or higher the resonant frequency, the larger the share of the IGBT reverse current. IGBT reverse current is similar to SCR recovery current. From the loss point of view, however, there is no significant change whether the IGBT conducts reverse current or the antiparallel diode takes the share. This loss is therefore included in the diode conduction loss.

RANGAN er al. : APPLICATION OF INSULATED GATE BIPOLAR TRANSISTOR

Fig. 4. Reverse conduction of IGBT (GE-4El-slow IGBT). VGs: 5 V/div, Vas: 2 V/div, I,: 1 A/div, t: 500 ns/div.

Antiparallel Diode Recovery Loss In a full-wave resonant switch, the current in the switch flows in both directions. After the end of diode forward conduction, the resonant current flows through the diode in reverse direction until the diode recovers. Reverse recovery loss occurs during the recovery phase. This loss can become significant at very high frequencies as shown in Table IV. In calculations that lead to Tables I-IV, a 200-ns reverse recovery diode is used. Antiparallel diode reverse recovery becomes a very important consideration in a zero-current full-wave switching converter. It is not only the direct switching loss associated with the recovery but also the voltage spike it generates. A “snappy” diode generates large d v l d t that, in tum, cause large power losses in the IGBT as will be explained in the next section. d v l d t Loss At the end of the reverse-recovery phase of the antiparallel diode,the device voltage starts to rise. A large rate of voltage rise induces a current spike in the IGBT device which leads to a power loss. The magnitudes of the current spike depends on the amount of charge stored in the IGBT at this instant. The slower the device and/or the higher the resonant frequency, the worse the current spike will be. Three factors are involved in determining the d v l d t current spike. One factor is the magnitude of d v l d t caused by the recovery of the diode and the dildt of the resonant current at zero crossing. A second factor is the device recombination rate and the third factor is the time determined t, by the circuit operating conditions. Fig. 5 shows that reverse recovery generates a large d v l d t and Fig. 6 shows d v l d t cause a large current spike in the IGBT. This current spike causes a large power loss and is the major device loss in very high-frequency resonant operation. When a slow IGBT is used in a high-frequency resonant circuit, the device is never really turned off even though the current goes through zero-current reversal. The current spike generated by d v l d t is reduced to zero by recombination, much like forced turn-off. Since the frequency is so high the current never reduces to zero before the device is turned on again. Fig. 7 shows waveforms under such a condition.

Fig. 5 . Antiparallel diode reverse recovery and drain-to-source dv/dt. V: 50 V/div, I: 1 A/div, t: 200 ns/div.

Fig. 6. dv/dr-generated current spike in COMFET. V: 50 V/div, I: 1 A/div, f: 200 ns/div.

Fig. 7 . Waveforms for slow device operating at high frequency.

Capacitive Losses Charging and discharging the device parasitic are necessary to switch the IGBT. The parasitic capacitances of an IGBT is inherently smaller than an equivalently rated MOSFET because a smaller chip area is required. The parasitic capacitances are highly nonlinear in nature. The capacitive losses are estimated using the “charge” concept to take nonlinearity into account. The loss, however, is insignificant in IGBT’s even up to 500 kHz at 200 V. In case of a MOSFET, the capacitive loss is a considerable portion of the total loss at high-frequency and highvoltage application. Resonant Operation Versus PWM Operation Table V summarized a comparison of semiconductor device losses in a resonant operation and a PWM operation. IGBT devices are used as the power switch. Device losses depend on current conduction level, duty cycle,

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 4, NO. I , JANUARY 1989

V. CONCLUSION Several conclusions are drawn from this investigation. The first is that the problems associated with IGBT turnPWM Resonant off in a PWM converter, namely, turn-off current tailing Operation Operation and latching, have been alleviated in a resonant operation. (40 kHz ) (200 kHz) Existing IGBT devices can be operated in a zero-curConduction loss 3 3.7 rent switching quasi-resonant converter in the 100-kHz Tum-on loss negligible negligible range with good conversion efficiency. As a point of refTurn-off loss 3 0 erence, at 200 V, 3 A (3 A for PWM, 8 A for the resonant Diode conduction loss (antiparallel diode) 0 0.2 converter), 50 percent conduction duty cycle, the converReverse recovery loss sion efficiency for a 40-kHz PWM converter is about the antiparallel diode) 0 0.5 same as that of a 200-kHz quasi-resonant zero-current d v / d t loss 0 2.4 Reverse recovery loss switching converter. However, it should be pointed out (commutation that additional components are needed in a resonant condiode) 1.8 negligible verter and additional components power losses, particuCapacitive loss negligible 1.4 larly the resonant capacitor loss must be taken into acTotal 7.8 8.2 count in making the choice betwen the PWM converter Operation conditions: V, = 200 V, Io = 3 A, I,,, = 5 A, D = 0.50, I, and the resonant converter. = 3 A , f , = 40 kHz, D = 0.50,I,,, = 5 A, ( D a = 0.36, D, = 0.14),f, = 200 kHz. The comparison of MOSFET’s and IGBT’s in a resonant operation differs with frequency of operation, conduction duty cycle, source voltage, and conduction curvoltage level, and switching frequency. The results shown rent level. At 200 V, 10 A peak, and 40-percent duty in the table are based on the assumptions that both PWM cycle, the crossover frequency is about 250 kHz. Note converter and zero-current resonant converter have iden- that in this comparison, the MOSFET chip size is about tical input and output conditions (input voltage, output three times that of the IGBT chip. As the frequency involtage, and output power). At 200-V source voltage, 3- creases or the current level decreases, the comparison is A conduction current level ( 3 A in PWM converter, 8 A tilted in favor of MOSFET’s. The effect of source voltage peak in resonant converter), conversion efficiency of a on the comparison is more difficult to assess. As the volt40-kHz PWM converter is about the same as a 200-kHz age level becomes higher, the conduction resistance inzero-current quasi-resonant converter. It should be pointed creases if the chip area remains the same. The conduction out that the comparison shown in the table is concerned loss increases at a much faster rate in MOSFET’s. Howwith semiconductor losses only. In a quasi-resonant con- ever, dv/dt loss in an IGBT also increases with voltage verter, resonant capacitor loss could be very significant, level. depending on the capacitor type used and the magnitude The major limitation of the IGBT in a very high-freof resonant current. Nevertheless, Table V gives a quick quency ( < 250 kHz) zero-current switching full-wave comparison between the two operations. resonant operation are as follows. 1) dv/dt Induced Power Loss: The loss is determined by four factors. IV. LIMITATIONS OF IGBT’s IN HIGH-FREQUENCY OPERATION a) Reverse characteristics of the anti-parallel diode: In a conventional PWM converter operation, the turnThe softer the recovery, the lower the loss. off loss of an IGBT is the major portion of the total losses b) Frequency of operation: The loss is more than at high frequency. This prevents the devices from being proportional to frequency. operated beyond the 50-kHz range. For very high-frec) Device characteristics: The longer the minority quency resonant operation ( > 250 kHz), however, three carrier lifetime, the more the loss. limiting factors were identified for existing devices. One d) Source voltage level: The higher the voltage level, of the limiting factors is the turn-on loss. The other two the more loss occurs. factors are related to the dv/dt phenomenon occuring at 2) dv/dt Induced Latching: The four factors menthe instant of reverse recovery of the antiparallel diode. One of the two factors is the power loss associated with tioned in 1) all contribute to the possibility of latching. the dv/dt induced current conduction. As can be seen However, compared to forced turn-off latching in a PWM from Table I, dv/dt loss becomes predominant power loss operation, the device is less likely to latch under dv/dt beyond 250 kHz. At even higher switching frequency, the condition. 3) Turn-On Loss: For frequency beyond 500 kHz, device never really cuts off due to the current tailing of the dv/dt induced current conduction. The other factor turn-on loss becomes significant. To improve high-frequency operation performance of limiting the frequency of operation is the dvldt-caused latching. The higher the resonant frequency, the smaller an IGBT in a full-wave zero-current switching resonant operation, dv/dt performance is inevitably accompanied the tq and the more prone the device is to latching. TABLE V COMPARISON OF DEVICE LOSSESI N PWM OPERATION AND I N RESONANT OPERATION IGBT (GEAD10)

RANGAN et al. : APPLICATION OF INSULATED GATE BIPOLAR TRANSISTOR

by a rise in device conduction resistance. The trade-off between d v / d t loss and conduction loss depends on application conditions. A practical way to reduce the problems associated with dv/dt is to use a soft recovery antiparallel diode.

REFERENCES [ l ] K. Liu, R. Oruganti, and F. C. Lee, “Resonant switches-Topologies and characteristics,” in IEEE PESC Rec., 1985, pp. 106-1 16. [Z] T. Zheng, D. Chen, and F. C. Lee, “Variations of quasi-resonant dcdc converters topologies,” in IEEE PESC Rec., 1986. [3] K. H. Lui and F. C. Lee, “Secondary-side resonant for high frequency converters,” in IEEE Applied Power Electronics Con5 Proc., 1986, pp. 83-89. Ramasamy Rangan was bom in Tiruchy, India. He received the B.S.E.E. degree from Regional Engineering College, Surathkal, India, and the M.S.E.E. degree from the Virginia Polytechnic Institute and State University, Blacksburg, in 1987. He was employed as a Design Engineer from 1985 to 1986 with TRW, Electronic Assembly Division, Schaumburg, IL, and from 1987 to 1988 with NJE Corporation in Dayton, NJ. He is currently a Design Engineer with Converter Concept

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Blacksburg, where he is presently a Professor. Since the start of his graduate study in 1970, he has been working in various fields of power electronics. His activities have included work in power semiconductor circuits, circuit-device interactions, device characterization, magnetic devices, magnetic amplifier applications, and product applications such as brushless motor robotic drive, electronic ballast, appliance power supply, electric car drive, etc. Dr. Chen has published more than 50 papers and has been awarded four U.S. patents and has one patent pending, all in the field of power electronics.

Jian Yang received the B.S.E.E. degree from Chan Sha University, Hunan, China, in 1984. He was a Visiting Scholar at the Virginia Polytechnic Institute and State University, Blacksburg, between 1986 and 1987. He is presently an Instructor at Chan-Sha University, Hunan, China.

in Pardeeville, WI. Dan Y. Chen (S’72-M’75-M’79-SM’83) received the B.S. degree from National Chiao-Tung University, Taiwan, in 1969 and the Ph.D. degree from Duke University, Durham, NC, in 1975, both in electrical engineering. From 1975 to 1979, he was employed as a Member of the Research Staff at the General Electric Research and Development Center, Schenectady, NY. Since 1979, he has been on the faculty of the Department of Electrical Engineering, Virginia Polytechnic Institute and State University,

John C. Lee was bom in Taiwan in December 1953. He received the B.S. degree from ChunShin University, Taiwan, in 1976 and is currently working toward the M.S. degree in electrical engineering at the Virginia Polytechnic Institute and State University, Blacksburg. He joined the Virginia Power Electronics Center in 1986.