Bidirectional Current-Fed Resonant Inverter for ... - IEEE Xplore

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 1, JANUARY 2015

Bidirectional Current-Fed Resonant Inverter for Contactless Energy Transfer Systems Alireza Namadmalan

Abstract—This paper presents a bidirectional currentfed parallel resonant push–pull inverter (CFPRPI) for contactless energy conversions. In conventional CFPRPIs, each switch is in series with a blocking diode. Direct zero voltage switching (ZVS) is proposed to remove the series blocking diodes for higher efficiency and boosting ratio. In addition, the modified inverter has bidirectional power flow capability when these blocking diodes are omitted. Resonant tuning of the system is investigated by using a phase-locked loop and a new self-oscillating switching technique (SST). By using the new SST, the inverter has direct ZVS in a wide range of operating frequencies, which is necessary for inductively coupled power transfer systems. Finally, a laboratory prototype with an input voltage of 12 V and a maximum output power of 60 W has been implemented. Index Terms—Bidirectional, contactless energy transfer, current-fed resonant inverter, phase-locked loop (PLL) circuits.

I. I NTRODUCTION

I

NDUCTIVELY coupled power transfer (ICPT) systems are widely utilized in industrial and commercial appliances such as robotics, hybrid vehicles, and mobile devices [1], [2]. Among them, there are some applications that require bidirectional power flow capability such as electric vehicles and smart-grid applications [3]–[6]. ICPT systems are commonly divided into series–series (SS), series–parallel, parallel–series (PS), and parallel–parallel (PP) resonant topologies (the first letter and the second letter denote primary and secondary compensations, respectively) [1], [2], [7], [8]. Parallel resonant topologies are widely applied in highpower applications because they are more reliable and costeffective. In comparison with the series topologies, a parallel resonant converter has smooth input current, low current stress for switches, no paralleling problems, and short-circuited protection capability [9]–[13]. Hence, parallel compensation is a suitable topology for the primary side of ICPT systems. In addition, the current ratio and the efficiency of the PS topology is the same as those of the SS topology [8]. Manuscript received October 9, 2013; revised March 26, 2014; accepted April 28, 2014. Date of publication May 30, 2014; date of current version December 19, 2014. The author is with the Department of Electrical and Computer Engineering, Jundi-Shapur University of Technology, Dezful 64615-334, Iran, and also with the Research and Development Center, Damavand Induction Furnace Company, Damavand 1657137336, Iran (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2014.2327593

Fig. 1. Two ICPT systems based on the modified CFPRPI: (a) with a bidirectional structure, and (b) with a unidirectional structure.

In practice, the parameters of high-frequency transformers (HFTs) extremely vary due to the air-gap variation and misalignment between the primary and secondary sides of the HFTs [2], [14]–[18]. Hence, improved tolerance to the misalignments and coupling variations are the major concerns in ICPT systems [1], [2]. The most important problems associated with the parallel resonant inverters are the resonant frequency, i.e., fr , the tuning loop, and unidirectional power conversion due to series blocking diodes [9]–[13]. The variations of an HFT make the control and tuning of the PP and PS topologies more complex [2], [7]. On the other hand, a tuning loop is necessary to track fr in a wide range of variations with direct zero-voltage switching (ZVS) at the start-up condition [2]. Under transient conditions, direct ZVS is not achieved due to the poor dynamics of phase-locked loops (PLLs) [2], [11], [19], [20]. Hence, in a conventional current-fed parallel resonant push–pull inverter (CFPRPI), two series-connected diodes are required to prevent the internal body diode of the switches from conducting [2], [11]–[13]. To improve the dynamics of the PLLs, the multiplier phase detector (MPD) method is utilized. However, a PLL based on the MPD method has significant steady-state phase error for large deviations of the tank circuit parameters [11]. The series blocking diodes increase the conduction losses and decrease the boosting ratio. By achieving direct ZVS, the switches are used without the series blocking diodes; hence, the modified CFPRPI can be utilized in bidirectional ICPT systems. Fig. 1 shows two applications of the modified inverter in ICPT systems, where Fig. 1(a) is based on the bidirectional capability,

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NAMADMALAN: BIDIRECTIONAL CURRENT-FED RESONANT INVERTER FOR ENERGY TRANSFER SYSTEMS

Fig. 3.

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Drain-to-source voltage and anode-to-source voltage of S1 .

Fig. 2. Conventional CFPRPI operation for a half-cycle. (a) Operation during 0 < t < T /4, where S1 is OFF, S2 is ON, and CDS is charged. (b) Operation during T /4 < t < T /2, where D1 is turned OFF, and CDS keeps the drain-to-source voltage of S1 near the peak value, i.e., Vm .

and Fig. 1(b) is based on an unregulated dc bus for fixed airgap applications. Conventionally, the primary and secondary compensations of the bidirectional ICPT systems are based on LLC circuits [3]–[5], whereas the proposed bidirectional ICPT system, as shown in Fig. 1(a), has a simple structure, and there is no series matching inductor. This paper is organized as follows. Section II describes the system modeling of the modified CFPRPI and the ICPT systems shown in Fig. 1. Section III investigates the direct ZVS using the proposed self-oscillating switching technique (SST) and the PLL circuit based on the MPD. Sections IV and V present the experimental results and main conclusions of this paper, respectively. II. M ODIFIED CFPRPI A. Review of Conventional CFPRPI In order to introduce the modified CFPRPI, first, a brief review of the conventional CFPRPI is set forth in this section. Due to the poor dynamic response of a PLL circuit, seriesconnected diodes similar to the diodes shown in Fig. 2 (D1 and D2 ) are required. To demonstrate the adverse effects of the blocking diodes, the operation of this inverter for the first half of a switching period, i.e., T , is studied. Fig. 2 shows the circuit operation of the conventional inverter when its output is directly connected to the PS resonant circuit (load). According to Fig. 2(a), during 0 < t < T /4 when switch S1 is OFF and switch S2 is ON, the drain-to-source capacitor, i.e., CDS , of S1 is charged. When the voltage of the load reaches its peak value, i.e., Vm , at t = T /4, D1 is turned OFF, and based on Fig. 2(b), CDS keeps the drain-to-source voltage of S1 nearly at the peak value for T /4 < t < T /2. However, CDS is instantly discharged at t = T /2 when S1 is turned ON, which increases the switching losses. Fig. 3 displays the drain-to-source voltage, i.e., VDS , and the anode-to-source voltage, i.e., VAS , waveforms for switch S1 during two switching periods with an operating frequency of 100 kHz and an input voltage of 12 V (S1 and S2 are IRFP250, D1 and D2 are BYV32E). B. Modified CFPRPI The aim of this paper is to achieve direct ZVS using a novel tuning loop in order to eliminate the need for the series blocking

Fig. 4. (a) and (b) Forward operation of the modified CFPRPI. (c) and (d) Backward operation of the modified CFPRPI.

diodes. Based on this idea, the modified CFPRPI shown in Fig. 4 is devised. Unlike the conventional topology, only two active switches are used in the modified topology. Inductors Ld1 and Ld2 (in Fig. 4) are designed to have much larger inductance than the inductance of the resonant inductor in order to have an almost constant dc link current, i.e., Id . Therefore, during the steady-state operation, the switching network injects an alternating square-wave current into the resonant tank. If the modified inverter is used in a topology similar to the topologies shown in Fig. 1, the resonant inductance is equal to the inductance of the primary coil. Fig. 4(a) and (b) illustrates the two modes of operation of the modified CFPRPI in the forward current flow direction. The two active switches (S1 and S2 ) are switched with duty cycles that are slightly more than half in order to achieve the soft switching and prevent the internal body diode of the power MOSFETs from conducting [11], [13]. During the first mode of operation, i.e., 0 < t < T /2, shown in Fig. 4(a), S1 is OFF, S2 is ON, and the current of inductor Ld1 is injected into the parallel resonant tank and returned back to the dc source through S2 . Similarly, during T /2 < t < T , which is shown in Fig. 4(b), S1 is ON, S2 is OFF, and the current of inductor Ld2 is injected into the parallel resonant tank and

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Fig. 7. Steady-state model of the inverter and the Is and PS compensation topology of the ICPT system based on the contactless HFT parameters. Fig. 5.

Drain-to-source voltage of S1 for the modified CFPRPI.

Fig. 6. Equivalent circuit of the CFPRPI seen from the dc-link side. (a) Conventional inverter. (b) Modified inverter. R2 is the value of the MOSFETs’ ON-resistance, R1 is the value of the dc-link inductor resistance, and VD is the value of the series diodes’ forward voltage drop.

returned back to the dc source through S1 . In each mode of operation, the peak current of the switches is equal to Id . Fig. 4(c) and (d) illustrates the two modes of operation of the modified inverter in the backward current flow direction. In the backward direction, the modified CFPRPI operates similar to a synchronous rectifier and transfers the power from the output terminals to the dc-link terminals. The modes of operation of the inverter in the backward direction are similar to the forward direction, meaning that the soft switching is achieved in the backward operation. Unlike the conventional CFPRPI, the drain-to-source voltage of the modified inverter is gradually decreased during T /4 < t < T /2 because there is no series blocking diode. Fig. 5 shows VDS for the modified inverter. Fig. 6 compares the steady-state equivalent circuit, which is seen from the dc-link side, of a conventional CFPRPI with the modified CFPRPI. The circuit in Fig. 6(a) is related to the conventional inverter and models the effect of the forward voltage drop of the series blocking diodes, i.e., VD , with an R2 resistance term of the power MOSFETs. By removing the series diodes, the saved power is VD Id . For a Schottky diode, which is typically used in this kind of inverter, VD is around 0.625 to 1.5 V, depending on the current rating. Hence, the net efficiency of the conventional inverter is decreased by 5.2%–12.5% for Vin = 12 V. As a result, the net efficiency and boosting ratio of the modified inverter is significantly higher than those of the conventional inverter for low-input-voltage applications. C. ICPT System Based on Modified CFPRPI As aforementioned, this inverter behaves as an alternating square current source with a peak value equal to the half of the dc-link current. The circuit in Fig. 7 is comprised of a squarewave current source Is representing the switching network, which is connected to a PS compensating circuit. Lm , L11 , and L22 are the magnetizing inductance referred to the primary of

the contactless HFT, the leakage inductance of the primary of the contactless HFT, and that of the secondary of the contactless HFT, respectively.Rac , which is shown in Fig. 7, is the ac-load resistance at the secondary side of the contactless HFT (see Fig. 1). Considering that the HFT behaves similar to a symmetrical two-port network, the relationship between the HFT parameters is derived as follows: L1 = L11 +Lm , L2 = L22 +n2 Lm

n2 nLm ; k= √ . (1) n1 L1 L2

L22 = n2 L11 ; n =

where k, n, L1 , and L2 are the coupling factor, the turn ratio of the HFT, and the inductance of the primary and secondary coils, respectively. Series capacitor Cs compensates the series impedance seen from the secondary side or the series reactance caused by L22 . Hence, a resistive load is referred to the primary side of the HFT. Consequently, the resonant frequency fr of the ICPT system is the resonant frequency of the parallel LC at the primary side in Fig. 7, which can be found from 1 1 = . (2) fr ≈ 2π L1 Cp 2π (L11 + Lm )Cp The angular resonant frequency is then equal to ωr = 2πfr .

(3)

The parallel tank circuit at the primary side behaves similar to an open circuit at the resonant frequency. Hence, the impedance of Z2 is derived by the following, meaning that the series capacitor compensates the reactance of X22 : Z2 = jX22 ,

X22 = ωr L22 .

(4)

According to (3), the series capacitor is derived as follows: XCs = X22 ,

XCs =

1 . ωr C s

(5)

Considering (1), (2), and (5), the relationship between parallel capacitor Cp and series capacitor Cs is derived by Cs =

Cp . n2 (1 − k)

(6)

The ac-load voltage V2 to the primary voltage V1 ratio or the voltage gain, i.e., Gv , of the PS topology is derived as follows: jXm Rnac V2 2 = n Gv = jX11 + jXm Rnac V1 2 Rac jXm . (7) = n 2 jRac Xm + jX11 (Rac + n jXm )


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following equation formulates the dc-link current as a function of time, i.e., id (t), during the first half of the switching period, where Id is the average value of the dc-link current:

t id (t) =

Vin dt + Ld2

0

t

Vin −V1 dt + Id Ld1

for 0 < t
VZ . (31) 2n1 Qωr L1 2Cp Regarding (31), maximum quality factor Qmax , at which the system is stable, is derived as follows: Ld n3 π 3 Vin . (32) → Qmax = 2n1 VZ ωr L1 2Cp


Fig. 10. Voltage and current of S2 at the start-up condition by using the SST with an operating frequency of 100 kHz, Ld = 800 μH, and Q ≈ 5.

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Fig. 11. Experimental setup of the ICPT system and the tuning loop based on the SST. TABLE I D ESIGN I NFORMATION OF THE SST AND THE ICPT S YSTEM

Similar to other self-oscillating techniques, there is an intrinsic time delay Td mainly due to the turning on of the power MOSFETs, gate drivers, logic, and zero detector circuits [10]. Hence, there is an intrinsic phase error between the injected current and the output voltage, i.e., β . The intrinsic phase error is derived by the following when the system operates at an angular frequency of ω: β = exp(−jωTd ) (rad).

(33)

The intrinsic time delay is typically between 100 and 200 ns regarding the practical data sheets for the SST elements. Hence, according to (33), the intrinsic phase error will be between −3.6◦ and −7.2◦ at an operating frequency of 100 kHz. To attenuate the intrinsic phase error, an RC compensator is proposed, as shown in Fig. 9. Considering the compensator, final phase error β is derived by exp(−jωTd )(jRt2 ωτ + R) β ≈ , jRt2 ωτ + R + Rt2 τ = RCt ;

R = (Rt1 RZ )

(34)

where Rt1 and Rt2 are the tuning resistors, and Ct is the tuning capacitor. Regarding the RC compensator, the phase error can be suppressed in a wide range of operating frequencies. Similar to the work in [10], S1 and S2 are switched with a duty cycle that is slightly more than half for a wide range of operating frequencies. Regarding (34) and considering Rt1 = 5.6 kΩ, Rt2 = 1 kΩ, RZ = 10 kΩ, and Ct = 820 pF, the maximum phase error is less than 3◦ for operating frequencies of 20–100 kHz. In addition, the operating frequency of ICPT systems is typically under 100 kHz [1], [2]; hence, the phase error of the SST is negligible in these applications. Fig. 10 shows the voltage and current of S2 at the start-up condition using the proposed tuning loop. In the following simulation, L1 = L2 = 12 μH, Cp is 0.22 μF, k is 0.4, Rac is 6 Ω, and Ii ≈ 0.2 A. In Fig. 10, the tuning system has no RC compensator because the tuning loop is considered with no intrinsic time delay and the zero detector works ideally.

IV. E XPERIMENTAL R ESULTS A laboratory prototype has been developed with and without the series blocking diodes. For simplicity, the load is an acload resistor Rac in series with Cs without a rectifier unit. Fig. 11 shows the CFPRPI, the PS compensator, the ac load, and the HFT. The resonant tuning of the modified CFPRPI is investigated with the PLL (based on the MPD mode) and the proposed SST. Two separate ferrite pot cores with a diameter of 3 cm are used to construct the HFT. The primary and secondary coils are wound with seven insulated strands of AWG 27 to reduce the skin effect. The parameters of the inverter, the SST, and the PS compensator are given in Table I. For n1 = n2 = 12 turns and an air-gap length of 3.5 mm, L1 and the coupling factor of the HFT are about 12μH and 0.4, respectively. The resonant frequency of the system changes according to the air-gap length of the HFT. The performance of the SST is investigated in operating frequencies of 20–100 kHz. The maximum output power is about 60 W and occurs in a coupling factor of 0.75. Fig. 12(a) and (b) shows the current and voltage waveforms of S1 at the start-up and steady-state conditions with fr ≈ 100 kHz. Fig. 12(a) shows that the modified CFPRPI has direct ZVS using the proposed SST. To show the bidirectional capability of the modified inverter, the step response of Id

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Fig. 12. (a) Voltage and current of S1 at the start-up using the SST with a quality factor of about 5, (Upper trace: 10 V/Div), (Lower trace: 1 A/Div), and (Time base: 10 μs/Div). (b) Voltage and current of S1 at the steady-state condition using the SST, with (Upper trace: 10 V/Div), (Lower trace: 5 A/Div), and (Time base: 1 μs/Div). (c) and (d) Step response of dc-link current Id at a light-load condition using the SST, with (c) based on the modified CFPRPI, (d) based on the conventional CFPRPI, (Trace: 1 A/Div), (Time base: 100 μs/Div), Cp ≈ 1 μF, Rac = 25 Ω, and Vin = 24 V. (e) and (f) Voltage and current of S1 at the start-up condition based on the modified inverter by using the SST, with Vin = 24 V, Q ≈ 55, and a resonant frequency of about 22 kHz, (Upper trace: 50 V/Div), (Lower trace: 2 A/Div), and [Time base: 50 μs/Div for (e) and 25 μs/Div for (f)].

Fig. 13. Voltage and current of S2 at start-up. (a) Using the PLL. (b) Using the SST. (Upper trace: 10 V/Div), (Lower trace: 10 A/Div), and (Time base: 5 μs/Div).

is investigated with a light-load condition. Fig. 12(c) and (d) shows Id with and without the series blocking diodes using the SST. Fig. 12(c) shows that the energy of the tank circuit can return to the dc-link terminals. To show the performance of the SST at high quality factors, a different parallel load with Q ≈ 55 and fr ≈ 22 kHz is connected to the modified CFPRPI. Fig. 12(e) and (f) shows the current and voltage of S1 under the load condition. This shows that the proposed tuning loop is not sensitive to a large deviation of the resonant tank parameters and that the system has the bidirectional capability. Fig. 13(a) shows the current and voltage of S2 at the start-up condition by using the PLL with fr ≈ 100 kHz, Δf ≈ 5 kHz, and ΔF ≈ 180 kHz [for the best case, see (23)]. The PLL circuit is implemented using an HCT4046A integrated circuit

Fig. 14. Efficiency of the ICPT system based on the conventional and modified CFPRPIs with the design parameters in Table I.

and operates based on the MPD mode. Fig. 13(a) shows that the PLL circuit cannot be used for an ICPT system with larger deviations; hence, switch failure can occur. Fig. 13(b) shows the performance of the SST with the same load. Fig. 14 shows the efficiency of the ICPT based on the modified and conventional CFPRPIs. In this condition, the coupling factor is 0.4 with an operating frequency of about 100 kHz. Regarding Fig. 6(b) and (25), the total power losses, i.e., Ploss , of the modified inverter is derived from the following: R1 Ploss ≈ 2Psw + Id2 R2 + + PHFT 2 2 Rw (πVin )2 Rw πVin PHFT ≈ ≈ . (35) 2|Z1 |2 2 ω r L1


PHFT and Rw are the power loss and ac-resistance of the primary winding of the HFT at an operating frequency of 100 kHz, respectively. In this condition, Vin = 12 V, R1 = 0.2 Ω, R2 = 0.08 Ω, and Cp and Cs are about 0.22 and 0.33 μF, respectively. The switching power loss, i.e., Psw , is derived by (25) considering the worst case (i.e., β = 3◦ ). Regarding (35), the power loss of the HFT is mainly due to the ohmic loss caused by the magnetizing current at the primary side, and Rw is about 45 mΩ. As shown in Fig. 14, the efficiency of the system significantly increases for an input voltage of 12 V. V. C ONCLUSION This paper has presented a bidirectional ICPT system based on the CFPRPI with a new resonant frequency tuning loop. PLLs that are based on integral phase detectors have poor dynamics in comparison with self-oscillating techniques. Although the MPDs have relatively fast dynamics, there are large phase errors for typical deviations and misalignments. By using the proposed SST, direct ZVS is achieved in a wide range of operating frequencies with negligible phase error at steady-state and transient conditions. Hence, the series-connected diodes are removed to achieve better efficiency, a higher boosting ratio, and bidirectional capability. In addition, the proposed SST can be also utilized for full-bridge parallel resonant inverters for electric vehicle applications. R EFERENCES [1] M. P. Kazmierkowski and A. J. Moradewicz, “Unplugged but connected: Review of contactless energy transfer systems,” IEEE Ind. Electron. Mag., vol. 6, no. 4, pp. 47–55, Dec. 2012. [2] G. A. Covic and J. T. Boys, “Inductive power transfer,” Proc. IEEE, vol. 101, no. 6, pp. 1276–1289, Jun. 2013. [3] A. K. Swain, M. J. Neath, U. K. Madawala, and D. J. Thrimawithana, “A dynamic multivariable state-space model for bidirectional inductive power transfer systems,” IEEE Trans. Power Electron., vol. 27, no. 11, pp. 4772–4780, Nov. 2012. [4] U. K. Madawala, M. Neath, and D. J. Thrimawithana, “A power-frequency controller for bidirectional inductive power transfer systems,” IEEE Trans. Ind. Electron., vol. 60, no. 1, pp. 310–317, Jan. 2013. [5] D. J. Thrimawithana, U. K. Madawala, and M. Neath, “A synchronization technique for bidirectional IPT systems,” IEEE Trans. Ind. Electron., vol. 60, no. 1, pp. 310–317, Jan. 2013. [6] K. Thirugnanam, T. P. E. R. Joy, M. Singh, and P. Kumar, “Modeling and control of contactless based smart charging station in V2G scenario,” IEEE Trans. Smart Grid, vol. 5, no. 1, pp. 337–348, Jan. 2014. [7] A. J. Moradewicz and M. P. Kazmierkowski, “Contactless energy transfer system with FPGA-controlled resonant converter,” IEEE Trans. Ind. Electron., vol. 57, no. 9, pp. 3181–3190, Sep. 2010.

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Alireza Namadmalan received the B.Sc. degree from Isfahan University of Technology, Isfahan, Iran, in 2009 and the M.Sc. and Ph.D. degrees (with honors) in electrical engineering from Amirkabir University of Technology, Tehran, Iran, in 2011 and 2014, respectively. He is currently an Assistant Professor with the Department of Electrical and Computer Engineering, Jundi-Shapur University of Technology, Dezful, Iran. He also conducts research in the Research and Development Center, Damavand Induction Furnace Company, Damavand, Iran, where he is working on industrial induction heating systems. His current research interests include power electronics, induction heating, inductive power transfer, and renewable energy.