Compact wideband differential bandpass filter based on ... - IEEE Xplore

www.ietdl.org Published in IET Microwaves, Antennas & Propagation Received on 21st August 2011 Revised on 26th October 2011 doi: 10.1049/iet-map.2011.0400

ISSN 1751-8725

Compact wideband differential bandpass filter based on the double-sided parallel-strip line and transversal signal-interaction concepts W.J. Feng1 W.Q. Che1 T.F. Eibert 2 Q. Xue3 1

Department of Communication Engineering, Nanjing University of Science and Technology, 210094 Nanjing, People’s Republic of China 2 Lehrstuhl fu¨r Hochfrequenztechnik, Technische Universita¨t Mu¨nchen, Germany 3 State Key Laboratory of Millimeter Waves (Hong Kong), City University of Hong Kong, Hong Kong, People’s Republic of China E-mail: [email protected]

Abstract: Two compact wideband differential bandpass filters based on the double-sided parallel-strip line (DSPSL) and transversal signal-interaction concepts are proposed in this study. Two transmission zeros and good harmonic suppression can be achieved for the first wideband differential DSPSL filter, because of the two transmission paths with different lengths for input/output ports. In addition, another improved common suppression wideband differential DSPSL filter with two 1808 phase inverters is introduced. Four l/4 shorted lines are added to improve the passband transmission characteristic for the two differential DSPSL filters. Two prototypes with 3 dB fractional bandwidth of 94 and 117.6%, and return loss greater than 15 dB for differential mode are designed and fabricated. The measured results show good agreement with the theoretical expectations.

1

Introduction

The recent fast growth in new applications for modern wireless communication systems has increased the demand for microwave-balanced circuits. Differential mode filters with advantages of high immunity to environmental noise and good dynamic range are needed in the design of balanced circuits. For a well-designed differential modebalanced bandpass filter, only the desired differential mode frequency response is allowed to pass through the filter, whereas common mode signals should be suppressed. In addition, the balanced filter should possess excellent out-ofband rejection and high selectivity. In the past few years, several approaches have been demonstrated for single-band, dual-band and wideband differential filters [1–6]. In [6], a new microstrip line differential mode filter with good common mode suppression was designed. However, multi-section branch lines should be cascaded to achieve wideband performance, resulting in bigger size and unsatisfactory out-of-band common mode suppression. The double-sided parallel-strip line (DSPSL), as one kind of balanced transmission lines, is quite useful and convenient for the balanced microwave component designs. Compared with other balanced transmission lines, such as the microstrip and coplanar stripline, DSPSL has important advantages of easy realisation of low and high characteristic impedance and simple circuit structures of wideband transitions [7–10]. Using these advantages, a novel double186 & The Institution of Engineering and Technology 2012

sided parallel-strip line (DSPSL) differential ultra-wideband (UWB) bandpass filter using 1808 DSPSL swap with good common mode suppression was illustrated [11]. Recently, some filter structures using transversal signalinteraction concepts are proposed [12 – 17]. In these structures, the input signal is split into two subcomponents, which propagate through different feed-forward signal paths. By introducing intentionally a passband constructive interference and out-of-band signal energy cancellations to produce power transmission zeros, high-selectivity filtering responses and harmonic suppression can be achieved. However, few studies have described the application of transversal signal-interaction concepts in the differential broadband bandpass filters. In this study, we firstly propose two compact wideband differential DSPSL bandpass filters based on transversal signal-interaction concepts. On the basis of the two-port differential/common S-parameters formulas in [18], two different transmission paths are designed to realise the signal transmission from ports 1 and 1′ to ports 2 and 2′ for the differential DSPSL bandpass filters. Four l/4 shorted lines are introduced to improve the selectivity and the passband performance for the differential mode. Detailed theoretical design, simulation and experiment results for the two wideband differential filters are demonstrated and discussed. All the structures are simulated with Ansoft HFSS v.10.0 and constructed on a dielectric substrate with 1r ¼ 2.65 and h ¼ 0.5 mm. IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 2, pp. 186 –195 doi: 10.1049/iet-map.2011.0400

www.ietdl.org ⎡

2 Design of the differential filter with two differential transmission lines Figs. 1a and b show the three-dimensional view and top view of the first wideband differential DSPSL bandpass filter. The structure consists of two layers, and the shadow stands for the metallisation parts on different layers. The ideal equivalent circuit of the structure is shown in Fig. 1c. Two different transmission paths (with electrical lengths u, 3u, and characteristic impedance Z1 and Z2) are introduced to realise the signal transmission from ports 1 and 1′ to ports 2 and 2′ . In addition, four shorted lines with electrical length of u and characteristic impedance Zs are connected among the two transmission lines with different lengths independently. Four microstrip lines with characteristic impedance Z0 ¼ 50 V are connected to ports 1 and 1′ and ports 2 and 2′ , performing as input and output ports. The two-port differential/common mode S-parameters (Smm) can be extracted from the four-port S-parameters (Sstd) of Fig. 1 [18] ⎡

S mm

Sdd11 ⎢ Sdd21 =⎢ ⎣ Scd11 Scd21

Sdd12 Sdd22 Scd12 Scd22

Sdc11 Sdc21 Scc11 Scc21

⎤ Sdc12 Sdc22 ⎥ ⎥ = TSstd T −1 Scc12 ⎦ Scc21

(1)

⎤ ⎡ 1 −1 0 0 S11 ⎢ ⎥ ⎢ 1 ⎢ 0 0 1 −1 ⎥ ⎢ S1′ 1 T = √ ⎢ ⎥, Sstd = ⎢ ⎣ ⎦ ⎣ S21 2 1 1 0 0 0 0 1 1 S2′ 1

⎤ S11′ S12 S12′ S1′ 1′ S1′ 2 S1′ 2′ ⎥ ⎥ ⎥ S21′ S22 S22′ ⎦ S2′ 1′ S2′ 2 S2′ 2′ (2)

The differential/common mode S-parameters can thus be obtained as Sdd21 = (S21 − S2′ 1 − S21′ + S2′ 1′ )/2

(3)

Sdd11 = (S11 − S1′ 1 − S11′ + S1′ 1′ )/2

(4)

Scc21 = (S21 + S2′ 1 + S21′ + S2′ 1′ )/2

(5)

Scc11 = (S11 + S1′ 1 + S11′ + S1′ 1′ )/2

(6)

Next, the theoretical analysis of differential mode and common mode will be given below, aiming to determine the corresponding parameters and dimensions of the resulting filter.

Fig. 1 Different views of proposed differential bandpass filter a Three-dimensional view of proposed differential bandpass filter with two different transmission lines b Top view of proposed differential filter c Ideal equivalent circuit of proposed differential filter IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 2, pp. 186– 195 doi: 10.1049/iet-map.2011.0400

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www.ietdl.org 2.1

Differential mode analysis

On the basis of the transversal signal-interaction concepts, a passband can be achieved by the following relationship [19]

u12 (f0 ) = u1′ 2 (f0 ) + 2np (n = 0, 1, 2, . . .)

(7)

When the differential mode signals are excited to transmit from ports 1 and 1′ to ports 2 and 2′ , because of the two different transmission paths with electrical lengths of u and 3u (u ¼ 908 at the centre frequency f0), u12 ( f0) ¼ 908, u1′ 2( f0) ¼ 4508, a passband performance for the differential mode can thus be easily realised from ports 1 and 1′ to ports 2 and 2′ . The ABCD matrix of the shorted lines and the two transmission lines of different lengths are

cos u Ms = , M1 = 1/jZs tan u 1 jY1 sin u

jZ2 sin 3u cos 3u M2 = jY2 sin 3u cos 3u 1

0

jZ1 sin u cos u

(8)

From port 1 to port 2, the ABCD parameter matrix is Ms × M1 × Ms; and for port 1′ to port 2, the ABCD parameter matrix is Ms × M2 × Ms . In Fig. 1, the two

transmission paths from ports 1 and 1′ to port 2 (ports 1 and 1′ to port 2′ ) are connected in parallel similar to a two-port network, so the overall admittance matrix for Sdd21 in (3) can be calculated by summing their individual admittances. After the Y-, ABCD-, and S-parameter conversions, when Sdd21 ¼ 0, we can obtain the following relationship −1/(jZ1 sin u) + 1/(jZ2 sin 3u) = 0

(9)

The locations of the transmission zeros can thus be determined as follows utz1 = arcsin (3Z2 − Z1 )/4Z2 , utz2 = p − arcsin (3Z2 − Z1 )/4Z2

(10)

Fig. 2a shows three different frequency responses of the transmission zeros for the differential mode (simulated with Ansoft Designer v3.0). Table 1 shows the relationship between the transmission zeros and characteristic impedance Z1 and Z2 . In addition, the transmission zeros for the differential mode do not change with the characteristic impedance Zs of the shorted line (as shown in Fig. 2b). In this way, the 3 dB bandwidth for the differential mode of Fig. 1 can be adjusted easily by changing the characteristic impedance Z1 and Z2 of the two different transmission paths.

Fig. 2 Simulated frequency responses of the differential mode a |Sdd21| (Zs ¼ 80 V) b |Sdd21| (Z1 ¼ 110 V, Z2 ¼ 91.6 V) c |Sdd11| 188 & The Institution of Engineering and Technology 2012

IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 2, pp. 186 –195 doi: 10.1049/iet-map.2011.0400

www.ietdl.org Table 1 Z1 ¼ Z2 Z1 . Z2 Z1 , Z2

Transmission zeros for different cases in (10)

utz1 ¼ 458, utz2 ¼ 1358 utz1 , 458, utz2 . 1358 utz1 . 458, utz2 , 1358

ftz1/f0 ¼ 0.5, ftz2/f0 ¼ 1.5 ftz1/f0 , 0.5, ftz2/f0 . 1.5 ftz1/f0 . 0.5, ftz2/f0 , 1.5

The transmission zero appearing at 2f0 is created by the l/4 shorted lines. To satisfy (10) and realise wider bandwidth for the differential mode, Z1 , 3Z2 should be satisfied. In Fig. 1, from port 1 to port 1′ , there are also two transmission paths, the ABCD matrix for the upper path is Ms × M1 × Ms × M2 × Ms; and the one for the lower path is Ms × M2 × Ms × M1 × Ms . After matrix conversions, when Sdd11 ¼ 0, we can obtain the following relationship sin u × cos u × [16(m + n + p) sin4 u − 8(m + 2n + 3p) sin u + m + 3n + 9p] = 0 2

m = Z12 Zs2 + Z12 Z2 Zs ,

(11)

n = Z1 Z2 Zs2 + Z1 Z22 Zs + Z22 Zs2

p = Z22 Zs2 + Z1 Z22 Zs

(12)

Fig. 2c illustrates the simulated results of the transmission poles for the differential mode in three different cases. In addition, the number of the transmission poles for (11) is illustrated in Table 2. Obviously, the introduction of the four shorted lines with Zs can result in several transmission poles and thus help to improve the passband transmission characteristic for the differential mode (as shown in Fig. 2b, Z1Z2 ¼ Z0 ¼ 50 V). In addition, to obtain the desired better specifications for the differential mode, the loaded quality factor QL for the differential mode against characteristic impedance Z1 , Z2 and Zs are shown in Fig. 3. The loaded quality factor QL and the 3 dB bandwidth Df for the differential mode can be related by [20] QL = f0 /Df

(13)

Further the loaded quality factor QL can be expressed as QL ¼ f (Z1 , Z2 , Zs), once the centre frequency f0 of the differential mode is determined. We can adjust the characteristic impedance Z1 , Z2 and Zs to satisfy the demand of QL . The required 3 dB bandwidth Df and the transmission characteristic for the differential mode can be simultaneously obtained based on above-mentioned discussion.

Fig. 3 Simulated QL for the differential mode against Z1 , Z2 and Zs (Z2 ¼ 80 V)

the two transmission paths with different electrical lengths of u and 3u (u ¼ 908 at the centre frequency f0), u12 ( f0) ¼ 908, u1′ 2( f0) ¼ 2708, a stopband performance for the common mode can be easily realised from ports 1 and 1′ to ports 2 and 2′ . Using the similar analysis method for the differential mode, when Scc21 ¼ 0, the transmission zeros for the common mode filter can be achieved as below utz1 = arcsin (3Z2 + Z1 )/4Z2 utz2 = p − arcsin (3Z2 + Z1 )/4Z2

The simulated frequency responses of the common mode are shown in Fig. 4a. When Z1 ¼ Z2 , utz1 ¼ utz2 ¼ 908, there is only one transmission zero in the passband ftz1/f0 ¼ ftz2/ f0 ¼ 1; when Z1 , Z2 , there are two transmission zeros in the passband, utz1 , 908, utz2 . 908, ftz1/f0 , 1, ftz2/f0 . 1; when Z1 . Z2 , there is no transmission zero in the stopband for the common mode. In addition, from (6), when Scc11 ¼ 0, the transmission poles for the common mode closed to the stopband are

utp1

utp2 2.2

Common mode analysis

From the transversal signal-interaction concepts, a stop band can be achieved by the following relationship [19]

u12 (f0 ) = u1′ 2 (f0 ) + np (n = 1, 3, 5, . . . )

(14)

In this case, when the common mode signals are excited to transmit from ports 1 and 1′ to ports 2 and 2′ , because of Table 2 Zs ¼ 0 Zs = 0 Zs = 0

(15)

(2Z2 − Z1 ) = arcsin Zo /Zs 2Z2 (2Z2 − Z1 ) = p − arcsin Zo /Zs 2Z2

(16)

Two passbands are realised by the two transmission poles for the common mode (as shown in Fig. 4b), the 3 dB bandwidth of the two passbands for the common mode becomes narrower with the decrease of the characteristic impedance Zs . In Section 3, to improve the common suppression, another new wideband differential filter structure using two ideal 1808 phase inverters will be introduced.

Numbers of transmission poles in (11) – Z1Z2 ¼ Z0 Z1Z2 ¼ Z0

m¼n¼p¼0 m = 0, n = 0, p = 0 m = 0, n = 0, p = 0

IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 2, pp. 186– 195 doi: 10.1049/iet-map.2011.0400

utp ¼ 908 utp1 , utp2 ¼ 908 , utp3 utp1 , utp2 , 908 , utp3 , utp4

ftp ¼ f0 ftp1 , ftp2 , ftp3 ftp1 , ftp2 , ftp3 , ftp4

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Fig. 4 Simulated frequency responses of the common mode a Zs ¼ 75 V b Z1 ¼ 110 V, Z2 ¼ 91.6 V

2.3 Proposed differential bandpass filter with two different transmission lines On the basis of the above-discussed theoretical analysis, the design procedures of wideband differential bandpass filter are summarised as follows: 1. On the basis of the transversal signal-interaction concepts, calculate the electrical lengths of the two transmission paths and the four shorted stubs for the desired centre frequency f0 of the differential bandpass filter. 2. Determine the characteristic impedance Z1 , Z2 of the two transmission paths corresponding to the 3 dB bandwidth for the differential/common mode. To realise wider 3 dB bandwidth for the differential mode, Z1 . Z2 could be chosen. 3. Adjust the characteristic impedance Zs of four shorted stubs to realise three/four transmission zeros for the differential mode. 4. Calculate the parameters of the DSPSL transmission line according to the formulas in [21], and optimise the whole transmission characteristics of the wideband differential filter.

The final parameters for the circuit of Fig. 1c are Zs ¼ 80 V, Z1 ¼ 120 V, Z2 ¼ 85.7 V, Z1Z2 ¼ Z0 ¼ 50 V, and f0 ¼ 6.85 GHz. The optimised structure parameters for the first wideband differential DSPSL bandpass filter in Fig. 1 are W0 ¼ 1.85 mm, W1 ¼ 0.72 mm, W2 ¼ 0.92 mm, W3 ¼ 0.53 mm, l1 ¼ 3.1 mm, l2 ¼ 3.63 mm, m1 ¼ 0.68 mm, m2 ¼ 2.77 mm, m3 ¼ 5.32 mm, m4 ¼ 4.69 mm, m5 ¼ 3.01 mm, m6 ¼ 1.6 mm, n1 ¼ 0.5 mm, n2 ¼ 3.32 mm, n3 ¼ 3.3 mm, t1 ¼ 0.98 mm, t2 ¼ 1.8 mm, t3 ¼ 0.84 mm, t4 ¼ 3.2 mm, S1 ¼ 1.41 mm, S2 ¼ 1.69 mm, S3 ¼ 0.99 mm, d ¼ 0.54 mm. The simulated results for the wideband differential filter are shown in Fig. 5. For the differential mode, two transmission zeros are located at 2.9 and 10.7 GHz, respectively, whereas three transmission poles are realised in the passband (3 dB fractional bandwidth is approximate 94.5%). The insertion loss is less than 0.7 dB whereas the return loss is over 15.5 dB from 3.9 to 9.6 GHz. Furthermore, over 14.5 dB roll-off skirt rejection is achieved from 11 to 15.5 GHz. For the common mode, the stopband is from 4.3 to 9.8 GHz, the insertion loss is greater than 17 dB at the centre frequency of the stopband.

Fig. 5 Measured and simulated results for the proposed differential filter with two different transmission lines a Differential mode b Common mode 190 & The Institution of Engineering and Technology 2012


www.ietdl.org 3 Design of the differential filter with 18088 phase inverter Although the first proposed wideband DSPSL filter in Fig. 1 has better skirt selectivity, and improved second-harmonic suppression compared with former differential UWB structures in [6, 11], the common mode out-of-band suppression and bandwidth should be further improved. Fig. 6 shows the structure and the ideal equivalent circuit of another new compact wideband differential DSPSL bandpass filter with two 1808 phase inverters. From port 1 to port 2 (port 1′ to port 2′ ), the transmission path electrical length is 2u with characteristic impedance Z1 , and from port 1′ to port 2 (port 1 to port 2′ ) the transmission path electrical length is also 2u with a 1808 phase inverter at the centre of the transmission line (characteristic impedance Z2). The other parts are similar with the structure and circuits in Fig. 1. Next, similar theoretical analysis of differential mode and common mode with those in Section 2 will be given.

3.1

Differential mode analysis

From (7), when the differential mode signals are excited to transmit from ports 1 and 1′ to ports 2 and 2′ , because of the two different transmission paths with electrical lengths of 2u (u ¼ 908 at the centre frequency f0) and a 1808 phase inverter, u12 ( f0) ¼ 1808, u1′ 2( f0) ¼ 5408, a passband performance for the differential mode can thus be realised from ports 1 and 1′ to ports 2 and 2′ . For the two transmission paths of the circuit in Fig. 6, from port 1 to port 2, the ABCD matrix is Ms × M1 × Ms; and for ports 1′ to 2, the ABCD matrix is Ms × M2 × Mps × M2 × Ms , and Ms =

1 0 , 1/jZs tan u 1

M ps =

cos 2u jZ1 sin 2u , M1 = jY1 sin 2u cos 2u

−1 0

M2 =

0

−1 cos u jY2 sin u

jZ2 sin u

cos u (17)

Fig. 6 Structure and the ideal equivalent circuit of new compact wideband differential DSPSL bandpass filter with two 1808 phase inverters a b c d

Three-dimensional view of the proposed differential bandpass filter with two 1808 phase inverters Top view of the filter 1808 phase inverter Ideal equivalent circuit of the proposed differential filter


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www.ietdl.org The overall admittance matrix for Sdd21 in (3) for the differential filter with 1808 phase inverter can be also calculated using the method in Section 2. In addition, M1 ¼ 2M2 × Mps × M2 if Z1 ¼ Z2 , so after the Y-parameter conversions, we can obtain Ydd21 ¼ 2[1/(jZ1sin 2u) + 1/(jZ2sin 2u)], so |Ydd21| . 0, and there are no transmission zeros created by the two transmission paths for the differential mode [22]. The two transmission zeros appearing at DC and 2f0 are created by the l/4 shorted lines. Fig. 7a shows three different frequency responses of the differential mode, and the bandwidth for the differential mode is decreased with the increase of Z1 and Z2 . In addition, in Fig. 7, from port 1 to port 1′ , there are also two transmission paths, the ABCD matrix for upper path is Ms × M1 × Ms × M2 × Mps × M2 × Ms , whereas the one for down path is Ms × M2 × Mps × M2 × Ms × M1 × Ms . After the Y-, ABCD-, and S-parameter conversions, when Sdd11 ¼ 0, we can obtain the following relationship sin u × cos u × (2p cos2 u − q) = 0 p = Z1 Z2 + Z1 Zs + Z2 Zs ,

q = Z1 Zs + Z2 Zs

(18) (19)

Fig. 7b illustrates the simulated results of the transmission poles for the differential mode in two different cases. When Zs ¼ 0, q ¼ 0, there is only one transmission pole in the passband, and the frequency of the transmission pole ftp/f0 ¼ 1, utp ¼ 908; when Zs = 0, there are three transmission poles, and ftp1/f0 , ftp2/f0 ¼ 1 , ftp3/f0 , utp1 , utp2 ¼ 908 , utp3 and

inverters, u12( f0) ¼ 1808, u1′ 2( f0) ¼ 3608, a stopband performance for the common mode can be easily realised from ports 1 and 1′ to ports 2 and 2′ . In addition, owing to the ideal 1808 phase difference of the two 1808 phase inverters as discussed in [15], the common mode can be easily suppressed in the passband and out-of-band for the differential mode, so |Scc21| ¼ 0. Furthermore, because of the symmetry of the circuit in Fig. 6d, from (6), we have |Scc11| ¼ 1 (as shown in Fig. 8). 3.3 Proposed differential bandpass filter with two 1808 phase inverters On the basis of the theoretical analysis above, and the design produces in Section 2, the resulting parameters for the circuit of Fig. 6d are Zs ¼ 80 V, Z1 ¼ Z2 ¼ 120 V and f0 ¼ 6.85 GHz. The optimised structure parameters for the UWB differential DSPSL bandpass filter in Fig. 6 are W0 ¼ 1.85 mm, W1 ¼ 1 mm, W2 ¼ 0.53 mm, l1 ¼ 13.24 mm, l2 ¼ 5.83 mm, n1 ¼ 4 mm, n2 ¼ 2.7 mm, S ¼ 1.84 mm, g ¼ 0.25 mm, d1 ¼ 0.32 mm and d2 ¼ 0.6 mm. The simulated results for the differential UWB filter is shown in Fig. 9. For the differential mode, three transmission poles are realised in the passband (3 dB fractional bandwidth is

utz1 = arccos (Z1 Zs + Z2 Zs )/(2Z1 Z2 + 2Z1 Zs + 2Z2 Zs ) utz3 = p − arccos (Z1 Zs + Z2 Zs )/(2Z1 Z2 + 2Z1 Zs + 2Z2 Zs ) (20) Similarly, as illustrated in Section 2, the introduction of the four shorted lines with Zs can result in several transmission poles and thus help to improve the passband transmission characteristic for the differential mode. 3.2

Common mode analysis

From (14), when the common mode signals are excited to transmit from ports 1 and 1′ to ports 2 and 2′ , because of the two transmission paths and the two 1808 phase

Fig. 8 Simulated frequency responses of the common mode (Zs ¼ 75 V)

Fig. 7 Simulated frequency responses of the differential mode a Zs ¼ 75 V b Z1 ¼ Z2 ¼ 120 V 192 & The Institution of Engineering and Technology 2012


www.ietdl.org approximate 118%). The simulated insertion loss is less than 0.9 dB whereas the return loss is over 15 dB from 3.5 to 10.6 GHz. For the common mode, the simulated insertion loss is greater than 15.5 dB in the whole frequency band, having better common mode suppression compared with the first differential filter.

4

Measured results and discussions

Two prototypes of the proposed differential DSPSL UWB bandpass filters with the size of 27 × 27 mm and

27 × 27 mm are fabricated on the substrate with 1r ¼ 2.65, h ¼ 0.5 mm, and tan d ¼ 0.002. Fig. 10 illustrates the photos of the two wideband differential filters. For comparison with the theoretical expectations, the measured results are also illustrated in Figs. 5 and 9. From Fig. 5, we may note that, for the differential mode, two transmission zeros are located at 2.9 and 10.6 GHz (3 dB fractional bandwidth is approximate 94%), respectively. The measured insertion loss is less than 1.6 dB whereas the return loss is over 15 dB from 3.9 to 9.4 GHz. Furthermore, over 14 dB roll-off skirt rejection is achieved from 11 to 16 GHz. For

Fig. 9 Measured and simulated results for the proposed differential filter with two1808 phase inverters a Differential mode b Common mode

Fig. 10 Photograph of two proposed wideband differential filter a Filter with two different transmission lines b Filter with two 1808 phase inverters IET Microw. Antennas Propag., 2012, Vol. 6, Iss. 2, pp. 186– 195 doi: 10.1049/iet-map.2011.0400

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www.ietdl.org Table 3

Comparisons of measured results for several differential filter structures

Differential filter structures

centre frequency (f0 , GHz) effective circuit size (l0) fractional bandwidth, % transmission zeros (|Sdd21|) |Sdd11|, dB |Scc21|, dB harmonic suppression for |Sdd21|

Filter with two different transmission lines

Filter with two 1808 swaps

Ref. [6]

Ref. [11]

6.85 0.57l0 × 0.4 . l0 94 p

6.85 0.63l0 × 0.63l0 117.6 × .15.5 .15 p

4.0 0.70l0 × 0.53l0 65 × .20 .20 passband ×

3.0 0.75l0 × 0.71l0 110 × .15 .20 p

.15 .17 at f0 p

the common mode, the measured stopband is from 4.1 to 9.7 GHz, the measured insertion loss is greater than 17 dB at the centre frequency of the stopband. From Fig. 9, for the differential mode, the 3 dB fractional bandwidth is about 117.6%. The measured insertion loss is less than 1.75 dB, whereas the return loss is over 15.5 dB from 3.5 to 10.5 GHz. For the common mode, the measured insertion loss is greater than 15 dB in the whole frequency band, indicating better common mode suppression compared with the first differential filter with two different transmission lines. Comparing the measured and simulated results, a slight frequency discrepancy and bigger insertion loss for the differential mode can be observed in the measured results. Some explanations are given below. In the practical prototypes of the proposed wideband differential filters, the shorting via holes were soldered with the DSPSL ground, the imperfect soldering skill actually affects the equivalent wavelength of the four shorted lines. In addition, the limited fabrication precision, especially for the folded transmission lines, also contributes a little error to the measurement results. Table 3 illustrates the comparisons of measured results for several different wideband differential structures. Compared with three other differential filters, for the first wideband differential filter with two different transmission lines, two transmission zeros in the vicinity of the passband and additional two transmission zeros in the second harmonic for the differential mode are realised, better selectivity and good harmonic suppression for the differential mode can thus be achieved. However, the out-of-band common mode suppression should be improved. For such purpose, another wideband differential filter with two 1808 phase inverters is proposed, common mode suppression can be improved because of the ideal 1808 phase difference of the phase inverters, and the filter size is almost the same as that of the first filter structure. Compared with the former wideband differential filters in [6, 11], the bandwidth for the differential mode of the proposed two wideband differential filters can be easily adjusted by changing the impedances for the two transmission paths, in addition, the transmission characteristics for the differential/common mode can be easily analysed by the matrix conversions based on the two-port differential/common S-parameter formulas in [18].

5

Conclusion

In this study, two new and compact wideband differential DSPSL bandpass filters based on transversal signalinteraction concepts are firstly proposed and designed. Two transmission paths are used to realise the signal transmission from ports 1 and 1′ to ports 2 and 2′ , 194


according to the two-port differential/common S-parameters formulas for the two differential filters. Good selectivity and harmonic suppression can be realised for the first wideband differential filter with two different transmission lines. Another new wideband differential filter with two 1808 phase inverters is introduced to improve the common mode suppression. The bandwidth for the differential mode of two differential filters can be easily adjusted by changing the impedances of the two transmission paths. Compared with former wideband differential filter structures, the proposed two filter structures have simpler filter structures and design theory. Good agreements between simulated and measured responses of the filter are demonstrated, indicating the validity of the design strategies.

6

Acknowledgments

This work is supported by the National Natural Science Foundation of China (60971013) and the Natural Science Foundation of Jiangsu Province (BK2008402). The authors thank the support from the IGSSE Project 5.02 of Technische Universita¨t Mu¨nchen (TUM), Germany.

7

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