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Design and Loss Comparison of Matrix Converters and Voltage-Source Converters for Modern AC Drives Steffen Bernet, Member, IEEE, Srinivas Ponnaluri, and Ralph Teichmann, Student Member, IEEE
Abstract—This paper compares a matrix converter (MC) and a dc-voltage link converter with an active front end for a 7.5-kW, 460-V induction motor drive. Part count, semiconductor losses, input filter design, and protection aspects are discussed. It is shown that the matrix converter’s semiconductor losses are smaller only at full load operation for the same silicon area in both converters. A 33% reduction of the device current rating of the MC is possible, resulting in comparable thermal device stress. The overall passive component count and rating is only slightly better for the MC.
schematics of the VSC- and the MC-based drives are depicted in Figs. 1 and 2, respectively. The comparison is done for a 460-V, 7.5-kW induction motor drive, a power range that is representative of the highly competitive drives market for industrial and consumer goods.
Index Terms—AC motor drives, input filter design, matrix converter, semiconductor losses, voltage-source converter.
A. Mains
I. INTRODUCTION
I
N RECENT years, the hard switching three-phase to threephase matrix converter (MC) has received considerable attention as an alternative to the dc-voltage-link converter with active front-end. Numerous publications have dealt with modulation schemes [1]–[3], operation at unbalanced input voltages [4], [5], semiconductor device and packaging technology [6], [7], gate drive concepts [8], and the commutation procedure for bi-directional switches (BDSs) [9]. MC technology has matured and is considered for a variety of industrial applications where a substantial amount of energy can be fed back to the mains, such as in rolling mills, conveyor belts, and elevators. The attractiveness of MC technology is increasing due to several factors, including the rapid decline in semiconductor costs, the advent of novel packaging concepts, and improvements in both on-state and switching characteristics of the semiconductors. For example, an insulated gate bipolar transistor (IGBT) module containing nine forward and reverse blocking BDSs using state-of-the-art trench gate technology was presented in [7]. The objective of this paper is to discuss key design criteria for a system comparison conducted to evaluate the market potential of the MC technology. In this comparison, the MC topology is set against a pulsewidth modulation (PWM) rectifier/dc-voltage-link/PWM inverter structure [subsequently referred to as a voltage-source converter (VSC)]. The Manuscript received April 14, 2001; revised August 24, 2001. Abstract published on the Internet January 9, 2002. S. Bernet is with the Department of Electrical Engineering and Computer Science, Technical University of Berlin, 10587 Berlin, Germany (e-mail:
[email protected]). S. Ponnaluri is with ABB Corporate Research, 68526 Ladenburg, Germany. R. Teichmann is with the Department of Electrical Engineering, Technical University of Dresden, 01062 Dresden, Germany. Publisher Item Identifier S 0278-0046(02)02892-7.
II. EXAMPLE DRIVES SPECIFICATIONS
Both drives are designed to run on a balanced, three-phase 460-V, 60-Hz grid voltage . The converters will operate at unity displacement power factor at the grid terminals. The drives are to meet the stringent EMC standards of EN55014 (class B) typically required for domestic appliances. For convenience, the line impedance stabilization network (LISN) and the voltage limits set forth in the standard are replicated in Fig. 3 and Table I, respectively. B. Converter Specifications Both converters are designed to deliver a short-time overload of three times the rated current current . This corresponds to a converter power of 28.6 kVA. 10 kHz. For The target switching frequency was set to comparison purposes, the losses have also been calculated at 20 kHz. Both converters feature a dynamic control reserve of 5%. Incorporating the voltage drop across the input filter will 750 V. A rated rms set the dc-link voltage for the VSC to 380 V leaves a 5% control reserve for output voltage of the MC, assuming it is connected to a 460-V mains. Further data are shown in Table II. C. Induction Machine Two different types of induction machines are to be fitted to the MC and the VSC based drives. A standard 7.5-kW, 460-V induction machine can be used for the VSC drive. Since the maxof the MC for linear continuous imum output voltage modulation schemes is limited to 86.6% of the input voltage [1], an induction machine with lower rated voltage must be chosen. A special 7.5-kW, 380-V induction machine design which features the same per unit parameter as the , , , 460-V motor ( ) [10] is assumed. The motor terminal data of a standard NEMA class B induction machine are summarized in Table II.
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Fig. 1. Induction motor drive with VSC.
converters must meet EN55014 and feature an equal mains current ripple. 1) VSC: An LCL filter structure consisting of , , and was chosen (Fig. 1). For a given attenuation, it can be shown is retheoretically that the minimum filter capacitor value and the converter side inquired if the grid-side inductance are equal. However, to decrease the dc and ac side ductance filter capacitor current ratings and to suppress the generation of is to EMI on the source, the inductor on the converter side . be larger than the inductor on the grid side To ensure a reasonable magnetic design, it is assumed that shall not exthe peak ripple current through the inductor 2.2 A (15% of peak fundamental current ceed of 14.55 A). For no-load conditions, neglecting the inductor resistance, assuming a closed loop control and a rectifier input voltage fundamental equal to the mains voltage, the inductance is given by (1) [13]. With the data in Table II, the inductor value is found to be 2.5 mH Fig. 2. Induction motor drive with MC.
III. EXAMPLE DRIVE DESIGN A. Modulation Scheme To compare semiconductor losses and input filter design of both converters, two closely related continuous modulation schemes were selected. While the inverter and rectifier of the VSC use sine wave modulation with added third harmonics [11], a direct modulation was assumed for the MC [1]. Aside from the placement of the zero states, the former can directly be derived from the latter [12]. B. Input Filter Both converters require an input filter that shapes the input current such that it meets the desired standard. A variety of optimization criteria exist such as minimum cost, losses, volume, and weight. Active damping is to be applied which requires a in the controllable range. In ,practice resonance frequency this is typically ensured if the resonant frequency is not higher than 20% of the switching frequency which was set to 10 kHz. For the comparative analysis, it is assumed that both
(1) , (2) is To determine the value of the grid-side inductor used. The iterative optimization process observing inductance and capacitance values, rms and peak current stresses, and res0.8. onant frequency resulted in an inductor split factor was set to 0.62 mH Thus, (2) For a substantial attenuation of the switching current ripple, at switching frequency should be at least the impedance of at this frequency. five times lower than the impedance of 12, the value For a current ripple attenuation factor of of the filter capacitor is 4.91 F (3) (3) leads to harmonic currents drawn Since a large value of from the potentially distorted mains, the filter capacitance value
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Fig. 3.
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 49, NO. 2, APRIL 2002
LCL filter and line impedance stabilization network recommended by EN55014.
TABLE I EN55014 CONDUCTED EMI LIMITS (CLASS B)
should be kept small. Thus, a compromise must be achieved to meet the EMC standard and keep the input switching current ripple small. The filter component values and their ratings are summarized in Table III. The current ratings were determined by simulation at rated and overload operation. up to 150 kHz were evaluated. All rectifier harmonics are assumed. In practice, the Ideal switches with infinite PWM voltage pulses will be trapezoidal and the overall harmonic spectrum will be better than that of the computed values. of Fig. 3 is For a negligible line impedance, the voltage depicted in Fig. 8 for rated operation. The current waveforms on the converter and on the grid side are shown in Figs. 4 and 6, respectively. 2) MC: The MC acts like a current source converter if viewed from the supply side. Therefore, an LC filter is required. The converter side current at rated load is shown in Fig. 5. 0, the grid inductance is selected to be roughly With the same as the total inductance of the LCL filter of the VSC. The filter capacitor value is designed such are equal for the MC and the that the current ripples in VSC at rated operation. This yields filter component values 3.1 mH and 9.9 F. The filter capacitors of are subject to a high-frequency rms current of 10.8 A and 32.3 A for rated and overload operation, respectively. The component ratings are summarized in is shown in Fig. 7. Table III. The current in the inductor The EMC compatibility at rated operation is shown in Fig. 9. The compatibility is achieved throughout the entire operation range.
TABLE II SPECIFICATIONS OF EXAMPLE DRIVES
C. Design of DC-Link Components and Clamp Circuitry To protect the MC, a clamp circuit (shown in Fig. 2) is normally used. If the converter is shut down due to an overcurrent or a short-circuit current, the clamp circuit provides a current path to de-energize the load. With the converter being shut off at the hardware current limit, the minimum value of the capacitor in the clamp circuit can be calculated with (4) which was derived 68 A in [16]. The hardware current limit is set to by the peak current value at overload condition plus a margin of 10%. The maximum clamp capacitor voltage was chosen to
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(f
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TABLE III FILTER COMPONENT COMPARISON OF VSC AND MC 10 kHz, V 460 V/380 V, I 12 A/14.5 A, V 750 V, V 460 V, I 36 A/43.5 A)
=
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Fig. 6. Grid-side current i 4.91 F, V 750 V, V 10 kHz).
=
Fig. 4. Rectifier currents i , 0.62 mH, C 4.91 F, V 8.2 kW, and f 10 kHz).
= =
i
, and
i
= 750 V, V
of VSC (L 460 V,
=
= 2.5 mH, L = f = 60 Hz, P =
of VSC (L 460 V, f
=
Fig. 7. Grid-side current i of MC (L 460 V, V 380 V, I 14.55 A, f and f 10 kHz).
=
=
=
= 2.5 mH, L = 0.62 mH, C = = 60 Hz, P = 8.2 kW, and f =
= 3.1 mH, C = 9.9 F, V = = 60 Hz, f = 50 Hz, P = 8.2 kW,
to be 134.5 F formed by a series connection of two board-mounted 270- F, 450-V capacitors (4)
Fig. 5. Converter side current i V 460 V, I 14.55 A, f f 10 kHz).
=
=
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of MC (L = 3.1 mH, C = 9.9 F, = 60 Hz, f = 50 Hz, P = 8.2 kW, and
be 900 V, which allows the application of a series connection of two readily available 450-V electrolytic capacitors and provides a voltage margin of 300 V to the maximum device blocking voltages. The steady-state voltage in the clamp is given by the peak value of the line-to-line capacitor input voltage. For the induction machine parameters shown in the previous section, the minimum capacitor value was found
The design of the dc-link capacitor in the VSC depends on the desired dc-link voltage ripple, supply harmonics, ride-through capability, the control structure and the dynamics of the control hardware, and the rms current through the capacitor. To achieve minimum costs, electrolytic capacitors are usually applied. Aside from ride-through specifications, the minimum capacitance value is typically set by the rms current rating at the switching frequency harmonics in conjunction with temperature and lifetime specifications. The dc-link capacitor current was investigated using a simulation tool by sweeping the parameters in the speed–torque plane. The capacitor currents reached A at switching maximum rms values of roughly 40 kHz at several points in the frequency multiples up to speed–torque plane including overload conditions. value was determined to be 171 F for a The minimum maximum dc-voltage ripple of 5% using the method described in [13]. Using the capacitor design tool provided by [14], the nearest capacitor value that still meets the current requirements
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Fig. 8. Computed EMC spectrum and EN 55 014 Class B limit for VSC versus 2.5 mH, L 0.62 mH, C 4.91 F, V number of harmonic n (L 750 V, V 460 V, f 60 Hz, P 8.2 kW, and f 10 kHz).
=
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was found. A 1500- F, 450-V board-mounted capacitor type was selected with a useful lifetime of 8 years at an average ca60 C. Thus, the effective dc-link pacitor temperature of capacitance of the series connection of the two 1500- F capacitors is 750 F. D. Semiconductor Design Approximately, the same installed switch power and silicon area was assumed for both converter topologies. Therefore, despite the higher rated output current, the current ratings of the switches in the MC were reduced by about 30% compared to the switches in the VSC. The current utilization of the semiconductors in the VSC was chosen to match the current utilization of equivalent industrial drives. The data of the EUPEC NPT-IGBTs BSM50GB120DN (1200 V, 50 A) and BSM35GB120DN (1200 V, 35 A) were chosen for the VSC and the MC, respectively. Both are modules containing IGBTs and diodes of the same structure and technology. Since the safe operating area (SOA) of the IGBTs and diodes enables the conduction and switching of two times the rated device current, the semiconductors in the MC can handle the maximum converter currents without problems. The semiconductor design must also incorporate thermal limits. The semiconductor losses of the converters were investigated as a function of the modulation depth, motor displacement power factor , switching frequency , and rms motor (output) current . The loss simulations validated the design of the MC with devices of smaller current rating. The average junction temperatures and the corresponding stress of each of the semiconductors in the MC is almost the same as in the VSC applying NPT-IGBTs with 30% increased current rating. The losses were simulated using a power converter loss model as described in [6] and [11]. The switching energies and the on-state voltages are obtained from the manufacturer’s data sheet and/or loss measurements. The switching energies are distributed to the switches based on the commutation type (natural or forced). The on-state losses are added whenever the device is conducting current. If not stated otherwise, the
Fig. 9. Computed EMC spectrum and EN 55 014 Class B limit for MC versus 3.1 mH, C 9.9 F, V 460 V, f number of harmonic n (L 60 Hz, f 50 Hz, P 8.2 kW, and f 10 kHz).
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semiconductor losses are calculated for a constant junction 125 C. The thermal time constant temperature of was assumed to be much larger than the period of the output frequency. Effects of thermal impedances at low output fre5 Hz) were not taken into consideration. The quencies ( following acronyms are used: PconT (IGBT on-state losses), PconD (diode on-state losses), PonT (IGBT turn-on losses), PoffT (IGBT turn-off losses), PoffD (diode turn-off losses), and Ptot (total semiconductor losses). 1) MC: A back-to-back configuration of two IGBTs was assumed. Due to the symmetry of the switches, both the common emitter and the common collector configuration [6] feature the same losses. Figs. 10 and 12 show the on-state, switching, and total converter losses of the MC as a function of the modulaand the output displacement power tion depth factor , respectively. All loss components are constant due to the symmetrical structure of the switches. Pulse dropping reduces the switching and total converter losses slightly at the maximum modulation depths. The IGBT and diode losses are not always equally distributed among the three switches comprising a switch group. Depending on the operation parameters and the modulation scheme, higher losses can occur in the diodes and IGBTs of one switch compared to the diode and IGBT losses of the other two switches. The losses as a function of the output rms current are depicted in Fig. 11. The total converter losses increase nearly linearly with increasing output current. Fig. 13 shows the losses of the MC as a function of the normalized speed and torque. Both speed and torque are and , respectively. If at conbased on their rated values stant torque the speed rises, the modulation index , the phase shift , and the real power increase at constant output current. Since the MC losses depend only on the output current at constant mains voltage, the losses remain constant at increasing speed and constant torque. The induction machine is symmetrical, field-oriented controlled, and the rotor flux is constant. 2) VSC: Fig. 14 shows the sum of the semiconductors losses in rectifier and inverter versus the inverter modulation . The modulation index of the index rectifier is almost constant. Slight adaptations to adjust unity
BERNET et al.: DESIGN AND LOSS COMPARISON OF MCs AND VOLTAGE-SOURCE CONVERTERS
Fig. 10. MC semiconductor losses as a function of the modulation depth q (BSM35GB120, V 460 V, I 25 A, 30 , f 10 kHz, and T 125 C).
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Fig. 11. MC semiconductor losses as a function of the rms output current (BSM35GB120, V 460 V, q 0.75, 30 , f 10 kHz, and T 125 C).
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power factor at the mains for different load conditions are negligible. On-state and switching losses increase with increasing inverter modulation index. On the one hand, the rectifier input current increases with increasing inverter output voltage and constant output current and output displacement power factor. On the other hand, the increasing conduction times of the IGBTs and the decreasing conduction times of the inverse diodes increase the total conduction losses of the inverter. Pulse dropping decreases the switching losses slightly at the maximum modulation index. The sum of the semiconductor losses in the rectifier and the inverter is displayed in Fig. 15 as a function of the output phase shift . The total semiconductor losses in the inverter vary slightly with the displacement power factor due to a variation of the conduction times of IGBTs and diodes and the higher on-state voltages of the IGBTs. The 180 where the inverter realizes the lowest total losses at diodes reach the maximum conduction time. The losses of the 90 and 270 where no real rectifier reach zero at power is transmitted. The rectifier conduction losses are higher 180 than at 0 . At 180 , energy is fed at
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Fig. 12. MC semiconductor losses as a function of the output phase angle (BSM35GB120, V 460 V, q 25 A, f 10 kHz, 0.75, I and T 125 C).
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Fig. 13. MC semiconductor losses versus normalized speed and torque (BSM35GB120DN2, V 460 V, P 7.5 kW, f 10 kHz, and T 125 C, constant rotor flux, stationary operation).
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back to the grid and the IGBTs conduction time predominates. The individual losses of the rectifier and inverter for the same system are shown in [15]. The losses of the rectifier plus inverter increase almost linearly as a function of the rms output current as shown in Fig. 16. Fig. 17 shows the simulated semiconductor losses of the VSC as a function of the speed and torque for a field-oriented controlled induction motor with constant stator flux. Both speed and torque are based on their rated values. If the speed rises at con, the phase shift , and stant torque the modulation index the real power increase at constant output current. The losses of the VSC clearly increase with increasing input current of the of the inverter. rectifier and increasing modulation index In contrast to the MC, the losses are equally distributed in steady-state operation among the six IGBTs and among the six diodes of the rectifier. The same is true for the inverter. IV. EXAMPLE DRIVE COMPARISON A. Switches, Gate Drives, Control The MC needs six additional IGBTs, diodes, and gate units compared to the VSC for the main switches. However, the same
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Fig. 14. VSC semiconductor losses as a function of the inverter modulation 460 V, V 750 V, I 25 A, index (BSM50GB120, V 30 , 0 ,f 10 kHz, and T 125 C).
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Fig. 16. VSC semiconductor losses as a function of the rms output current 460 V, V 750 V, q 0.75, 30 , 0 , (BSM50GB120, V f 10 kHz, and T 125 C).
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number of isolated power supplies is required in both systems. This assumes a common collector configuration for the MC and nonisolated power supplies for all devices attached to the negative dc-rail in the VSC. With the de-rating of the semiconductor devices for the MC, roughly the same switch power (i.e., chip area) is installed. Twelve additional diodes are required for the clamp circuit. The control of the staggered switching in the MC is more complex than a simple dead-time control in the VSC. If all circuit states are to be measured, the MC must be fitted with two input-current, two output-current, and two input voltage sensors. In contrast, the VSC must be fitted with four input-current, two output-current, and two input voltage sensors. An additional voltage sensor is required to measure the dc-link voltage in the VSC. The feedback of all circuit states enables a simple implementation of active damping. B. Filter Components The filter component values and the corresponding rms and peak-to-peak (pp) ripple current ratings are summarized in
=
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Fig. 17. VSC semiconductor losses versus normalized speed and torque 460 V, P 7.5 kW, f 10 kHz, and T (BSM50GB120, V 125 C, constant rotor flux, stationary operation).
=
Fig. 15. VSC semiconductor losses as a function of the output displacement 460 V, V 750 V, q 0.75, 0 , angle (BSM50GB120, V I 25 A, f 10 kHz, and T 125 C).
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Table III for rated operation (subscript ) and overload operation (subscript ). The MC requires only six filter components while the VSC system requires nine filter components. Since the of the VSC, a special iron powder current ripple is high in can be a core is needed for these inductors. The inductor standard three-phase inductor in both systems. The filter capacitor value of the MC is twice that of the VSC and its current rating is higher by a factor of 12 and 36 for rated and overload operation, respectively. Its high-frequency rms current rating is of the same magnitude as the rms input current. C. DC-Link and Clamp Components Comparing the ratings of the electrolytic dc-link capacitor of the VSC and the clamp capacitor of the MC, it can clearly be seen that the dc-link capacitor value of the VSC is roughly six times that of the clamp capacitor value of the MC. In both systems, the gate units can be fed from this dc-link. The clamp capacitor in the MC is subject to a very small rms current. The capacitance value is set by the energy stored in the leakage inductance of the electric motor, the trip current, and the voltage
BERNET et al.: DESIGN AND LOSS COMPARISON OF MCs AND VOLTAGE-SOURCE CONVERTERS
Fig. 18. Percent loss difference of a VSC and an MC versus speed and torque 10 kHz, P 7.5 kW, (MC: BSM35GB120, VSC: BSM50GB120, f V 460 V, and T 125 C).
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Fig. 19. Percent loss difference of a VSC and an MC versus speed and torque (MC: BSM35GB120, VSC: BSM50GB120, P 7.5 kW, V 460 V, f 20 kHz, and T 125 C).
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margin. In contrast, the dc-link capacitor value of the VSC is mainly set by current stress and life-time considerations. D. Semiconductor Losses The losses of both systems were discussed for different operation conditions. The MC clearly features fewer switching losses. This is due to only nine switching transitions per switching cycle in comparison to twelve for the VSC for a continuous modulation scheme. Additionally, the average commutation voltage of the MC is roughly 63% of that of the VSC. The average commutation voltages are given by the average absolute line-to-line voltage and the dc-link voltage for the MC and VSC, respectively. One important criterion is the semiconductor loss performance of the MC versus the VSC in an actual drive application. Figs. 18 and 19 show the semiconductor loss difference defined in (5) in a speed–torque plane for a field-oriented controlled induction machine for a switching frequency of 10 kHz and 20 kHz, respectively. The semiconductor loss
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Fig. 20. Percent loss difference of a VSC and an MC versus speed and torque 10 kHz, T 125 C, (MC: BSM50GB120, VSC: BSM50GB120, f P 7.5 kW, and V 460 V).
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Fig. 21. Percent loss difference of a VSC and an MC versus speed and torque (MC: BSM50GB120, VSC: BSM50GB120, f 20 kHz, T 125 C, P 7.5 kW, and V 460 V).
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differences are normalized to the semiconductor losses of the VSC at rated operation (5) The losses in the MC and in the VSC increase at constant speed and rising torque especially due to the increase of the output current. In addition, the input current at the rectifier of the VSC increases due to the higher real power being transferred. Although the MC uses IGBTs with about 33% lower current ratings and the rated current of the induction machine driven by the MC is about 21% higher, the total losses in the MC are about 11% less than those in the VSC 10 kHz) at rated speed and torque (Fig. 18). For ( 20 kHz, the total losses in a switching frequency of the MC are reduced by 20% at rated speed and torque in comparison to the VSC (Fig. 19). However, at partial load, in particular at low speed and full torque, the losses in the VSC are substantially lower due to the
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Fig. 22. Junction temperatures of the MC (equal loss distribution) and VSC for rated and overload operation (MC: BSM35GB120, VSC: BSM50GB120, V 460 V, 25 , V 750 V, T 80 C, T 125 C, f 10 kHz, V 380 V/460 V, and I 43.5 A/36 A).
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low input current drawn by the rectifier of the VSC. In contrast to the VSC, the switches of the MC have to carry and to switch the load current independent of the converted real power. Obviously this is a drawback of the MC at high reactive power of the load. Figs. 20 and 21 show the percent loss difference between the MC and VSC if both converter systems apply the same (1200 V, 50 A)-IGBTs. The MC generates fewer losses than the 10 kHz VSC in about 50% of the torque/speed range at (Fig. 21). The loss savings of the MC reach a maximum of 22% at rated torque and speed. Applying a switching frequency of 20 kHz, the losses of the MC are lower than those of the VSC in nearly the entire torque/speed range (Fig. 20). The total losses are reduced by 32% at rated output power. The maximum achievable switching frequency is strongly dependent on the thermal characteristics of the modules. For the MC, the distribution of the losses among the switches will additionally affect the maximum switching frequency. The distribution of the losses is determined by the ratio of input to output frequency, the modulation depths, the displacement power factor, and the modulation scheme. For most operating conditions, an equal loss distribution is achieved. Only in certain operating points, like zero speed or identical input and output frequencies, are the losses not equally distributed. This is due to the fact that some switches are more often activated during the peak values of the input voltage/output current while others are gated on only during small values of input voltage/output current. For equal loss distribution and overcurrent condition , at a constant heat sink temperature 80 C, a maximum junction temperature 125 C, and the modules being discussed, the switching frequency is limited to 20.6 kHz and 42.0 kHz for the VSC and the MC, respectively. If (1200-V, 50-A) IGBTs are applied in the MC, its maximum switching frequency is about 70 kHz for the same conditions.
Fig. 23. Composition of MC (equal loss distribution) and VSC losses at different switching frequencies and overload conditions (MC: BSM35GB120, 460 V, 25 , V 380 V/460 V, VSC: BSM50GB120, V I 43.5 A/36 A, and T 125 C).
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For an unequal loss distribution caused by an identical input and output frequency or an output frequency of zero, the maximum switching frequency of the MC reduces to values between kHz depending on the setting of the output displacement power factor and the modulation depths. Fig. 22 displays the junction temperature of the diode and IGBT chips in the VSC and the MC at rated and overload operation for a 10 kHz, a constant heatsink temswitching frequency of 80 C, and equal semiconductor loss distribution perature in the MC. The thermal stress of the devices in both converters are very similar. The losses of the MC and the VSC at overload conditions are depicted in Fig. 23 for various switching frequencies. While the losses of the MC and the VSC are about 5 kHz, the MC generates 25% fewer losses at the same at 20 kHz. V. SUMMARY The MC requires 50% more semiconductors and gate drives, excluding the clamp circuit. In spite of that, the active silicon area and the number of gate unit power supplies are comparable to that of the VSC of the same power rating. Fewer passive components are needed for the MC. The total input inductance value is comparable in both converters. While the complete MC input inductance is subject to a low current ripple stress, the input inductance of the VSC is split into two inductances (ratio 20%/80%) with the larger one subject to a high current ripple of 15% of the peak fundamental current. The input filter capacitors of the MC are double in capacitance value and subject to a high-frequency current stress of the same magnitude as the fundamental rms input current. In contrast to that, the current stress of the input filter capacitors of the VSC is basically independent of the converter power and smaller by a factor of 12 at rated current. The dc-link capacitor value of the VSC (750 F/900 V) is about six times larger than the clamp capacitor of the MC (135 F/900 V).
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TABLE IV COMPARISON
OF AN MC AND A VSC FOR A 7.5-kW FIELD-ORIENTED CONTROLLED INDUCTION MACHINE
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tion machine at given electrical mains, the MC requires motors which realize full speed and torque at 86.6% of the mains voltage. The control of the MC is certainly more complex. This is mainly due to the higher number of controllable switches with the demand for staggered switching. The higher number of gate units will have a negative impact on the overall failure in time (FIT) rate of the converter. However, with the advent of single modules containing a complete set of main switches of an MC, a useful converter FIT rate should be possible. Based on the characteristics discussed above, the MC is obviously advantageous if a small size/weight and/or a high switching frequency is required. It realizes lower losses than the VSC in applications where high real power must be converted from ac to ac. The limited output voltage of the MC makes it a potential topology for integrated motor-converter applications. REFERENCES
The MC generates 11%/20% fewer losses at 10 kHz/20 kHz than the VSC at rated torque and speed with the same chip area installed. Applying devices with the same current rating in the VSC and the MC, which translates to 50% more silicon area in the MC, the semiconductor losses of the MC are reduced by 22%/32% (10 kHz/20 kHz). The losses of both converter systems are roughly the same at the typical 70% of rated torque and speed and a operating range of 40 10 kHz. Especially at full torque switching frequency of and speeds less than 40% of the rated speed, the semiconductor losses of the VSC are smaller. At higher switching frequencies, the losses of the MC become smaller than that of a VSC in a wider operation range. The advantage of the MC in areas 20 kHz) is due to fewer of high switching frequencies ( commutations and an average commutation voltage of 63% of that of the VSC. The typically higher possible switching frequency of the MC is limited by an unequal loss distribution among the nine switches in certain operating conditions where a few semiconductors are loaded with the majority of losses. Table IV summarizes important features of the MC- and VSC-based drive systems. In contrast to the VSC, the MC cannot be operated at singlephase mains. To get the full mechanical power of an induc-
[1] M. G. B. Venturini and A. Alesina, “Intrinsic amplitude limits and optimum design of 9-switches direct PWM-AC–AC Converters,” in Proc. IEEE PESC’88, 1988, pp. 1284–1291. [2] L. Huber and D. Borojevic, “Space vector modulated three-phase to three-phase matrix converter with input power factor correction,” IEEE Trans. Ind. Applicat., vol. 31, pp. 1234–1246, Nov./Dec. 1995. [3] J. Oyama, X. Xia, T. Higuchi, and E. Yamada, “Displacement angle control of matrix converter,” in Proc. IEEE PESC’97, 1997, pp. 1033–1039. [4] P. Nielsen, “The matrix converter for an induction motor drive,” Ph.D. dissertation, Aalborg Univ., Aalborg East, Denmark, 1996. [5] P. Nielsen, D. Casadei, G. Serra, and A. Tani, “Evaluation of the input current quality by three different modulation strategies for SVM controlled matrix converters with input voltage unbalance,” in Proc. PEDES’96, vol. 2, 1996, pp. 794–800. [6] S. Bernet, T. Matsuo, and T. A. Lipo, “A matrix converter using reverse blocking NPT-IGBT’s and optimized pulse patterns,” in Proc. IEEE PESC’96, Baveno, Italy, 1996, pp. 107–113. [7] M. Hornkamp, M. Loddenkoetter, M. Muenzer, O. Simon, and M. Bruckmann, “EconoMAC the first all-in-one IGBT module for matrix converters,” in Proc. PCIM, 2001. [8] C. Klumpner, F. Blaabjerg, and P. Nielsen, “Speeding-up the maturation process of the matrix converter technology,” in Proc. IEEE PESC, vol. 2, 2001, pp. 1083–1088. [9] M. Ziegler and W. Hoffmann, “Semi-natural two steps commutation strategy,” in Proc. IEEE PESC’98, 1998, pp. 727–731. [10] D. W. Novotny and T. A. Lipo, Vector Control and Dynamics of AC Drives. Oxford, U.K.: Oxford Univ. Press, 1997, vol. 41. [11] F. Blaabjerg, U. Jaeger, S. Munk-Nielsen, and J. K. Pedersen, “Power losses in PWM–VSI inverter using NPT or PT IGBT devices,” IEEE Trans. Power Electron., vol. 10, pp. 358–367, Mar. 1995. [12] D. G. Holmes and T. A. Lipo, “Implementation of a controlled rectifier using AC–AC matrix converter theory,” IEEE Trans. Power Electron., vol. 7, pp. 240–250, Jan. 1992. [13] S. Ponnaluri and A. Brickwedde, “Generalized system design of active filters,” in Proc. IEEE PESC, vol. 3, 2001, pp. 1596–1601. [14] (2001) AlCap Calculation 3.01. Development Tools. [Online]. Available: www.epcos.com [15] S. Bernet and R. Teichmann, “Potential and risks of matrix converters for modern drives,” in Proc. 5th Brazilian Power Electronics Conf. (COBEP), Belo Horizonte, Brazil, 1997, pp. A1–A10. [16] P. Nielsen, F. Blaabjerg, and J. K. Pedersen, “New protection issues of a matrix converter—Design considerations for adjustable speed drives,” IEEE Trans. Ind. Applicat., vol. 35, pp. 1150–1161, Sept./Oct. 1999.
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Steffen Bernet (M’97) was born in Ilmenau, Germany, in 1963. He received the M.S. degree from the Technical University of Dresden, Dresden, Germany, in 1990 and the Ph.D. degree from the Technical University of Ilmenau, Ilmenau, Germany, in 1995, both in electrical engineering. His doctoral thesis contained the investigation of power semiconductors in soft switching converters. He was a Development Engineer with the Department of Private Communication Systems, Siemens, from 1994 to 1995. During 1995 and 1996, he was a Post-Doctoral Researcher in the Electrical and Computer Engineering Department, University of Wisconsin, Madison. In 1996, he joined ABB Corporate Research, Heidelberg, Germany, where he led several power electronics research projects and developed various novel high-power converters including resonant snubber based current source converters, matrix converters, and three-level voltage-source inverters. He led the Electrical Drive Systems Group, ABB Corporate Research, Heidelberg, from 1998 to 2001. In 1999, he was appointed Subprogram Manager for Power Electronics Systems, ABB Corporate Research Worldwide. In 2001, he joined the Technical University of Berlin, Berlin, Germany, as a Professor of Power Electronics. His main research areas are high-power converter topologies, power semiconductors, and motor drives.
Srinivas Ponnaluri was born in Guntur, India, on August 31, 1970. He received the B.E. degree from Ravishankar University, Raipur, India, and the M.E. degree in 1994 from Indian Institute of Science, Bangalore, India, both in electrical engineering. He is currently working toward the Ph.D. degree in the area of power quality as an external registrant at the Technical University of Aachen, Aachen, Germany. Currently, he is a Senior Scientist with ABB Corporate Research, Ladenburg, Germany, working in the area of power quality. He has approximately nine years experience in the industry, and he has been with ABB since 1994. He is the holder of several patents and has authored several publications. At ABB Corporate Research, he has worked on traction line-side converter design and control, energy metering, system design and control of STATCOMs, active filters, and UPS. His other areas of interests are ac motor drives and renewable energy systems.
Ralph Teichmann (S’96) was born in 1972 in Dresden, Germany. He received the M.S. degree in electrical engineering in 1997 from Dresden University of Technology, Dresden, Germany, where he is currently working toward the Ph.D. degree. From 1995 to 1996, he studied as a guest student at the University of Wisconsin, Madison. He has worked as an independent consultant for ABB Corporate Research, Germany. From 1998 to 1999, he was also a Research Assistant at Nagasaki University, Nagasaki, Japan. His research interests include high-power conversion, hard and soft switching, ac/dc, dc/ac, and ac/ac converter topologies, as well as converter controls.
