Dual-Mode Time-Sharing Sinewave-Modulation Soft ... - IEEE Xplore

IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 43, NO. 3, MAY/JUNE 2007

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Dual-Mode Time-Sharing Sinewave-Modulation Soft Switching Boost Full-Bridge One-Stage Power Conditioner Without Electrolytic Capacitor DC Link Nabil A. Ahmed, Hyun Woo Lee, and Mutsuo Nakaoka, Member, IEEE

Abstract—This paper is aimed at presenting a novel system topology and control scheme of a selective dual-mode pulse-modulated high-efficiency single-phase sinewave powerconversion circuit for the new energy generation and storage applications. This power-conversion system is composed of a timesharing-operated sinewave-absolute-modulation boost chopper with a bypass diode in the first power conditioning and processing stage and time-sharing sinewave partially pulse-modulated full-bridge inverter in the second stage. The proposed power conditioner is operated by a selective time-sharing dual-mode pulse pattern signal processing control scheme without electrolytic capacitor dc link. The unique operating principle of the two-power conditioning and processing stages with sectional time-sharing dual-mode partial sinewave-modulation scheme is described and discussed with a design example. In addition, this paper proposes also a sinewave tracking voltage controlled soft switching pulsewidth-modulation boost chopper with a bypass-diode loop, which includes a passive auxiliary edge-resonant snubber. The new conceptual operating principle and the control implementation of this novel power conditioner are presented and evaluated through experimental and simulation results. Index Terms—Boost chopper with bypass diode, full-bridge (FB) inverter, selective dual-mode control, sinewave absolute modulation, soft switching, time-sharing operation.

I. I NTRODUCTION

T

HE SMALL-SCALE distributed power generation systems for residential power utilizations and utility interactive as solar photovoltaic (PV) and fuel cell (FC) generation systems in addition to energy storage conditioning systems as super capacitor bank and new-type batteries have been become

Paper IPCSD-06-113, presented at the 2005 Industry Applications Society Annual Meeting, Hong Kong, October 2–6, and approved for publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Industrial Power Converter Committee of the IEEE Industry Applications Society. Manuscript submitted for review October 15, 2005 and released for publication January 18, 2007. This work was supported by the Ministry of Commerce, Industry and Energy (MOCIE) through the Industry and Energy Research Centre (IERC) program. N. A. Ahmed is with the College of Technological Studies, Public Authority of Applied Education and Training, Shuwaikh 70654, Kuwait, and also with the Department of Electrical and Electronics Engineering, Faculty of Engineering, Assiut University, Assiut 71516, Egypt (e-mail: [email protected]). H. W. Lee and M. Nakaoka are with the Electric Energy Saving Research Center (EESRC), Department of Electrical and Electronics Engineering, Kyungnam University, Masan 631-701, Korea (e-mail: lhwoo@ kyungnam.ac.kr). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIA.2007.895803

more and more popular from the earth environmental point of view. For further practical applications and cost-effective requirements, new concepts of topology and control implementation should be developed and evaluated, as compared with the high- or low-frequency transformer-linked converters. In general, the nonisolated single-phase sinewave power conditioning topologies have practical advantages such as lower cost, smaller size, higher power density, higher response, and higher efficiency [1]–[3]. These nonisolated system topologies that are used for new energy generation and energy storage systems are composed of two cascaded power conditioning and processing stages [4]. In the first stage, a high-frequency pulsewidth-modulation (PWM)-controlled boost-chopper-type dc–dc converter with an electrolytic capacitor dc link is required for boosting the low dc output voltage from PV modules, FC stacks, super capacitor, and/or chemical batteries bank. In the second stage, a sinewave-modulated full-bridge (FB) inverter, which is connected to the commercial utility ac power grid or stand-alone ac power effective utilization loads, is widely used. However, this conventional PWM boost chopper and sinewave PWM-inverter-based system has some disadvantages that must be solved as relatively poor power-conversion efficiency particularly in the low-output power-setting ranges due to switching and conduction power losses in boost chopper, electrolytic capacitor dc link, and the cascaded sinewave inverter system. Further, a bulky and temperature-dependent unreliable electrolytic dc smoothing capacitor bank, which has a short lifetime, impossible recycle easiness is actually required for constant voltage in the first power stage based on the PWM-controlled boost chopper. Moreover, this electrolytic dc capacitor bank in dc busline link must have a sufficiently large capacitance, relatively large volumetric physical size, heavy weight, and high-frequency ripple-current-related power loss due to its equivalent series resistance. Consequently, it is more difficult to implement cost-effective, compact, and highefficiency solar PV or FC power generation systems that are acceptable and suitable for miniaturization in size and light in weight [5]–[7]. In this paper, a novel prototype of selective time-sharing dual-mode sinewave PWM single-phase inverter and timesharing sinewave-absolute-modulated boost chopper with a bypass diode is proposed. This proposed power conditioner can, particularly, achieve high-efficiency power-conversion processing for wide range power-setting requirements. In addition, the dc-link capacitor between the first boost chopper and the second

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Fig. 1. Conventional single-phase power conditioner. (a) Circuit configuration. (b) Operating principle.

inverter power-conversion processing stages can be largely reduced from the practical point of view. Furthermore, this paper proposes an auxiliary edge-resonant passive snubbercircuit-assisted soft switching operation of the PWM dc boost chopper as a front stage of the proposed time-sharing dualmode sinewave tracking power conditioner for further efficiency improvement. The operating principle of this novel power conditioner and its dual-mode PWM time-sharing control scheme are evaluated and verified experimentally with a practical design model in terms of switching voltage and current waveforms, ν–i switching trajectory, and actual power-conversion efficiency. The practical effectiveness of the proposed time-sharing-controlled sinewave power conditioner without transformer isolation as compared with the conventional-type ac-link power conditioner is verified experimentally and through simulation results. It is proved that the proposed power conditioner is suitable and acceptable for the residential applications and stand-alone new energy-related power systems. II. C ONVENTIONAL T WO -S TAGE P OWER C ONDITIONER The basic system configuration of conventional single-phase power conditioner is schematically shown in Fig. 1(a). This power conditioner consists of boost-chopper-type dc–dc converter in addition to the FB single-phase sinewave inverter with low-pass filter in parallel with load. Its operating principle is also shown in Fig. 1(b). The boost chopper in the first power processing stage is used to boost the low and unregulated dc voltage from the PV module arrays or FC stacks up to a constant output voltage (dc 350–400 V). The active power switch SWC in this boost chopper always operates at high-frequency switching modulation to keep a constant output voltage in accordance with the

Fig. 2. Novel time-sharing single-phase power conditioner. (a) Circuit configuration. (b) Operating principle.

fluctuation voltage from the new energy sources. In general, the boost-chopper stage causes switching and conduction losses because of high-frequency switching. The output side of this boost chopper needs a bulky and large-volumetric electrolytic dc capacitor to keep constant dc voltage. It is actually impossible to implement a smaller and lighter power conditioner. In addition to these, the bulky electrolytic dc capacitor provides lower reliability such as power loss of equivalent shunt resistance based on ripple current, degradation, and short lifetime. The FB inverter in the second power processing stage is to produce 100/200-Vrms ac voltage for residential power applications or connect to the utility interactive ac power grid and release all available output power into the utility grid under sinewave carrier-based high-frequency PWM control. Consequently, the active power switches (SW1 –SW4 ) in the FB inverter produce switching and conduction losses. As a result, the total system efficiency of this power conditioner is very poor. III. B YPASS -D IODE -A SSISTED B OOST -C HOPPER C ASCADED S INEWAVE I NVERTER A. Circuit Description The circuit topology of the novel selective dual-mode timesharing sinewave-controlled soft switching PWM power conditioner presented in this paper is shown in Fig. 2(a). The proposed power conditioner is composed of the selective timesharing absolute sinewave voltage tracking boost chopper with bypass diode Db and complementary time-sharing sinewave voltage source FB inverter with low-pass filter. The selective time-sharing sinewave voltage tracking PWM boost chopper with a bypass-diode loop is used for boosting and converting the intermediate input side dc-link voltage from the PV modules or FC stacks to a constant quasi-sinewave ac absolute value.

AHMED et al.: DUAL-MODE TIME-SHARING SINEWAVE-MODULATION SOFT SWITCHING POWER CONDITIONER

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The complementary sinewave PWM-controlled FB inverter with a low-pass filter is operated selectively by a time-sharing dual-mode sinewave control processing scheme to produce the required ac output voltage (100/200 Vrms) for utility interface or for residential applications. B. Operation Principle The unique operating principle of this power conditioner with a bypass that is diode assisted is basically shown in Fig. 2(b), which is mainly based on the selective dual-mode time-sharing operation of the boost chopper and the FB inverter. As it is shown in Fig. 2(b), when the PWM boost chopper operates under the condition of instantaneous absolute sinewave pulse modulation to boost the input voltage, the single-phase FB inverter is designed as to operate as a synchronous polarity switching. On the other hand, when the sinewave voltage source FB inverter operates under the condition of instantaneous absolute sinewave pulse modulation to obtain the desired output voltage, the boost PWM chopper is designed to have a nonoperation condition. Therefore, as the proposed sinewave PWM power conditioner is not required to operate simultaneously both its power conditioning and processing stages like the conventional sinewave PWM conditioner, the total number of switching operation in the power conditioner can be reduced substantially. Consequently, the switching and conduction losses of its power conversion and processing stages, in addition to the power loss of electrolytic dc capacitor, can be all substantially reduced for the newly developed power conditioner. Moreover, the time-sharing sinewave voltage tracking PWM boost chopper is not required to keep a constant output voltage at the intermediate link between the chopper and the inverter, so the large-volumetric electrolytic dc capacitor bank between the first boost-chopper stage and the second FB inverter stage is conveniently unnecessary in practice. A small film capacitor that is commonly used for high-frequency ac power can be employed in place of the conventional bulky electrolytic dc capacitor. The capacitance of the reduced-scale film capacitor is substantially about 1/1000 times as compared to the conventional electrolytic capacitor that is used commonly for dc physical voltage smoothing. The film capacitor as a nonsmoothing dc rail link can effectively realize small and thin physical size, lighter weight, low power losses, high reliability, and long life. Further, during the operation of the FB inverter, the input current from dc supply does not flow through the boost inductor Lb and free-wheeling diode Dc , but it continuously flows through the bypass diode Db of the boost chopper. Therefore, the conduction losses of boost inductor Lb and free-wheeling diode Dc in the first power delivery stage are reduced, and consequently, high power-conversion efficiency can be achieved. IV. A BSOLUTE S INEWAVE V OLTAGE T RACKING C ONTROL S CHEME The selective time-sharing dual-mode absolute sinewave voltage tracking control strategy for the proposed single-phase sinewave PWM inverter and time-sharing sinewave absolute

Fig. 3. Boosted voltage conversion ratio versus duty cycle characteristics of PWM boost chopper.

PWM boost chopper in Fig. 2(a) is clearly summarized in the following. A. Operation Mode of PWM Boost Chopper When the input dc voltage Vin is less than the absolute value of the required sinewave output voltage νout , the switch SWC of the boost chopper operates at high-frequency switching mode for boosting the input voltage and producing the quasi-sinusoidal pulse-modulated waveform at the intermediate link between the two stages. On the other hand, the switches (SW1 –SW4 ) in the voltage-source-type FB inverter operate under the commercial frequency-based synchronous polarity switching. For example, when the positive half sinewave of output voltage is required, the switches SW1 and SW4 only are designed as to be in ON-state. When the negative half sinewave of output voltage is required, the switches SW2 and SW3 only are designed to be conducted. B. Operation Mode of FB Inverter When the input dc voltage Vin is greater than or equal to the absolute value of the required sinewave output voltage νout , the switch SWC in the boost chopper is always in OFF-state. The switches (SW1 –SW4 ) in the voltage source FB inverter operate under a condition of high-frequency switching sinewave carrier-based PWM switching mode. In this case, the input current from dc supply does not flow through the boost inductor Lb and free-wheeling diode Dc . It continuously flows through the bypass diode Db of the boost chopper. Therefore, the conduction losses of boost inductor Lb and free-wheeling diode Dc do not occur in this operating mode. C. Time-Sharing Sinewave Modulation The steady-state voltage conversion characteristic of the typical PWM boost chopper can be generally represented by νout =

Vin 1−D

(1)

where D is the duty cycle of the PWM boost-chopper switch SWC , Vin is the input voltage, and νout is the absolute output

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Fig. 4. Block diagram of the control circuit of time-sharing sinewave modulation.

voltage (absolute value of the desired sinusoidal output voltage) obtained from the PWM boost chopper. Rearranging (1), the duty cycle of the boost PWM chopper can be calculated from

D =1−

1 . νout /Vin

(2)

Using (2), the duty cycle D of the switch SWC , which is in the boost-chopper circuit, can be specified from the input voltage Vin and the absolute value of the desired sinusoidal output voltage νout . Fig. 3 shows the steady-state-boosted voltage conversion ratio (νout /Vin ) versus the duty cycle characteristics of the boost chopper. This operating characteristic is important, and it will be used for the experimental breadboard setup. Fig. 4 shows the block diagram of control circuit that is used to generate the gate pulse timing sequences for the proposed power conditioner. When Vin < |νout |, the boost chopper operates for boosting and producing the quasi-sinusoidal pulsemodulated waveform with duty ratio calculated from (2). The FB sinewave inverter operates by comparing a highfrequency triangular carrier signal with a sinusoidal reference signal, where the modulation index is designed to be more than unity. The gate pulse timing sequences of the proposed PWM boost-chopper switch SWC and the FB PWM inverter switches SW1 –SW4 are shown in Fig. 5. Besides the unique features and excellent advantages of the proposed time-sharing single-phase power conditioner with electrolytic dc capacitor less link discussed previously, the time-sharing sinewave PWM FB inverter operates around zeroor low-current value. Therefore, the switching and conduction power losses of the voltage source FB inverter stage are kept to be low as compared with the conventional one.

Fig. 5.

Gate pulse timing sequences of proposed power conditioner.

V. S OFT S WITCHING PWM B OOST C HOPPER A. Circuit Conﬁguration For further improvement of the efficiency of the proposed time-sharing power conditioner, this paper also presents a soft switching PWM boost chopper instead of the hard switching PWM boost chopper shown in Fig. 2(a). Fig. 6 shows the modified time-sharing dual-mode-controlled power conditioner, in which a soft switching time-sharing sinewave voltage tracking PWM boost chopper replaces the hard switching PWM boost chopper using a passive auxiliary resonant snubber circuit. The passive auxiliary resonant snubber circuit is composed of the resonant inductor Lr , the resonant capacitor Cr , the lossless snubber capacitor Cs , and the auxiliary diodes D1 –D3 . This


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Fig. 6. Modified time-sharing power conditioner with soft switching PWM boost chopper.

Fig. 8. Operating modes and operating waveforms. (a) Gate voltage pulse sequences. (b) Operating voltage and current waveforms. Fig. 7. Mode transitions and equivalent circuits of soft switching PWM boost chopper.

soft switching boost PWM chopper can operate at zero-voltageswitching (ZVS) commutation in the turn-OFF switching transition of the boost-chopper switch SWC . B. Operation and Mode Transition of Soft Switching PWM Boost Chopper The mode transitions and equivalent circuits of the sinewave voltage tracking soft switching PWM boost chopper during one switching period are depicted in Fig. 7. The gate pulse timing sequences of the active power switch SWC in the auxiliary passive resonant snubber assisted boost chopper shown in Fig. 5 is enlarged in Fig. 8, and the operating voltage and current waveforms of each component of the soft switching PWM boost chopper are shown in this figure. The operating principle in each switching mode transition of the soft switching PWM boost chopper is explained as follows. Mode 0 The stored energy of the boost inductor Lb and the input voltage Vin is transferred to the load side through the free-wheeling diode Dc . When the boost-chopper switch SWC is turned ON, Mode 0 shifts to Mode 1.

Mode 1 When the power switch SWC is turned ON, the current begins to flow in the passive auxiliary resonant snubber circuit. The resonant current flows through the resonant inductor Lr , the resonant capacitor Cr , and the lossless snubber capacitor Cs . In this mode, the resonant capacitor Cr is charging, and the lossless snubber capacitor Cs is discharging. The switch SWC is turned ON with the hard switching transition because the voltage and current of switch SWC have rapid dν/dt and di/dt performances. Mode 2 When the voltage across the snubber capacitor Cs becomes zero and the voltage across the resonant capacitor Cr is equal to the output average voltage Vo , the auxiliary diode D2 is turned OFF. Therefore, the resonant current that is flowing through the inductor Lr and the capacitor Cr and Cs becomes zero. All the circuit operations are identical to the conduction state of the conventional boost chopper with the boost inductor Lb . Mode 3 Due to the snubber capacitor Cs , it is safe to turn OFF the boost-chopper switch SWC with ZVS. When the power switch SWC is turned OFF with ZVS, the current flowing through the boost inductor Lb flows through the snubber capacitor Cs and the resonant capacitor Cr . Therefore, the

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TABLE I DESIGN SPECIFICATIONS AND CIRCUIT CONSTANTS

Fig. 9.

Simulated current waveforms through boost inductor Lb .

lossless snubber capacitor Cs starts charging gradually from zero voltage, and the resonant capacitor Cr starts discharging. When the voltage across the lossless snubber capacitor Cs is equal to the output average voltage Vo and the voltage across the auxiliary resonant capacitor Cr becomes zero, the diodes D1 and D3 are naturally turned OFF. At the same time, the diode Dc is turned ON, and Mode 3 shifts to Mode 0. VI. E XPERIMENTAL R ESULTS AND E VALUATIONS An experimental setup assembly is proposed to validate the steady-state performance evaluations of the proposed timesharing single-stage power conditioner. The design specifications and circuit parameters are listed in Table I. In this section, the performance evaluations of the converter under consideration will be evaluated and investigated for the given loading conditions through a digital computing program describing the different operating modes of the proposed power converter. In addition, simulation results based on the PSIM software are used too, and the results are nearly agreed. The computed, simulated, and experimental results will be discussed in the following.

Fig. 10.

Simulated voltage waveforms across capacitor Cc .

Fig. 11.

Simulated voltage waveform across FB inverter.

A. Simulation Results Fig. 9 shows the simulated time-sharing sinewave tracking current waveform through the boost inductor Lb . Fig. 10 shows the voltage waveform across the intermediate dc-link capacitor Cc . As shown in Figs. 9 and 10, the sinewave tracking boost PWM chopper with an auxiliary passive resonant snubber operates only when the input dc voltage Vin is less than the required absolute sinewave output voltage νout . Fig. 11 shows the simulated voltage waveform across the time-sharing dual-mode sinewave-modulated FB inverter before the low-pass filter. Observing this waveform, when the desired sinusoidal ac output voltage νout is greater than the

input dc supply voltage Vin , the boost PWM chopper is not operated. In this case, the bypass diode only works, the boostchopper stage is bypassed, and the FB inverter operates with the time-sharing partially controlled sinewave PWM strategy. The simulated output voltage and current waveforms of the proposed time-sharing power conditioner are shown in Fig. 12. As shown in this figure, a high-quality sinusoidal voltage and current waveforms can be obtained.


Fig. 12. Simulated output voltage and current waveforms.

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Fig. 15. Experimental current waveform through boost inductor Lb .

Fig. 13. Voltage and current waveforms at turn-OFF transition of SWC . (a) Hard switching. (b) Soft switching.

Fig. 16. Experimental voltage waveform across capacitor Cc .

Fig. 14. ν–i trajectory during turn-OFF transition of SWC . (a) Hard switching. (b) Soft switching.

B. Experimental Results and Discussions Figs. 13 and 14 show the experimental voltage and current operating waveforms and ν–i trajectory in case of turn-OFF switching commutation of the boost-chopper power semiconductor switch SWC of the boost chopper under hard switching and soft switching commutation conditions, respectively. From the voltage and current switching waveforms at turn-OFF hard switching commutation, as shown in Fig. 13(a), there is an overlapping region in the voltage and current switching transient waveforms. Moreover, the ν–i trajectory in Fig. 14(a) spreads out in the first quadrant in the ν–i plane, which increases the switching power losses. Observing the switching voltage and current operating waveforms shown in Fig. 13(b) under soft switching commutation

condition, except the overlapping period of the switching voltage and falling current during the tail current of insulatedgate-bipolar-transistor switch, there is no overlapping region of the switching voltage and current. In addition, the relevant ν–i trajectory shown in Fig. 14(b) is nearly moving along the voltage and current axes of the ν–i plane. Therefore, under soft switching, the switching power losses of the power switch SWC can be essentially reduced as compared to the hard switching operation. Figs. 15–18 show the experimental voltage and current operating waveforms. Comparing these figures with the simulated waveforms of Figs. 9–12, a good agreement between the simulated and experimental results is evident. C. Harmonic Contents and Actual Conversion Efﬁciency The output current harmonic contents are shown in Fig. 19. The ac output current waveforms of this sinewave power conditioner can produce high-quality sinusoidal waveforms with the maximum total harmonic current distortion of 2.82%. Fig. 20 shows the comparative actual power-conversion efficiency of the proposed time sharing and the conventional power conditioner. Observing the results in Fig. 20, the actual efficiency of the proposed-type power conditioner is much higher than that of the conventional one for all the required output

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Fig. 17. Experimental voltage waveform across FB inverter. Fig. 20.

Actual efficiency versus output power characteristics.

VII. C ONCLUSION

Fig. 18. Experimental output voltage and current waveforms.

Conventional two-stage energy conversion systems are bulky, expensive, provide low efficiency, and are, thus, not suitable for the small-scale new energy conversion utilizations. In this paper, a novel circuit topology of high-efficiency energy conversion system with single-stage architecture has been presented, which is suitable for the small-scale solar PV or FC power generation systems or new storage applications. The proposed power conditioner is composed of the time-sharing sinewave-absolute-modulated boost chopper with a bypass diode and the time-sharing single-phase sinewave PWM FB inverter. The unique operating principle and control implementation of the proposed power conditioner system have been described and discussed based on a design example in experimental point of view. In addition, this paper has introduced a front-stage soft switching sinewave tracking voltage controlled soft switching PWM boost chopper using the passive auxiliary edge-resonant snubber circuit to achieve further highefficiency power conversion. The excellent performance and the practical effectiveness of the proposed system as high power-conversion efficiency, high-quality sinusoidal voltage and current waveforms with low total harmonic distortion, more reduction in the intermediate capacitor size, and elimination of the electrolytic capacitor are proved and verified as compared to the conventional power conditioner from both the simulated and experimental results. R EFERENCES

Fig. 19. Output current harmonic contents.

power ranges. In particular, for the low-output power-setting condition, this boost chopper cascaded single-phase inverter that is operating at the time-sharing dual-mode sinewave PWM control scheme can realize higher efficiency characteristics as well as high-power density or system miniaturization. Moreover, the actual efficiency of the soft switching boost PWM is higher than that of the hard switching PWM for all the output power ranges.

[1] Y. Nishida, S. Nakamura, N. Aikawa, S. Sumiyoshi, H. Yamashita, and H. Omori, “A novel type of utility-interactive inverter for photovoltaic system,” in Proc. Annu. Conf. IEEE IECON, Nov. 2–6, 2003, vol. 4, pp. 2338–2343. [2] S. Saha, N. Matsui, and V. P. Sundarsingh, “Design of a low power utility interactive photovoltaic inverter,” in Proc. Int. Conf. Power Electron. Drives and Energy Syst. Ind. Growth, Perth, Australia, Nov. 1–3, 1998, vol. 1, pp. 481–487. [3] F. Kudo, T. Uematsu, T. Irihama, N. Yamada, and T. Ninomiya, “Experimental investigation of double-bridge inverter,” in Proc. Jpn. Nat. Conv. Rec. IEE, Sendai, Japan, Mar. 2003, vol. 4, p. 108. [4] H. Terai, S. Sumiyoshi, T. Kitaizumi, H. Omori, K. Ogura, H. Iyomori, S. Chandhaket, and M. Nakaoka, “Utility-interactive solar photovoltaic power conditioner with soft switching sinewave modulated inverter for residential applications,” in Proc. IEEE 33th Annu. PESC, Cairns, Australia, Jun. 23–27, 2002, vol. 3, pp. 1501–1505.


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[5] K. Ogura, N. A. Ahmed, O. Noro, H.-W. Lee, T.-K. Kang, T. Ahmed, and M. Nakaoka, “Dual mode time-sharing sinewave single-phase sinewave inverter with partially controlled sinewave absolute PWM soft switching boost chopper for new energy system,” in Proc. IPEC, Niigata, Japan, Apr. 4–8, 2005, pp. 1629–1631. [6] N. A. Ahmed, J.-Y. Lee, K.-Y. Suh, H. W. Lee, and M. Nakaoka, “High efficiency power conditioner using bypass diode assisted sinewave pulse modulation boost chopper-fed inverter with electrolytic capacitorless dc link,” in Proc. 8th ICEMS, Nanjing, China, Sep. 27–29, 2005, pp. 959–964. [7] N. A. Ahmed, T. Ahmed, H.-W. Lee, and M. Nakaoka, “Dual-mode timesharing one-stage single-phase power conditioner using sinewave tracked soft switching PWM boost chopper,” in Proc. 40th Annu. Meeting IEEE IAS, Hong Kong, Oct. 2–6, 2005, pp. 1612–1617.

Hyun-Woo Lee received the B.E. degree in electrical engineering from Dong-A University, Pusan, Korea, in 1979, the M.S. degree in electrical engineering from Yuing-Nam University, Kyungbook, Korea, in 1984, and the Ph.D. degree in electrical engineering from Dong-A University in 1992. Since 1985, he has been with the Electric Energy Saving Research Center, Department of Electrical and Electronics Engineering, Kyungnam University, Masan, Korea, where he is currently a Professor. His research interests include power electronics and new power generation systems. Dr. Lee is a member of the Korean Institute of Electrical Engineers (Academic Director).

Nabil A. Ahmed received the B.Sc. and M.Sc. degrees in electrical engineering from the Electrical and Electronics Engineering Department, Faculty of Engineering, Assiut University, Assiut, Egypt, in 1989 and 1994, respectively, and the Dr.-Eng. degree in electrical engineering from Toyama University, Toyama, Japan, in 2000. Since 1989, he has been in the Department of Electrical and Electronics Engineering, Faculty of Engineering, Assiut University, where he is currently an Associate Professor. He was a Post-Doctorate Fellow at the Electric Engineering Saving Research Center, Kyungnam University, Korea, from October 2004 to April 2005, and he was a Japan Society for the Promotion of Science (JSPS) Visiting Professor at Sophia University, Japan, from July 2005 to September 2006. He is currently an Associate Professor at the College of Technological Studies, Public Authority of Applied Education and Training, Shuwaikh, Kuwait. His research interests are in the area of power electronics, variable speed drives, soft switching converters, and renewable energy systems. Dr. Ahmed was the recipient of the Japanese Monbusho Scholarship, the JSPS Fellowship, the Best Paper Awards from the IEEE-ICEMS’04, IEEEICEMS’05, and IATC’06 conferences, and the Egypt State Encouraging of Research Prize 2005.

Mutsuo Nakaoka (M’83) received the Dr.-Eng. degree in electrical engineering from Osaka University, Osaka, Japan, in 1981. He was with the Department of Electrical and Electronics Engineering, Kobe University, Kobe, Japan, in 1981 and served as a Professor in the Department of Electrical and Electronics Engineering, Graduate School of Engineering, until 1995. He was also a Professor in the Department of Electrical and Electronics Engineering, Graduate School of Science and Engineering, Yamaguchi University, Yamaguchi, Japan. He is currently with the Electric Energy Saving Research Center, Department of Electrical and Electronics Engineering, Kyungnam University, Masan, Korea. His research interests include the application developments of power-electronic circuits and systems. Dr. Nakaoka is a member of the Institute of Electrical Engineering Engineers of Japan, Institute of Electronics, Information and Communication Engineers of Japan, Institute of Illumination Engineering of Japan, European Power Electronics Association, Japan Institute of Power Electronics, Japan Society of the Solar Energy, Korean Institute of Power Electronics, and IEE-Korea. He is currently a Chairman of the IEEE Industrial Electronics Society Japan Chapter. He was the recipient of more than ten awards such as the 2001 Premium Prize Paper Award from IEE-U.K., the 2001 and 2003 Best Paper Awards from IEEEIECON, the 2000 Third Paper Award from IEEE-PEDS, and the 2003 James Melcher Prize Paper Award from IEEE-IAS.