Electromagnetic Field Simulation of MMICs Including ...

Proceedings of the 47th European Microwave Conference

Electromagnetic Field Simulation of MMICs Including RF Probe Tips D. M¨uller∗ , J. Sch¨afer∗ , D. Geenen∗ , H. Massler† , A. Tessmann† , A. Leuther† , T. Zwick∗ and I. Kallfass‡ ∗ Institute

of Radio Frequency Engineering and Electronics, Karlsruhe Institute of Technology, 76131 Karlsruhe, Germany † Fraunhofer Institute for Applied Solid State Physics, 79108 Freiburg, Germany ‡ Institute of Robust Power Semiconductor Systems, University of Stuttgart, 70569 Stuttgart, Germany

Abstract—RF probe measurements are widely used for characterizing circuits in the millimeter-wave frequency range. Especially above 200 GHz the large dimensions of the RF probe, in comparison to the wavelength, lead to parasitic effects which will effect the device under test. Since the measurement is influenced by the electro-magnetic coupling between probe tip and the test structures on the substrate, the standard on-wafer calibration becomes unprecise. This paper discusses a method of simulating this influence by means of 3D electromagnetic simulations. Confirmation of the approach was done by comparing measured and simulated results of impedance substrate standards and galium arsenid (GaAs) monolithic millimeter-wave integrated circuits (MMICs). A very good agreement to measured results in the investigated WR3 frequency range (220 - 325 GHz) is presented, showing effects which were unobserved in earlier simulations.

Fig. 1: Probe-tip CAD model and microscopic photograph of a WR3 Cascade Microtech® Infinity Probe®

II. M ODELING OF THE PROBE TIP

I. I NTRODUCTION The trend towards higher frequencies in antenna and circuit design is increasing the simulation challenges drastically. With dimensions of circuit components and transmission lines in the same order as the wavelength, the behavior of the components show and increase in parasitic effects, such as stray fields or the excitation of unwanted modes. Modern monolithic millimeter-wave integrated circuit design (MMIC) tries to constantly miniaturize the circuits and therefore shift these issues above the frequency of interest. An important bottleneck of this miniaturization is the contact to the outer world, the RF contact pad. The minimum dimensions of the contact pads depend on the boundaries given by the measurement system and available packaging technology. Limiting factors are either imposed by the pitch of the RF probe or the minimum requirements in terms of wire bonding or flip-chip packaging. With this in mind it is obvious that the pad dimensions cannot be downscaled arbitrarily. In the WR3 frequency range (220 325 GHz) this leads to a situation where the dimensions of the RF contact pad and the RF probe tip itself are in the order of the wavelength, therefore showing unwanted parasitic effects such as radiation and coupling[1]. This often results in a large deviation between the measurement and initial simulation. Since these effects strongly depend on the coupling of the probe with its environment, a general model of the RF probe tip behavior is not possible. Therefore, a new approach is presented, which includes the probe tip into the simulation of the MMIC to recreate the real world procedure as close as possible, including the calibration.

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The probe tip used in the measurement setup consists of a micro-coax conductor which has a thin-film membrane soldered to it [2]. The membrane itself embeds a coax-tomicrostrip transition and the wafer contacts, which establish the interconnect to the MMIC. The modeling was simply done by using the information given in [2] and [3] together with several photographs taken by a conventional microscope which were evaluated and measured. Based on this data a 3D CAD model was built using CST Microwave Studio® , remodeling the first millimeter of the probe tip and neglecting all remaining parts such as casing and coax-to-waveguide transition. While this is sufficient for the application shown in this paper, to investigate the influence of the RF probe in antenna measurements, it may be necessary to model a larger portion of the probe. Fig. 1 shows the final CAD model next to a microscopic photograph of the probe tip. III. M EASUREMENT S ETUP The measurements were done using an on-wafer setup employing a set of WR3 Cascade Microtech® Infinity Probes® with a pitch of 100 μm. S-parameters were recorded using a Keysight PNA-X network analyzer with Oleson Microwave Labs WR3 extension modules. The devices under test (DUT) were the Cascade Microtech® impedance standard substrate (ISS) 138-357 and two MMICs which were fabricated by the Fraunhofer IAF on a GaAs mHEMT technology [4] and are more closely described in section IV-C and IV-D. The whole measurement system was calibrated to the probe tip

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10–12 Oct 2017, Nuremberg, Germany

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Line K6 Fig. 2: CAD model of two probe tips contacting the reference line of the TRL calibration standards.

Measurement Probe Tip Simulation

reference plane by the thru-reflect-line (TRL) method using the standards available on the ISS.

Fig. 3: 50 Ω Smith diagram showing measurement and simulation of the reflection coefficient S11 for different ISS calibration standards in the frequency range 200 - 330 GHz. Since the Line K6 was used in the calibration it is perfectly matched and therefore not visible.

IV. E VALUATION OF THE PROBE MODEL A. Calibration of the probe tips In order to reproduce the measurement conditions and for de-embedding the probe tips from the simulation results a TRL calibration with simulated standards was performed. For this purpose a CAD model of the ISS was modeled and the same standards which were used in the measurement calibration were simulated and used to calculate a corresponding error set. The material properties and dimensions of the ISS were taken from [5] and microscopic photographs. As an example Fig. 2 shows the CAD model for the simulation of the reference line contacted with a probe tip on each side. As recommended in the calibration manual, the ISS was placed on an absorptive mount for measurement and simulation.

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S21 in dB

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B. Evaluation using the ISS For the initial evaluation of the simulation method and the CAD probe model a simple DUT is favorable. This minimizes the risk of inaccurate modeling and material properties in the electromagnetic (EM) simulation. Luckily the ISS which is used in the calibration has a very simple layer stack and is therefore well suited for this task. The ISS standards Through, Line K6, Open, Short and Match were measured and simulated. Furthermore, to get a result which is only dependent on the probe model and not the contacted environment the so called ’open in air’ was measured and simulated. In this state the probe is not contacted to a standard but floats several 100 μm above the DUT. The probe spacing of the ’open in air’ measurement was 180 μm, what is equivalent to the probe spacing of the Line K6 standard. TRL calibrated simulation and measurement results are shown for S11 in the Smith diagram in Fig. 3 and for the transmission S21 in magnitude and phase in Fig. 4. Comparison of the simulation and measurement of S11 shows only little deviation in the magnitude while the measured phase shows more relative movement. The transmission coefficient S21 also shows very good agreement between simulation and measurement even at high attenuation levels. The remaining deviations are most probably due to uncertainties in the CAD probe model and unavoidable inaccuracies in the positioning of the probes while

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Fig. 4: Measurement (dotted) and probe tip simulation (solid) of the S21 magnitude and phase for different ISS standards.

recording the measurements. Even with high caution, it is challenging to manually reproduce contact position accuracies better than ±5 μm. At 330 GHz an offset of 1 μm results in a phase shift of roughly 0.9°, therefore certain deviations between two measurements are unavoidable. C. Evaluation of a GaAs transmission line To evaluate possible deviations between measurement and simulation, rather simple MMICs were chosen. A transmission line with RF probing pads on both sides and a passive reflective type phase shifter (RTPS) with RF probing pads at the in- and output. The reason for selecting these MMICs is their passive nature since they do not make use of active tran-

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S11 , S21 in dB

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D. Evaluation of a GaAs reflective type phase shifter To verify the probe tip simulation on a more complex device, a reflective type phase shifter (RTPS) was measured and simulated. An RTPS consist of a 90° hybrid coupler with two adjustable highly reflecting loads at the through- and coupledport. If both loads are identical, the reflected signals cancel each other out at the input, while constructive interference appears at the usually isolated port of the coupler. In practical systems, this leads to very good input- and output-matching

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Fig. 6: S-parameter results of measurement and different simulation methods for the transmission line GaAs DUT. 150 128 106 85 63 42 20

|E| in kV/m

sistors and therefore possible uncertainties due to the transistor models are eliminated. The pad’s dimensions are designed in a way that, when excited with a coplanar waveguide mode, the characteristic impedance is close to 50 Ω. Chip photographs of the MMICs are shown in Fig. 5. To make the EM simulation possible the layer stack of the Fraunhofer GaAs process was translated into CST Microwave Studio® . Afterwards the whole MMIC was imported and contacted with two of the probe tip models at the input and output. Additionally to this simulation, the MMIC was EM simulated using a so-called waveguide port, which is a standard approach for signal excitation in CST Microwave Studio® . The simulated results together with measurements are shown in Fig. 6. Due to the reciprocity of the circuit only S11 and S21 are shown. The simulation using waveguide port excitation predicts an input matching better than −18 dB with a transmission loss of less than 1 dB. While such characteristics would fit the expected behavior of a 50 Ω transmission line, the actual measured results show strong deviations. Especially around 300 GHz the measured transmission magnitude shows a drop down to −5 dB while the input matching is only at −11 dB. On the other hand, utilizing the simulation including the probe tips shows much better prediction of the measured behavior. The drop in transmission magnitude around 300 GHz as well as the behavior of S11 is predicted very accurately. Further investigations showed that at frequencies around 300 GHz below the RF pad a parasitic mode was excited which radiated towards the RF probe. Fig. 7 shows two z-axis cross sections of the electric field magnitude at the substrate center for 200 GHz and 300 GHz. For better orientation the field magnitude is overlayed with the simulated MMIC whereby the RF probes are hidden. The comparison of both figures shows clearly the unwanted parasitic mode around 300 GHz which radiates towards the RF probe.

S11 , S21 in Degree

Fig. 5: Chip photograph of the transmission line MMIC (a) and reflective type phase shifter MMIC (b). The chip size of each MMIC is 0.5 mm × 0.5 mm.

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Fig. 7: Cross section of the electric field distribution for the EM simulation of the transmission line MMIC.

and de-coupling of the reflective loads to the input and output of the circuit [6]. In comparison to the simulation setup of the transmission line DUT, the setup of the RTPS is more complex. The varactor diodes which are used as reflective loads cannot be simulated in the EM simulation. To overcome this issue, the varactor diodes are removed from the simulation and waveguide ports are inserted instead. Additionally to the RF probe models, a simple model of the DC probe, used for bias supply in the measurement, was added to the simulation. In post-processing the EM simulation results are de-embedded and connected to the varactor diode model using Keysight Advanced Design System. A picture of the final EM simulation setup is shown in Fig. 8. The results of the post-processed simulation together with the measurement is shown in Fig. 9. While the waveguide port excitation predicts a maximum

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relative S21 in Degree

insertion loss of 7 dB the measured loss increases up to 12 dB. On the other hand, the EM simulation including the probe tips predicts this behavior much better. Similar results are visible in the input matching S11 where the waveguide port excited simulation predicts a matching better than −13 dB while the measured results show a matching less than −7 dB at 300 GHz. This is also accurately predicted by the probe tip EM simulation. Evaluation of the EM simulation showed that, similar to the results shown in the previous section, a parasitic mode was excited, radiating towards the probe at frequencies around 300 GHz. Altogether the measurements shows strong deviations to the waveguide port excited simulations while the simulation embedding the probe tips follows the measurements in very good agreement. The simulated relative phase shift shows certain deviations between the measurement and both simulation methods. While the maximum relative phase shift of approximately 115° was predicted by both simulation methods, the shape over frequency is not met. Since both simulation methods show approximately the same phase shift, the assumption is, that the phase shift of the varactor model is not properly modeled.

S21 in dB

Fig. 8: Final setup of the reflective type phase shifter simulation, showing RF probes and DC probes contacting the MMIC.

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Fig. 9: Measured and simulated results of the RTPS.

ACKNOWLEDGMENT We express our gratitude to our colleagues in the IAF technology department for their excellent contributions during epitaxial growth and wafer processing. This work was supported by the Deutsche Forschungsgemeinschaft (Grant KA 3062/10-1).

V. C ONCLUSION

R EFERENCES

The much better agreement between measurement and simulation clearly proves the large advantage of simulating the RF probe in conjunction with the whole DUT. Especially the RF probe to pad interconnect seems to be sensitive for parasitic effects. As presented above these effects are not visible if the widely used waveguide port excitation is used. Embedding the RF probes into the EM simulation as described above on the other hand allows to precisely analyze this interconnect and optimize the RF probe and RF pad geometries. Further investigations showed that in the presented GaAs DUTs part of the deviation from measurement to waveguide port excited simulation could be accounted to a resonant mode which is excited below the RF pad. This kind of troubleshooting is not possible without including RF probe tips into the simulation. The presented method is therefore an optimal method to optimize pad structures for minimal RF probe-to-pad influence. Finally, the presented method can easily be adapted to probes and calibration substrates of different manufacturers and are therefore a versatile tool for future MMIC designs.

[1] R. Campbell, “Near and far field e and h in sub-mmwave on-wafer probes,” in Cascade Microtech COMPASS 2016 Users’ Conference, 2016. [2] L. S. Richard Campbell, Michael Andrews and A. Fung, “Membrane tip probes for on-wafer measurements in the 220 to 325 ghz band,” Proc. 18th Int. Symp. Space Terahertz Technology, 2007. [3] “The infinity probe for on-wafer device characterization and modeling to 110 ghz.” [Online]. Available: https://www.cascademicrotech.com/files/ INFINITY QA.pdf [4] A. Leuther, A. Tessmann, M. Dammann, C. Schworer, M. Schlechtweg, M. Mikulla, R. Losch, and G. Weimann, “50 nm mhemt technology for g- and h-band mmics,” in Indium Phosphide Related Materials, 2007. IPRM ’07. IEEE 19th International Conference on, may 2007, pp. 24 –27. [5] “Impedance standard substrate.” [Online]. Available: https://www. cascademicrotech.com/files/iss map 138-357.pdf [6] D. M¨uller, S. Reiss, H. Massler, A. Tessmann, A. Leuther, T. Zwick, and I. Kallfass, “A h-band reflective-type phase shifter mmic for ism-band applications,” in 2014 IEEE MTT-S International Microwave Symposium (IMS2014), June 2014, pp. 1–4.

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