2010 Asia-Pacific International Symposium on Electromagnetic Compatibility, April 12 - 16, 2010, Beijing, China
Electromagnetic Interference in Substrate-Integrated Waveguides Circuit and Its Suppression Technique Ruey-Bing Hwang 1 , Cheng-Yuan Chin 1 , Yu-De Lin 1 , Toshihide Kitazawa 2 , Chang-Yu Wu 3 1
2
3
Department of Electrical Engineering, National Chiao-Tung University Hsinchu, Chinese Taipei
Department of Electrical and Electronic Engineering, Ritsumeikan University Kyoto, Japan
Department of Electronic Engineering, Jinwen University of Science and Technology Taipei County, Chinese Taipei
[email protected]
Abstract—As was well known, the substrate-integrated waveguide can be easily implemented and integrated with active and passive devices using the printed circuit process. Thus, the integration circuit and system based on the substrateintegrated waveguide technique becomes achievable. However, in the integrated circuit environment, the close proximity between adjacent substrate-integrated waveguides may cause the electromagnetic coupling, and thus the electromagnetic interference problem arises accordingly. In this research, we demonstrate the electromagnetic coupling between two parallel substrateintegrated waveguide. Moreover, the theory of Fabry-Perot resonator was employed to explain the resonance coupling between the substrate-integrated waveguides. On the other hand, a new substrate-integrated waveguide equipped with moats [1] was utilized to examine the capability of cross-talk suppression. Interestingly, we find that in comparison with the conventionally used substrate-integrated waveguide the new one can effectively suppress the electromagnetic wave coupling. Therefore, the signal integrity can be further improved in a microwave or millimeter-wave circuit or system implemented using the substrate-integrated waveguide technique.
I. I NTRODUCTION Metal waveguides were popular during and after World War II. However, the bulky waveguides were gradually replaced by transmission lines printed on a circuit board, such as micro-strip line, strip line, coplanar waveguide, slot line, and et al. Recently, a printed waveguide mimicking the dispersion property of a metal waveguide was successfully developed [2]. Specifically, such a waveguide can be easily fabricated using printed circuit process, and therefore, attracts considerable attentions [3], [4]. It is termed as SIW (substrate-integrated waveguide) or post-wall waveguide. This waveguide uses the via-hole arrays to replace the metallic walls for guiding electromagnetic waves [5]. Besides, since the wave guided in the dielectric substrate with relative dielectric constant larger than that of air, the dimension of the waveguide can be further reduced. Moreover, several transition structures using printed transmission lines were developed to offer a much greater
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degree of flexibility in practical applications. Regarding the theoretical investigation relating the guidedwave and leaky-wave characteristics of the SIW [6], [7], various numerical method, such as the finite-difference frequencydomain method [8], [9], generalized multipole technique [10], and numerical multimode calibration procedure incorporating the finite-element method [11], were developed. In addition to the rigorous full-wave analysis [12], the empirical formula for practically designing the SIW was also developed. As was indicated in [11], the via-hole array is equivalent to a solid metal wall as the ratio of pitch width to via-hole radius is smaller than 4 and that of waveguide width to via-hole radius is greater than 8. In addition to the commonly used SIW, a new type of SIW equipped with moat (slit) outside the via-hole array [1] was developed. From the measured scattering parameters and calculated dispersion relation, it is apparent to see that the new structure can maintain an excellent guiding characteristic but using a sparser via-hole array compared with the conventionally used one. Besides, the cutoff frequency can be reduced to broaden its operational bandwidth. Incidentally, since numerous microwave and millimeterwave components, such as filter, resonator, multiplexer, and even slotted antennas, have been designed based on the SIW, the integration of SIW-based components and circuit in a printed circuit board becomes an option for microwave circuit and system. However, as far as the integration is concerned, the electromagnetic coupling between lines in close proximity shall be a problem. To evaluate the electromagnetic coupling between SIWs, the two parallel and identical SIWs (like a coupler) were implemented on a printed circuit board to measure its coupling coefficient. It is interesting to observe that although the via-hole array is dense enough, the resonance coupling occurs in the structure and, moreover, the coupling strength does not decrease as the separation distance between the two SIWs increases. As will become clear later on, the
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region bounded by the two SIWs can be regarded as a FabryPerot resonator for contributing the resonance coupling.
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Simulation d=7mm, w/o moats d=7mm, with moats
II. S TRUCTURE C ONFIGURATION
S21 and S31(dB)
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Frequency (GHz) Fig. 2. Simulated transmission and coupling coefficients for the SIWs with and without moats 0
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Measurement d=7mm, w/o moats d=7mm, with moats
S21 and S31(dB)
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Fig. 1. Structure configuration of the adjacent co-parallel substrate-integrated waveguides with moats: p = 1.5 mm, a = 0.75 mm, r = 0.5 mm, d = 7 mm, and w = 9.15 mm, respectively.
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Frequency (GHz) Figure 1 depicts the structure configuration to be analyzed and measured in this research. As shown in this figure, two adjacent and co-parallel SIWs was fabricated in a microwave dielectric substrate RO4003 with relative dielectric constant 3.55. The SIW was fed by a micro-strip line with tapering transitions at its both ends. The via-hole radius, pitch width between adjacent via-hole, moat width, and channel width of the waveguide are denoted by r, p, w and a, respectively. Parameter d represents the separation distance between the two SIWs. III. E LECTROMAGNETIC C OUPLING B ETWEEN SIW S AND A S TRATEGY FOR C ROSSTALK R EDUCTION From the results reported in [1], we may clearly observe that the moats can further suppress the electromagnetic-wave leakage from the via-hole array. Intuitively, such a new waveguide may mitigate the electromagnetic wave coupling between two adjacent and co-parallel waveguides.
Fig. 3. Measured transmission and coupling coefficients for the SIWs with and without moats
In the following example, we fabricated two SIW circuits with and without moats, respectively, shown in Fig. 1, each of which contains two SIWs separated by the same distance d. The structure parameters were attached in the figure caption for easy reference. It is noted that the pitch width in the two cases both are 1.5 mm, which was designated based on the criterion reported in the literature [11]. Figure 2 and 3 demonstrate the scattering parameters including the transmission- and coupling- coefficient for the coupled SIWs with and without moats, where Fig. 2 and 3 are simulatedand measured-results, respectively. The lines drawn in blue and black colors respectively represent the result of the structure with and without moats. The numerical simulation was
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the coupled-wave theory, we know that if the coupling is due to the evanescent wave outside the waveguide, the coupling coefficient decreases as the separation distance increases. However, those spikes are still present and do not decrease with the increase in the separation distance. We were inspired by this phenomenon to study the coupling mechanism between the SIWs. The electric field (strength of the Ez component) distributions at each resonant frequency in the SIWs were calculated. However, due to the page limitation only the case for the second resonant frequency was shown in Fig. 4. From this figure, it is apparent to see the resonant mode occurring in the region between the two waveguides. In the presence of the two via-hole arrays, the electric field distribution along the transverse plane resembles that of TE10 mode in the SIW. Moreover, the two open ends cause a standing wave along the x-axis. Therefore, the electric-field distribution exhibits a resonance behavior shown in the figure. It may be concluded that the resonance coupling is mainly due to the wave leaky from the first SIW, penetrating through the via-hole array into the Fabry-Perot resonator, and finally coupling to the second SIW. Thus, the resonant frequency should be correctly predicted by the formula given below.
(a)
f=
c √
2π r
(
mπ 2 nπ ) + ( )2 , l w
(1)
where l and w are the length and width of the cavity, c is the light speed, and m and n are integers representing mode indices along the x- and y-axis, respectively. Since the cavity height (substrate thickness) is much smaller than those of the other two dimensions, the fields along the thickness direction can be assumed to be uniform and the formula can be simplified as (1). TABLE I R ESONANCE FREQUENCIES OBTAINED FROM THE MEASUREMENT AND THE FORMULA IN (1).
Measurement (GHz)
Calculation (GHz)
Error (Percentage)
9.22
9.3004
0.87
(b)
9.885
9.8982
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Fig. 4. Contour map of the strength of electric field Ez (x, y) in coupled SIW structure (a) without moats, and (b) with moats. The operation frequency is 9.9 GHz.
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based on the Time-Domain Finite-Integration Method (CST: Microwave Studio). From Fig. 2 and 3, it is obvious to see that the coupling coefficient (S31 ) is significant although the via-hole array is dense enough. Specifically, there are spikes present in both the simulated and measured results. Although not shown in this paper, we have changed the separation distance between the two waveguide to measure its coupling coefficient. From
We have compared the resonant frequencies between the measured result and theoretical one obtained from (1), shown in Table 1. Because of an excellent agreement between the measured and calculated one for each frequencies, we may conclude that they are caused by the cavity resonance. The three resonant frequencies corresponds to the three resonant modes (m=0, n=1), (m=1, n=1), and (m=2, n=1), respectively. Additionally, from equation (1), we may conjecture that the resonance coupling will start at a lower frequency as the separation distance increases. It was confirmed by the nu-
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merical and experimental results to be demonstrated in our presentation. On the contrary, the resonance coupling does not occur in the case of SIWs with moats. Owing to the excellent confinement of the electromagnetic fields in the new SIWs, the cavity modes are hard to be excited and the electromagnetic coupling can be further suppressed.
[11] F. Xu and K. Wu, “Guided-wave and leakage characteristics of substrate integrated waveguide,” IEEE Trans. Microwave Theory Tech., vol. 53, no. 1, pp. 66–73, Jan. 2005. [12] N. Ranjkesh and M. Shahabadi, “Loss mechanisms in SIW and MSIW,” PIER B, vol. 4, pp. 299–309, 2008.
IV. C ONCLUSIONS In this paper, we numerically and experimentally examined the electromagnetic coupling between two SIWs that are close to each other. It is interesting to find that the coupling is considerable even the via-hole is dense enough. Through a systematical investigation in the electric field distribution, we may conclude that such a coupling is caused by the excitation of resonant modes in the cavity between the two SIWs. In addition to the demonstration for the electromagnetic coupling taking place in conventionally used SIWs, we also provide a strategy for suppressing the coupling by adding moats outside the via-hole array. The excellent performance in preventing the wave leakage from the waveguide effectively removes the path of electromagnetic interference. ACKNOWLEDGMENT The corresponding author of this paper wish to acknowledge R for their support in providing software for numerical CST simulation. R EFERENCES [1] R.-B. Hwang and C.-Y. Chin, “Substrate integrated waveguides with moats,” Journal of Electromagnetic Waves and Applications, vol. 23, pp. 1101–1112, 2009. [2] H. Uchimura, T. Takenoshita, and M. Fujii, “Development of a laminated waveguide,” IEEE Trans. Microwave Theory Tech., vol. 46, no. 12, pp. 2438–2443, Dec. 1998. [3] J. Hirokawa and M. Ando, “Single-layer feed waveguide consisting of posts for plane tem wave excitation in parallel plates,” IEEE Trans. Antennas Propagat., vol. 46, no. 5, pp. 625–630, May 1998. [4] D. Deslandes and K. Wu, “Integrated microstrip and rectangular waveguide in planar form,” IEEE Microwave Wireless Compon. Lett., vol. 11, no. 2, pp. 68–70, Feb. 2001. [5] K. Wu, “Integration and interconnect techniques of planar and nonplanar structures for microwave and millimeter-wave circuits-current status and future trend,” in Proc. IEEE APMC’01, vol. 2, Taipei, Taiwan, Dec. 3–6, 2001, pp. 411–416. [6] A. Zeid and H. Baudrand, “Electromagnetic scattering by metallic holes and its applications in microwave circuit design,” IEEE Trans. Microwave Theory Tech., vol. 50, no. 4, pp. 1198–1206, Apr. 2002. [7] Y. Cassivi, L. Perregrini, P. Arcioni, M. Bressan, K. Wu, and G. Conciauro, “Dispersion characteristics of substrate integrated rectangular waveguide,” IEEE Microwave Wireless Compon. Lett., vol. 12, no. 9, pp. 333–335, Sep. 2002. [8] F. Xu, Y. L. Zhang, W. Hong, K. Wu, and T. Cui, “Finite-difference frequency-domain algorithm for modeling guided-wave properties of substrate integrated waveguide,” IEEE Trans. Microwave Theory Tech., vol. 51, no. 11, pp. 2221–2227, Nov. 2003. [9] Y. L. Zhang, W. Hong, F. Xu, K. Wu, and T. J. Cui, “Analysis of guided-wave problems in substrate integrated waveguides - numerical simulations and experimental results,” in IEEE MTT-S Int. Micro. Symp. Dig., vol. 3, Jun. 8–13, 2003, pp. 2049–2052. [10] N. Talebi and M. Shahabadi, “Application of generalized multipole technique to the analysis of discontinuities in substrate integrated waveguides,” PIER 69, pp. 227–235, 2007.
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