High-Speed Electric Drive for Exhaust Gas Energy ... - Semantic Scholar

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 6, JUNE 2014

High-Speed Electric Drive for Exhaust Gas Energy Recovery Applications Fabio Crescimbini, Member, IEEE, Alessandro Lidozzi, Member, IEEE, Giovanni Lo Calzo, and Luca Solero, Member, IEEE

Abstract—High-speed electric drives play an important role in the field of power generating units on board vehicles and aircrafts. This paper deals with the solutions for developing the direct coupled electric drive to be used in combination with a radial turbo-expander for exhaust energy recovery in automotive applications. The descriptions of prototypal realization of both the axial-flux permanent-magnet (PM) generator and the three-level boost-rectifier converter, which results as the preferred topology for the controlled rectifier, are given. The high rotational speed of the direct-driven PM generator results in high electric fundamental frequency also, which is challenging for the electric drive control issues. Results of the electric drive prototype experimental activity are finally presented. Index Terms—AC–DC power electronic converters, automotive applications, axial-flux permanent-magnet (PM) (AFPM) machines, direct coupling.

I. I NTRODUCTION

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LECTRIC drives having very high rated speed are being proposed in the recent technical literature for use in combination with microgas or air turbines to arrange power generating units having rated power in the range from tens of watts to a few kilowatts [1], as well as for drives required by tooling machines and molecular pumps [2]. Considerable interest is recently raised for automotive applications, in which electric drives having rated speed in the range of thousands of revolutions per minute are expected to play an important role as power generating units required to meet the increasing demand of electric energy on board the nextgeneration vehicles [3]. This trend is driven by the increasing number of electrically powered ancillary devices as well as by the introduction of a wide range of new functionalities on board vehicles. As a result of having an increasing number of electrical components being used in automobiles for improving the vehicle’s performance, comfort, convenience, and safety, the rating power of electrical generating systems required on board vehicles is rapidly growing, and the next-generation

Manuscript received February 18, 2013; revised May 20, 2013; accepted June 9, 2013. Date of publication June 28, 2013; date of current version December 20, 2013. This work was supported primarily by the PRIN 2008 Program of the Italian Ministry for Education, University and Research (MIUR) under Award 20085BP47Z. The authors are with the Department of Engineering, University of Roma Tre, 00146 Rome, Italy (e-mail: [email protected]; [email protected]; [email protected]; [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2013.2271602

vehicles are expected to require electric power in the range from 4 to 6 kW to supply automobile loads such as air conditioners, electric steering systems, electric brakes, and high-energy discharge lamps. The envisaged new scenario is actually forcing car manufacturers to explore newly conceived solutions concerning the overall electrical system being utilized on board automobiles, as the present generating system based on using a Lundelltype alternator has become too inefficient whenever requested to deal with higher power output. In fact, as a result of the increased electrical power requested on board, the power loss in a Lundell-type alternator is too high, and the 14-V voltage level being used in today’s cars results in increased currents and, thereby, thicker wiring harnesses. As a consequence, the cost of the overall electrical system increases while the performance drops significantly. Based on the aforementioned considerations, electrical systems based on 42-V rating voltage are being widely accepted as an incoming standard for automotive applications, and various 42-V power-net architectures have been proposed since the last years. If a full 42-V architecture system has to be implemented in automobiles, there are many vehicle devices that would require a change of design and qualification to accept a 42-V power supply. Hence, to start with, most car makers are planning to incorporate dual voltage architectures being suitable to supply both 42- and 14-V loads. To date, automotive electrical systems likely draw the power required to drive the alternator from the mechanical shaft of the internal combustion engine (ICE), and thereby, the increased electrical power demand inevitably results in increased fuel consumption. However, from recognizing that the ICE exhaust gases still retain a significant amount of energy being usually wasted, substantial fuel saving can be achieved by using a radial turbo-expander which can provide recovery of the kinetic energy available from the ICE exhaust gases to directly drive an electrical generator as it is depicted in Fig. 1 [4]–[6]. Then, the electrical generator output is rectified through a controlled rectifier in order to suitably supply the 42-V power-net architecture within the variable rotating speed region of the radial turbo-expander. The turbo-expander is a radial turbine where the energy conversion relies on the component of the gas speed being perpendicular to the rotation axis. Hence, in the volute of the radial turbine, the exhaust-gas pressure is being converted into kinetic energy to move the turbine wheel, thus making mechanical power available onto the alternator shaft. As much as in the generating system being the subject matter of this

0278-0046 © 2013 IEEE

CRESCIMBINI et al.: HIGH-SPEED ELECTRIC DRIVE FOR EXHAUST GAS ENERGY RECOVERY APPLICATIONS

Fig. 1.

Schematic layout of generating unit for exhaust energy recovery.

paper, it is envisaged that automotive generating systems might use turbo-expanders having rated power output of about 4 kW at rated speed of 18 000 r/min. In consideration of the huge improvements achieved in permanent magnet (PM) materials in terms of both technical characteristics—such as high energy density and high operational temperature—and manufacturing costs, PM machines are being widely recognized to be eligible for use in 42-V automotive systems. Within the broad category of PM machines, several distinct machine arrangements can be identified, and these include the axial-flux PM (AFPM) machine topology which, in the last years, has drawn substantial interest concerning various applications [7]–[9]. Hence, for turbo-expanderdriven alternators being used in the 42-V generating system depicted in Fig. 1, the AFPM machine arrangement would be likely selected from the recognition of unique features such as high compactness and improved efficiency. Direct-drive arrangement should be considered in order to take advantage of avoiding the use of a gearbox along the transmission path, which greatly affects the system’s costs, reliability, and efficiency at the aforementioned high rotational speeds. However, the removal of the gearbox and the need for high compactness require that the electrical generator and the power electronic interface are both designed for operation with ac electrical quantities having high fundamental frequency. This paper deals with various configurations suitable for automotive generating systems devoted to recovering energy from exhausts. In particular, three alternative topologies for the controlled rectifier are investigated concerning the harmonic content of the generator output current which may be responsible for undesired effects such as noise and vibration on both mechanical coupling and turbo-expander blades. Concerning the development of a concept prototype of the generating system depicted in Fig. 1, this paper describes the technical solutions adopted for the AFPM generator and for the controlled rectifier. As the high fundamental-frequency output of the direct-driven AFPM generator is challenging for the electric drive control issues, therefore suitable arrangement is discussed for the control architecture to be used in the generator-rectifier system. Results taken from experiments carried out on a concept prototype of the generating system are finally presented.

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Fig. 2. Idealized representation of the PM generator—controlled rectifier generating unit. (a) Per-phase equivalent circuit. (b) Vector diagram.

II. H IGH -S PEED E LECTRIC G ENERATING U NIT The proposed high-speed electric generating unit is intended to operate within a 9000–18 000-r/min speed range with both rated power of 4 kW and overall efficiency of 90% at a 18 000-r/min rating speed. The minimum provided power output should be 500 W at 9000 r/min. At any rotational speed within the operating range, the generating unit is expected to supply a 42-V power-net architecture, with a maximum value of 48 V. According to that, a controlled rectifier with adjustable voltage gain is required to step up the PM generator three-phase output voltage against a 42-V rated voltage dc link [10]. In consideration of both the relatively low value of the alternator output voltage and the high fundamental frequency being considered in the envisaged application, it can be recognized that the synchronous inductance (i.e., Ls ) of the PM generator plays a key role as discussed in the following. The sinusoidal shaping for the PM generator phase current is considered for the discussed application. Boost-rectifier topologies with either a switching rectifier or diode rectifier followed by a dc–dc converter allow the effective regulation of the input current; as a consequence, this paper is focused on low-voltage machine solutions followed by boost-rectifier topologies. Assuming that the controlled rectifier is arranged by means of the three-phase pulsewidth-modulated (PWM) two-level voltage source inverter being operated in the regenerative mode (referred to as 2L-BR in the following) and that such a threephase boost rectifier behaves as an ideal sine wave converter to dc (i.e., the fundamental frequency ac power input is fully converted to dc power in the output), the fundamental frequency ac quantities in the alternator-rectifier system can be represented by means of the per-phase equivalent circuit shown in Fig. 2(a) and the vector diagram shown in Fig. 2(b). In such an equivalent circuit, the PM generator is also represented with an idealized form (i.e., any power loss mechanism in the alternator is neglected), and the dc-link voltage Vdc is taken into account by means of an ac voltage source that provides—through an adjustable ratio autotransformer thus used for representing the effects of the inverter modulation index ma —a phase rms voltage Uph at the alternator terminals. Hence, at any given output fundamental frequency ω and phase

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Fig. 3. Power generating unit with two-level boost-rectifier topology.

electromotive force (EMF) rms value (i.e., E0 ) set by the PM generator operation, the alternator phase rms current (i.e., Iph ) is suitably adjusted by regulating both the voltage Uph and the load angle δ, and as usual, the maximum torque per ampere condition is accomplished by having the vector of the phase current aligned with the vector of the phase EMF. Based on the schematic representation depicted in Fig. 2, it is easily found that the component of the alternator output current being responsible for the transfer of electrical power to the dc link can be written as π Pg Iph,P = Iph cos δ = √ 3 2ma Vdc

(1)

where Pg is the mechanical power that the turbo-expander delivers at the generator shaft. On the other hand, the alternator current Iph that the controlled rectifier is required to deal with is determined also by the current component Iph,Q indicated in Fig. 2, which is in quadrature with the voltage Uph and thereby is related to the exchange of reactive power between the alternator and the controlled rectifier. Simple math work yields √ 2 ma Vdc 2 ωLs Iph,P . (2) Iph,Q = Iph sin δ = π E02 Hence, the rms value of the fundamental-frequency current that has to circulate in the power switches and diodes of the controlled rectifier can be written as 2 ωLs Pg Iph = Iph,P 1 + . (3) 3 E02 From (3), it clearly appears that, for any operating condition set by the alternator input torque and speed and for a given voltage of the dc link, the lower is the alternator synchronous

inductance, the lower will be the rms value of the fundamental frequency current that circulates in the power switches and diodes of the controlled rectifier. Thereby, designing the PM generator with low value of the per-unit synchronous inductance is beneficial for the controlled rectifier in terms of reduced kilovoltampere rating and power loss. However, a low value of the synchronous inductance negatively affects the waveform of the alternator phase current as, for a given value of the switching frequency used in the controlled rectifier, the lower is the alternator synchronous inductance, the higher is the total harmonic distortion (THD) of the alternator current waveform. As a consequence, the rms value of the PM generator output current increases, and this may offset the advantages envisaged from the use of a low-inductance alternator. In other words, the use of an electrical generator having low synchronous inductance reduces the fundamental frequency component of the alternator output current while increasing the harmonic content in the same current. In order to retain the advantages resulting from a reduced value of the fundamental frequency component of the alternator output current, the power circuit arrangement used for the controlled rectifier should be appropriated. Thereby, it is useful making a comparison among the various power electronic converter topologies that could be used as power conversion interface between a turbo-expander-driven PM generator and a 42-V rated voltage dc link. To this goal, the envisaged electric drive has been suitably modeled in order to investigate, through computer simulations, three alternative topologies for the controlled rectifier, namely, the conventional 2L-BR shown in Fig. 3, the dc–dc boost converter in cascade with the diode rectifier (BOOST-DR), as depicted in Fig. 4, and the three-level neutralpoint-clamped (NPC) boost rectifier (3L-BR) shown in Fig. 5. Even though the Vienna topology is a well-known solution for


Fig. 4.

Power generating unit with dc–dc boost converter in cascade with diode rectifier topology.

Fig. 5.

Power generating unit with three-level boost-rectifier topology.

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rectification, the NPC configuration has been considered in this paper as three-level reference topology because of its widely recognized standard rule for many applications in generating units. Several manufacturers have developed packaging modules for the NPC multilevel phase leg, with the perspective of future modules based on semiconductor devices technologies also different than insulated gate bipolar transistors (IGBTs) for many applications in the field of automotive and distributed power generation. The comparison between the NPC configuration and Vienna rectifier has been deeply discussed in the literature [11] with the conclusion of substantial equivalence in total power losses. However, the different distribution of power losses among the semiconductor devices can make the NPC multilevel converter preferable, even if it shows a more complicated topology, when MOSFET devices are used because of their lower conduction losses with respect to both diodes and IGBTs. The comparison among the three alternative topologies for the controlled rectifier is carried out by considering 30 kHz as switching frequency, with this value being still congruent with the use of switching devices having 150-A rated current for low-voltage applications. For simulation purposes, a total amount of 2.7 mF is supposed as dc-link capacitance in order to assure the dc-link voltage ripple within 0.25% of the rated value for the conventional 2L-BR topology. A three-phase AFPM machine having a 4-kW rated power at a 18 000-r/min rated speed is considered with design characteristics such as the 17-V rms value of the phase EMF at a nominal output frequency of 1200 Hz and a 4-μH synchronous inductance. As usual in generating unit applications, a two-loop control architecture is envisaged by considering an outer voltage loop—which is in charge for regulating the dc-link voltage and, thereby, the 42-V battery charging/discharging operation—and an inner current loop devoted to controlling the three-phase output currents of the PM generator. The 2L-BR with sinusoidal PWM is the state-of-the-art solution in most electric drive systems. However, it is not naturally the best choice as it leads to quite high value of the phase current ripple and the power switches are operated in discontinuous conduction mode for a large fraction of the sinusoidal current period, with consequent increasing of both the rms value for the phase current and the switching stress of semiconductor devices. As a result, supplementary power loss in both the electrical generator and the controlled rectifier should be expected, unless a much higher switching frequency is utilized, thereby accepting higher power loss in the controlled rectifier due to switching. Despite the simple control structure, the BOOST-DR requires the additional boost inductor Lb to limit the current ripple. For the simulation purposes, a value of 12 μH has been considered for the boost inductor in order to reduce the peakto-peak current ripple within 15–20 A. As an additional disadvantage compared to the other two topologies being considered, the BOOST-DR does not allow vector control of the alternator phase current, so the maximum torque per ampere cannot be exploited. The use of a diode rectifier causes significant distortion of the generator current waveforms with respect to the sinusoidal shape, and as a consequence, the generator torque

contains a pulsating component having relatively high amplitude. This is a remarkable disadvantage as the presence of such a pulsating torque can significantly influence the durability and reliability of the turbo-expander/generator unit. Furthermore, the conduction power loss in the BOOST-DR is mainly related to the rectifier diodes, which show worse conduction performance with respect to low-voltage power switches as MOSFETs. The 3L-BR shows a more complex hardware and control structure, mainly due to both the number of switches and the third harmonic injection for the balancing of the dc-link capacitor middle point. However, the implementation of the control algorithm is still congruent with conventional industrial-grade digital signal processors (DSPs); moreover, future trends of the power electronics market could limit higher costs related to semiconductor devices and driving circuits. On the other hand, the use of a multilevel configuration for the controlled rectifier leads to effectively reducing the current ripple to an acceptable value, thereby allowing low values for the THD, which is an essential requirement for the desired high efficiency and to lower both mechanical vibrations and acoustic noise. The benefits resulting from the use of the 3L-BR topology in the proposed generating unit should be recognized from the simulation results shown in Figs. 6 and 7, which refer to the rated torque and speed condition for the generating unit and report waveforms of, respectively, the line-to-line voltage and the alternator phase current for each of the three controlled rectifier topologies being under comparison. It clearly appears at a glance that the 3L-BR is capable to provide reduced distortion of the current waveform even in the case of the low-inductance alternator. As a result, 3L-BR topology assures continuous conduction mode of operation for a significant part of the main frequency period even at reduced generated current (i.e., at reduced rotational speed), whereas 2L-BR would require either higher switching frequency (i.e., 60 kHz) or higher PM generator synchronous inductance to limit the discontinuous conduction mode for the same operating conditions. Both the solutions would increase power losses, switching losses in the first case, and conduction losses in both the PM generator and the power electronic converter in the second case. Results achieved from simulations in terms of peak-to-peak current ripple (ΔIpp ), current THD up to the 100th harmonic, and most significant harmonic amplitudes are listed in Table I to confirm the expected superior performance of the 3L-BR arrangement. The dependence of power factor PF and phase current total rms Iph−t values on current THD can be written as PF = Iph−t

1

· cos δ 1 + THD2 = Iph 1 + THD2 .

(4) (5)

On the basis of the values in Table I, the PF reductions of 1% and 2% are found respectively for 2L-BR and BOOST-DR with respect to 3L-BR. At the same time, the Iph−t increase leads to the PM generator stator winding loss increases of 0.35%, 1.36%, and 3.72% for the 3L-BR, 2L-BR, and BOOST-DR controlled rectifier topologies, respectively.


Fig. 6.

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Simulation results at rated speed of 18 000 r/min: Line-to-line voltage (10 V/div, 200 μs/div). (a) 2L-BR. (b) BOOST-DR. (c) 3L-BR.

The comparison of the theoretical power losses for the three investigated controlled rectifier topologies has also been accomplished. To this purpose, the top characteristics among a number of devices (i.e., MOSFETs, ultrafast diodes, and

Schottky rectifier diodes) from two of the most important manufacturers (i.e., International Rectifier and IXYS) have been selected. Data sheets of power devices in the voltage ranges of 40–75 and 80–130 V are considered respectively for 3L-BR

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Fig. 7. Simulation results at rated speed of 18 000 r/min: Phase current (50 A/div, 200 μs/div). (a) 2L-BR. (b) BOOST-DR. (c) 3L-BR.

and for 2L-BR and BOOST-DR. Results of the theoretical comparison show the slight superior performance for the 3L-BR configuration, as a consequence of the switching loss reduction and the lower on-resistance for 50-V class MOSFETs. The achieved results are resumed in the diagram of Fig. 8 for the maximum and the minimum rotational speed expected for the

PM generator. The maximum speed of 18 000 r/min is related to either the modulation index ma of 0.95 or boost duty cycle db of 0.232 as well as to the rated phase current; the minimum speed of 9000 r/min is instead related to either the modulation index of 0.5 or boost duty cycle of 0.6 as well as to 1/4 of the rated phase current.


TABLE I HARMONIC CONTENT OF PHASE CURRENT AT 4 kW AND 18 000 r/min

Fig. 8. Power loss theoretical calculation: Switch conduction losses Pcon−S , antiparallel diode conduction losses Pcon−D , switching losses Psw−S , antiparallel diode recovery losses Pr−D , clamp diode conduction losses Pcon−Dc , rectifier diode conduction losses Pcon−Dr , clamp diode recovery losses Pr−Dc , boost inductor losses PLb , and dc-link capacitor losses PC .

III. AFPM G ENERATOR P ROTOTYPE In the envisaged application for the alternator, the maximum radial dimension is somewhat constrained by the mechanical coupling with the turbo-expander. For the generating unit being considered in this paper, the design specifications have setdimensional constraints for the alternator in terms of overall outer diameter lower than 200 mm and overall axial length not exceeding 120 mm. The PM machine should have been designed to have a rated power of 4 kW at a rated speed of 18 000 r/min and with efficiency expected close to 95%. As early anticipated, the AFPM machine having the so-called “Torus” structure [7], [8] seems to be the most appropriate for meeting the design specification by recognizing that, in such a PM machine configuration, high compactness is achieved as a result of a wider air-gap surface area being made available for the electromagnetic interactions and improved efficiency results from a winding arrangement having substantially reduced length of the end windings with respect to the conventional radial flux machine. Typically, in AFPM machines, for a given value of the stator outer diameter, the axial length and the weight of the machine active parts reduce with the increase of the machine number of poles. However, for a given rated speed, the increase of the number of poles leads to an increase of the alternator output frequency, which has a significant impact on the overall performance of both the electrical generator and the controlled rectifier. Concerning the envisaged application,

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an AFPM machine having eight poles and a nominal output frequency of 1200 Hz is deemed appropriate for limiting both the machine axial length and the weight, provided that a suitable material is selected for the machine stator core in order to assure reduced power loss at reasonable costs and an appropriate solution is adopted for limiting both skin effect and eddy current in the conductors of the machine winding. As illustrated in the recent scientific literature [12], [13], in high-speed PM machines, the use of a slotless stator provides a number of advantages over the conventional arrangement of the stator with slots, and in consideration of the relatively high value of rated speed used, an AFPM generator with slotless stator arrangement is selected for the intended application with the turbo-expander. On the other hand, the “Torus” structure of the AFPM machine allows easier manufacturing of slotless winding with respect to the more conventional PM radial machine. Further than meeting the goal of having an alternator with low synchronous inductance, the slotless arrangement of the machine stator allows avoiding cogging torque and acoustic noise and eliminates the power loss that otherwise highfrequency flux pulsations due to stator slots would produce in the stator teeth and in the solid structure of the rotor disks. In addition, as the core is made of conventional nonoriented electrical steel, the slotless stator arrangement avoids punching, and the core manufacturing process does not actually expose the magnetic material to significant mechanical stresses. A design of a machine with a coreless stator [14] might be also considered, but it should be discarded due to a much higher mass of PM material usually required. In high-speed applications of PM machines, a design issue concerns the fixing of the PMs onto the rotor mechanical structure [15], [16] since the mounting of PMs onto the rotor surface by simply using the gluing technique is no more adequate whenever the rotational speed and the rotor outer diameter impose significant centrifugal forces. Thus, sleeves of nonmagnetic high-strength alloys are often used to achieve a structure that counteracts the centrifugal forces acting on the magnets, but in machines having the “radial flux” arrangement, this structure is inevitably exposed to the magnetic field being in the machine air gap. Due to the harmonic content of the air-gap field, eddy currents can be induced in the sleeve, and this generates significant heating at the surface of the PMs, resulting in further reduction in magnetic loading under operation. Bandage made of carbon fiber can even be used for magnet containment, thus significantly reducing the eddycurrent power loss issue but introducing further complexity in the manufacturing process and increasing manufacturing costs. Concerning the design issue described earlier, it is recognized that the AFPM machine topology has inherent advantages resulting from its own “axial flux” structure, because any suitable mechanical arrangement devoted to counteract the centrifugal forces acting on the magnets is not immersed in the axially directed magnetic field. Thus, in order to deal with the centrifugal forces acting on the magnets glued onto the rotor disk surface, a stainless annular sleeve can be mounted on each rotor disk as earlier described in [6]. In consideration of the severe environment conditions that are expected in the automotive applications, the machine

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TABLE II DESIGN CHARACTERISTICS OF EIGHT-POLE AFPM GENERATOR

cooling is another critical aspect to deal with since a PM machine with totally enclosed construction is required and the operating temperature of the magnets significantly influences the machine electromagnetic performance. Again, the AFPM machine topology offers distinct benefits since very effective heat removal from the machine stator can be accomplished by using the end windings placed at the outer diameter of the stator core. To this goal, the stator active parts comprising the winding and the iron-strip toroidal core are encapsulated by means of an epoxy resin which also joins the stator active parts onto an aluminum casing having suitable cooling fins. Based on the use of an epoxy resin having much higher thermal conductivity compared with conventional insulation resins, the described arrangement provides a heat flow path with low thermal resistance for the removal of the heating produced by the various power loss mechanisms in the machine stator. As a result of the high value of rotational speed, further contribution to the machine cooling should be expected from the rotor disks having surfacemounted magnets which naturally act as fans in sustaining air streams in the radial direction within the machine air gaps. Based on the various technical solutions described earlier, an AFPM machine having the electrical and geometrical design characteristics listed in Table II has been built for the purpose of testing a concept prototype of the generating unit. The cross-sectional view of the prototype is depicted in Fig. 9, whereas Fig. 10 shows a picture in which one side closing shield of the PM generator has been removed to allow a view inside the machine, with one of the rotor disks in the foreground and with the terminals of the three phases of the winding being visible in the upper left. Since the winding is fully immersed in the air-gap field as a consequence of the slotless arrangement, the three-phase winding is accomplished by means of litz-wire conductors each formed of 420 wires having 0.2-mm diameter so that the eddy

currents in the winding can be reasonably neglected and the power loss in the winding is due only to the Joule effect. At rated torque operating condition, the Joule power loss in the winding is estimated to be about 75 W with a working temperature of 130 ◦ C. Disregarding the friction in the bearings, other significant power loss mechanisms are related to eddy currents and magnetic hysteresis in the stator iron core, as well as the windage effect associated with the rotational speed of rotor disks and the stray load power loss. In order to have an acceptable amount of power loss in the iron core, a strip of soft magnetic steel having a 0.2-mm thickness is used for the stator core. With a toroidal core having a mass of about 0.62 kg, the power loss in the iron toroidal core is expected to be about 90 W. Windage power loss can be taken into account through first tentative theoretical expressions, and for the AFPM machine prototype, having 116.8 m/s as the rotor maximum peripheral speed, a rough estimation leads to about 30 W for each rotor. Further to that, stray load losses should be expected in the aluminum casing due to the leakage flux resulting from the load current flowing in the end winding [17], [18]. Such power loss mechanisms cannot be negligible in the proposed AFPM generator arrangement because of end windings being placed quite close to the aluminum casing for cooling purposes. A theoretical calculation of such stray load losses is quite elaborated, and a rough estimation usually relies on experiments. IV. T HREE -L EVEL B OOST-R ECTIFIER P ROTOTYPE For the design of the controlled rectifier, the value of the switching frequency should be selected with the purpose of limiting subharmonics in the phase current waveform, and thereby, it should be higher than 21 times the fundamental frequency if the conventional asymmetrical PWM is used. However, in electrical drives that operate with high fundamental frequency, the switching frequency may be required to be higher than usual, and this considerably affects the switching losses in the semiconductor devices. Hence, a compromise value must be selected depending on both the type of semiconductor devices and the power converter arrangement that are utilized. Following the considerations developed in Section II, a prototype of the three-level NPC boost rectifier has been accomplished as shown in Fig. 11. The purpose is to conduct experimental tests devoted to verify that such a controlled rectifier arrangement is the most suitable for minimizing the harmonic content of the current whenever operated with an AFPM generator having a value of the synchronous inductance on the order of few microhenries. In the three-level NPC converters, the dc-link voltage Vdc can be split into two equal voltage sources each having Vdc /2, and this allows the use of semiconductor devices having lower voltage rating, such as MOSFETs. However, a key problem with the selected converter topology is to control the neutral-point voltage at one half of the dc-link voltage. In fact, under certain operating conditions [19], a low-frequency voltage oscillation appears in the neutral point. Therefore, a current is drawn from the neutral point, causing one dc-link capacitor to be charged, while the other is discharged. Several methods for balancing the


Fig. 9.

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Cross-sectional view of the AFPM generator prototype.

Fig. 10. AFPM generator prototype.

Fig. 11. Prototype of three-phase three-level NPC boost rectifier.

neutral-point voltage have been proposed in the literature. To solve such an issue in the prototype of 3L-BR shown in Fig. 11, the error between the measured capacitor voltages is used as a simple controller, which modifies the neutral-point current and produces a charge balancing on the capacitor through the suitable injection of third harmonic currents.

The 3L-BR prototype has been realized based on the phaseleg concept. Each phase leg is mounted on a power printed circuit board (PCB), and it has been accomplished by means of power modules which integrate in a single three-level phaseleg package four OptiMOS 3 power MOS transistor chips IPC218N06N3 as well as conventional CAL diode chips used as both antiparallel diodes and clamp diodes. The combination of electrolytic capacitors with film capacitors is used in order to assure enough capacitance to limit the dc-link voltage ripple and, at the same time, to be consistent with the rms value of current ripple that the dc-link equivalent capacitor would be capable to deal with. Based on theoretical estimations of the power loss in the devices of the 3L-BR power circuit, at the rated operating condition, the controlled rectifier efficiency is expected to be close to 96%, as a result of the 10-W power loss in dc-link capacitors as well as the 25-W switching power loss and 125-W conduction power loss in semiconductor devices. Concerning the modulating strategy to be selected for the prototype of 3L-BR, it can be found in the literature that, particularly for power converters having the NPC topology, the phase disposition allows achieving better THD and reduced switching losses, in comparison with other types of multicarrier PWM [20]. The proposed 3L-BR is controlled by a DSP platform. The digital control platform has been designed for general-purpose applications, and it includes the 32-b Texas Instruments TMS320F28335 digital processor as well as the necessary devices to access DSP utilities. An important feature of this DSP is the enhanced pulsewidth modulator module that can generate 12 PWM signals, which can be easily used to control all the power switches of the three-phase 3L-BR. The DSP has a 16-channel 12-b analog-to-digital converter module that allows the acquisition of 16 measures. Eight of these channels are already equipped with second-order active Butterworth filters, placed on to the PCB, while the other eight channels are available on a connector and can be easily used through a suitable expansion PCB. Furthermore, the control platform includes two isolated controller area network (CAN) interfaces, a resolver-to-digital converter which is used to acquire the AFPM generator speed/position measurement and a digital-toanalog converter.

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Fig. 12. Simplified control block scheme for the generating unit.

V. C ONTROL OF THE H IGH -S PEED G ENERATING U NIT The electric drive utilizes control loops that are digitally implemented, and the synchronous frame control strategy is accomplished in recognition of its capability to regulate ac currents with high bandwidth. Fig. 12 shows the selected control architecture with the digital part and the power section of the electric drive highlighted. The voltage controller has been developed to avoid excessive voltage at the 42-V dc-link battery terminals. It acts to reduce the current injected into the storage system operating as a variable saturation block. The controller is a proportional– integral-type regulator which is adaptively tuned, according to the method discussed in [21], due to the nonlinear behavior of the system. Since a fully digital control structure is selected, an inevitable digital delay is detected in the control loop due to signal sampling and computational efforts. In electric drive applications where the ratio between the sampling frequency and the output electrical frequency is quite low, such as either high-power drive or high-speed drive, the effect of the dq-frame rotation during the delay time causes both phase and magnitude errors in the inverter output voltage [22]. The evaluation of the delay introduced by the digital implementation can be assumed equal to one sample period when the calculated duty cycles are updated at the end of the switching time, whereas the delay can be considered as half of the sample time in the double-updated sampling mode where the new dutycycle values are loaded also in the middle of the switching period. This feature is today available in industrial-grade DSP. However, the complete control algorithm has to be evaluated two times per switching period, increasing the computational effort. In the proposed application, the sampling frequency is set equal to the 3L-BR switching frequency of 30 kHz with single update mode, and the total computational time is 21 μs. Phase lag due to the digital implementation can be evaluated respectively at the minimum and the maximum operating speed of the electric drive. At 9000 r/min, the machine electrical frequency is equal to 600 Hz which yields a phase lag of 7.2◦ . At the maximum value of operating speed, the electrical frequency increases to 1200 Hz with a phase delay of 14.4◦ which causes a torque reduction of nearly 3%. This delay introduces a d-axis current component which is not detectable by the control algorithm. Furthermore, the q-axis current increases in order to have the same output power, and as a result, an increment of the power loss in both the generator and the controlled rectifier is expected.

Fig. 13.

Test rig.

Digital delay introduces also stability problems due to the reduction of the stability margins. In fact, the sample delay affects directly the whole system phase margin, changing the system dynamic behavior. The described effects can be compensated basically leading the achieved electrical flux position by the delay angle, which is related to the actual rotational speed [23]. This simple and effective method provides acceptable performances with acceptable additional computational cost. VI. E XPERIMENTAL R ESULTS In order to validate the technical solutions adopted for the turbo-expander-driven generating unit, an experimental investigation has been carried out through the test rig shown in Fig. 13. Aside from than the AFPM generator and three-level NPC converter having the design characteristics discussed earlier, the experimental setup includes a test bench motor devoted to emulate the prime-mover behavior of the turbo-expander. This motor is a liquid-cooled four-pole PM synchronous machine having a 20 000-r/min rated speed and a 40-N · m rated torque. The test rig is suitably instrumented with a torque meter, a power analyzer, and digital scopes, as schematically depicted in Fig. 14, to the goal of monitoring and recording various mechanical and electrical quantities so that the overall performance of the AFPM generator and the 3L-BR prototypes can be evaluated. The digital scope DL9140 is used for visualizing the waveforms of electrical quantities at the AFPM machine winding terminals, such as the phase current and line-to-line voltage. The four-channel power analyzer measures the mechanical power, the electrical power at the AFPM machine winding terminals,


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Fig. 14. Laboratory experimental setup.

and the dc-link electrical power. As a result, the efficiency value for both the electrical generator and controlled rectifier can be evaluated at various operating conditions. The power analyzer gives also data concerning the harmonic content of the phase current. Since the AFPM generator is equipped with K-type thermocouples being placed within the machine winding, the DL708 scope recorder is used for recording experimental traces of temperature which can provide useful information on the PM machine thermal behavior. Finally, the test rig is equipped with a control unit that, by means of the CAN bus communication layer, sets the desired operating conditions for the generating unit, detects failures in the devices under testing, and, through proper negative temperature coefficient (NTC) sensors, supervises the operating temperature inside the power modules of the controlled rectifier. Preliminary measurements have been finalized to evaluate the characteristics of the AFPM generator such as the EMF waveform and rms value, as well as the winding phase resistance and synchronous inductance of the AFPM generator. As shown in Fig. 15, the AFPM machine has very sinusoidal lineto-line EMF with a harmonic content of 0.54% only for the fifth harmonic order. At a 9000-r/min rotational speed (i.e., 600-Hz fundamental frequency), the peak of the line-to-line EMF waveform has a value of 21.1 V which is consistent with the rms value of 17.5 V expected for the phase EMF at rated speed. For the winding phase resistance, a value of 2.8 mΩ is measured at an ambient temperature of about 25 ◦ C, whereas the synchronous inductance results in a value of 4 μH. In order to provide grounds for the results discussed in Section II, the experimental investigation has been particularly addressed to verify the behavior of the 3L-BR when fed by the low-inductance AFPM generator at rated current and operated with a switching frequency of 30 kHz. Load conditions of the generating unit have been accomplished through a suitable arrangement of reconfigurable power resistors being connected

Fig. 15. AFPM generator line-to-line EMF at 9000 r/min.

at the dc link of 3L-BR. To the purpose of representing the actual modes of operation of the 3L-BR, the capacitors of the dc link have been first energized to set the dc-link voltage close to its rated value so that the controlled rectifier operates with the correct value of the modulation index. The waveform of the line-to-line voltage at the input terminals of the controlled rectifier and the waveform of the generator phase current waveforms are shown in Fig. 16, with such waveforms being referred to the load condition of the generating unit with a rated input torque of 2.2 N · m. Concerning the current waveform, a THD value of 3.80% is found by considering harmonics up to the 19th order. This gives a reason for the slightly different value if compared with respect to simulation results. The values measured for the amplitude of the most significant harmonics of the phase current are listed

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Fig. 16. Experimental traces of line-to-line voltage at 3L-BR input terminals (top trace, voltage scale: 20 V/div) and phase current (bottom trace, current scale: 50 A/div) resulting from load operation with a 2.2-N · m torque. TABLE III HARMONIC CONTENT OF THE PHASE CURRENT RESULTING FROM GENERATOR OPERATION WITH 2.2-N · m TORQUE

Fig. 18. Experimental traces of AFPM generator thermal transient resulting from load operation with 2.2-N · m torque at 9000 r/min (temperature scale: 12.5 ◦ C/div, time scale: 200 s/div).

operation with ambient temperature up to 80 ◦ C being expected for the turbo-expander generating unit and 70 ◦ C for the power electronic converter in the actual automotive environment, the overall thermal behavior demonstrates that safe operation can be accomplished with temperature values within the limits for both the generator and the power electronic converter active parts. The efficiency values of both the AFPM generator and the 3L-BR are found to be slightly lower than the design values. VII. C ONCLUSION

Fig. 17. Experimental results: Zoom of phase current (10 A/div).

in Table III, and they are very close to the ones achieved from simulation results. A zoom of the peak of the phase current is shown in Fig. 17, and a peak-to-peak current ripple of 16 A is found, which is consistent with the expected value. As mentioned, the AFPM generator is equipped with thermocouple sensors, which are positioned within the winding corresponding to the inner radius (T1 ) and medium radius (T2 ) of the stator core, respectively. In the operating conditions, for test purposes, the thermal transient was recorded, and it is shown in Fig. 18. The MOSFET power modules are equipped with NTC sensors, and the converter prototype is air cooled; it is equipped with fans which are turned on as the NTC measured temperature reaches 80 ◦ C. During the experimental tests, the measured steady-state overtemperatures are about 30 ◦ C and 25 ◦ C for the AFPM generator and the 3L-BR prototype with ambient temperature of 26 ◦ C, respectively. In consideration of the

With reference to 42-V onboard generating systems for automotive applications, this paper has described the technical solution used for a generating unit which uses a radial turbo-expander to recover energy from exhaust gases through the direct coupling with a PM generator. For the investigated application, the AFPM machine topology and the threelevel boost-rectifier configuration have been selected for the high-speed electric drive. For the rated torque operation with 1200-Hz electric frequency, an eight-pole generator assembly with litz-wire conductors for the stator winding and low-loss thin nonoriented electrical steel for the stator core is proposed. It is shown that the three-level boost-rectifier configuration is able to effectively limit the electric generator current ripple to an acceptable value, even though the PM alternator has a relatively low synchronous inductance. A low value of the THD is achieved for the alternator output current waveform, which, in fact, is an essential requirement for low mechanical vibrations and acoustic noise, as well as for high efficiency. The proposed generating unit arrangement proves to be a viable solution for improving the fuel saving on board road vehicles. ACKNOWLEDGMENT The authors would like to thank Lucchi Elettromeccanica Srl and Semikron Srl for the technical support provided for the electrical generator and power electronic devices, respectively.


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Fabio Crescimbini (M’90) received the degree in electrical engineering and the Ph.D. degree from the University of Rome “La Sapienza,” Rome, Italy, in 1982 and 1987, respectively. From 1989 to 1998, he was with the Department of Electrical Engineering, University of Rome “La Sapienza,” as the Director of the Electrical Machines and Drives Laboratory. In 1998, he joined the newly established University of Roma Tre, where he is currently a Full Professor of Power Electronics, Electrical Machines and Drives in the Department of Engineering. His research interests include newly conceived permanent-magnet machines and power electronic converter topologies for emerging applications such as electric and hybrid vehicles and electric energy systems for distributed generation and storage. Prof. Crescimbini served as a member of the Executive Board of the IEEE Industry Applications Society (IAS) from 2001 to 2004. In 2000, he served as Cochairman of the IEEE-IAS “World Conference on Industrial Applications of Electric Energy,” and in 2010, he served as Cochairman of the 2010 International Conference on Electrical Machines. He was a recipient of the IEEE-IAS Electric Machines Committee awards, including the Third Prize Paper in 2000 and the First Prize Paper in 2004.

Alessandro Lidozzi (S’06–M’08) received the Electronic Engineering degree and the Ph.D. degree from the University of Roma Tre, Rome, Italy, in 2003 and 2007, respectively. Since 2010, he has been a Researcher with the Department of Engineering, University of Roma Tre. His research interests are mainly focused on multiconverter-based applications, dc–dc power converter modeling and control, control of permanentmagnet motor drives, and control aspects for power electronics in diesel-electric generating units. Dr. Lidozzi was a recipient of a Student Award and a Travel Grant at the International Symposium on Industrial Electronics in 2004. During 2005–2006, he was a Visiting Scholar at the Center for Power Electronics Systems, Virginia Polytechnic Institute and State University, Blacksburg, VA, USA.

Giovanni Lo Calzo received the Electronic Engineering degree from the University of Roma Tre, Rome, Italy, in 2010, where he has been working toward the Ph.D. degree in the Department of Engineering since 2012. From 2010 to 2011, he was a Research Assistant with the University of Roma Tre. His research interests are mainly focused on the control and modeling of grid-tied and isolated inverters and on inverter output filter topologies.

Luca Solero (M’98) received the Electrical Engineering degree from the University of Rome “La Sapienza,” Rome, Italy, in 1994. Since 1996, he has been with the Department of Engineering, University of Roma Tre, Rome, where he currently is an Associate Professor in charge of teaching courses in the fields of power electronics and industrial electric applications. His current research interests include power electronic applications to electric and hybrid vehicles as well to distributed power and renewable energy generation units. He has authored or coauthored more than 100 published technical papers. Prof. Solero is a member of the IEEE Industrial Electronics, IEEE Industry Applications, and IEEE Power Electronics Societies.