High-Torque-Density High-Efficiency Flux-Switching PM Machine for ...

IEEE JOURNAL OF EMERGING AND SELECTED TOPICS IN POWER ELECTRONICS, VOL. 1, NO. 4, DECEMBER 2013

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High-Torque-Density High-Efficiency Flux-Switching PM Machine for Aerospace Applications Christian Sanabria-Walter, Henk Polinder, Member, IEEE, and Jan Abraham Ferreira, Fellow, IEEE

Abstract— Recently, flux-switching permanent magnet (FSPM) machines have been increasingly investigated due to its torque density and fault tolerance. Because of this, they are attractive for aerospace applications where weight, installation spaces, and safety play a critical role. Recent research has investigated a variation of the conventional FSPM machine called the C-core FSPM machine, which offers 40% greater electromagnetic torque, while using half of the magnet mass. This paper proposes a new topology called Halbach-FSPM capable of achieving an electromagnetic torque 20% higher than the C-core FSPM machine, while maintaining machine volume and phase current constant and using no more magnet material than a conventional FSPM machine. First, a physical explanation of the advantages are developed for use in all FSPM machines using Halbach arrays. An analysis and comparison of torque, power and efficiency are made. Further, magnet losses and segmentation applicable to all FSPM topologies are discussed, plus an analysis on magnet usage is given. Index Terms— Brushless, direct drive, efficiency, flux switching, halbach array, permanent magnet machines, torque density.

N OMENCLATURE As,ph Bm Bpeak Br bt D FSPM g hc Id Iq Iq,rated Jrated

Slot area/phase. Magnet flux density. Peak magnetic flux density (array’s strong side). Remanent flux density. Stator tooth width. Halbach array thickness. Flux-switching permanent magnet. Airgap length. Stator back-iron thickness. d-axis current. q-axis current. Rated current. Rated current density.

Manuscript received March 8, 2013; revised July 12, 2013; accepted August 21, 2013. Date of publication August 30, 2013; date of current version October 29, 2013. This work was supported in part by the International Conference on Electrical Machines 2012, Marseille, France, in part by EADS Innovation Works in cooperation with the Delft University of Technology, and in part by the Future Aircraft Research Project. Recommended for publication by Associate Editor Yen-Shin Lai. C. Sanabria-Walter is with EADS Innovation Works, Munich 81663, Germany (e-mail: [email protected]). H. Polinder and J. A. Ferreira are with the Delft University of Technology, Delft 2628 CD, The Netherlands (e-mail: [email protected]; [email protected]). Digital Object Identifier 10.1109/JESTPE.2013.2280183

kCu kw La Le Ld Lq Lm N n Nph Nr Ns PCu,dc PCu,ac PFe,rotor PFe,stator Pin PLoss Pmag,1 Pmag, p Pout p Ri Ro SR SRM y βpole ϕarray ϕpm λ μr η ψpm

Copper fill factor. Winding factor. Axial length of the machine. Average end-winding length. d-axis inductance. q-axis inductance. Permanent magnet thickness. Number of blocks per wavelength. Rotational speed. Number of turns per phase. Number of rotor poles. Number of stator poles. dc copper loss. ac copper loss. Rotor iron loss. Stator iron loss. Input power. Total loss. Total magnet loss without segmentation. Total magnet loss with p segments ( p>1). Output power. Number of segments per magnet. Rotor radius. Stator outer radius. Split ratio. Switched reluctance machine. Distance perpendicular to the array surface. Rotor pole width. Halbach array flux. d-position magnet flux. Halbach array wavelength. Magnet recoil permeability. Efficiency. Permanent magnet flux-linkage. I. I NTRODUCTION

A

LL-ELECTRIC flight is currently probably the most important long-term goal of the aerospace industry. Although achieving full electric flight is not yet in sight in the near future, since battery and fuel cell technologies are not yet mature enough to provide the needed energy and power densities, a reasonable intermediate step would be to consider hybrid drives [1]–[4]. For this purpose, electrical machines

2168-6777 © 2013 IEEE

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Fig. 1.


Conventional 12/10 (left) and C-core 6/13 (right) FSPM machines.

with a high torque, high power density, and suitability for fault tolerant applications are imperative. Increasingly popular in recent years, FSPM machines are considered a versatile type of fractional-slot PM machines. They comply inherently with the requirements mentioned above, since the use of concentrated windings, as seen in Fig. 1, also implies an enhanced performance related to weight and fault tolerance when compared with traditional machines with distributed windings ([6], [23]). Further features contributing to the interest in this topology are stator-mounted magnets and a robust rotor as highlighted in [7]. Up to now most literature addresses two particular FSPM machine configurations: 1) the 12 stator pole/10 rotor pole and 2) 12 stator pole/14 rotor pole conventional FSPM machines. Several researches refer to the working principle, main characteristics, and design approaches as can be seen in [11]–[16]. Thermal aspects, efficiency, and losses have also been examined in [17]–[20] and their applicability in electric and hybrid drives has been analyzed also in [21] and [22]. In addition, great attention has been given to FSPM machine variations in the search for enhancement of its characteristics: 1) multitooth FSPM machines have been designed and investigated in [24]; 2) E-core machines’ suitability for fault tolerance has been highlighted as noted in [12], [13], [25], and [26]; 3) alternative hybrid excitation methods and configurations have been proposed in [27]; and 4) even an outer-rotor FSPM machine has been proposed in [28]. A very comprehensive overview that covers most of these designs can be found in [6], and [8]–[10]. Among all these investigations the C-core FSPM machine displays features that result very attractive for machine developers and manufacturers: 40% higher torque than a conventional 12/10 FSPM machine can be achieved by a 6 stator pole/13 rotor pole configuration, therefore using only half of the magnet material (Fig. 1). In [29], [32], and [33] solid foundations for this topology were laid that have made possible the following research. This paper proposes an alternative new topology that extends the performance potential of FSPM machines making use of Halbach arrays and analyzes thoroughly its performance in terms of power, torque, and efficiency. The performance of the 6/13 C-core machine is used as a reference for comparison, since this configuration was already studied independently in literature, and presents reliable and validated results that serve as a solid starting point. To make an adequate presentation of the new architecture, this paper is structured according to the following outline.

As a first step, Section III will explain how the magnetic excitation flux from the magnets is used to generate torque in FSPM machines: the relevant equations will be examined, and in particular, how the torque and magnetic flux equations are influenced by the machine’s rotor radius. In Section IV, the flux compensation principle will be introduced which allows for a redistribution of the magnet material in the stator. In FSPM machines, the torque is determined by the compromise between the amount of magnet material and rotor size: a bigger rotor implies less magnet material in the stator and vice versa. Using the flux compensation principle presented here, the magnet material is rearranged and a bigger rotor size is possible without loss of excitation flux. Once the design basics have been explained, a C-core FSPM machine using flux compensation is designed and compared with the baseline machine presented in [29]. In Section IV main electrical parameters are compared and in Section V the efficiency of both machines is calculated. Section VI presents a cost comparison of used magnet material. Finally, conclusions are drawn in Section VII. II. T ORQUE P RODUCTION M ECHANISM IN FSPM M ACHINES It is known that despite its resemblance to SRMs and the reliance of the torque production mechanism on the machine’s double saliency, the torque in FSPM machines is predominantly composed of the electromagnetic torque resulting from the excitation flux from the magnets, first sum term in (1). The reluctance torque is very small and therefore all torque can be produced with a q-axis current [15] 3 Nr ψpm Iq + L d − L q Id Iq . 2 Taking Id = 0, (1) is simplified to T =

(1)

3 3 Nr ψpm Iq = Nr Nph ϕpm Iq kw . (2) 2 2 Equation (2) gives a well-known insight on how the magnetic flux provided by the magnet influences the torque. In FSPM machines this variable strongly depends on the machine’s SR, defined as the ratio of Ri to Ro , since it determines the amount of magnet material embedded in the stator, and therefore, the maximum flux linked by a coil as seen in Fig. 2 where the rotor is in the d-position. From [34], this dependence can also be expressed through the equation (leakage ignored for the sake of simplicity) T =

(1 − SR) Ri L a (3) SR where Bm (SR) is the operational flux density of the magnet as a function of the SR. The magnet’s flux density is written as a function of the SR, since the SR determines the magnet area, which in turn determines the magnet permeance, and as a consequence the flux density. It was already determined in [29] that a SR of 65% gives the maximum electromagnetic torque for a chosen phase current of 10 Arms (Table I). The results from this optimum give an airgap flux density of ∼2T, a value characteristic of FSPM machines. ϕpm = Bm (SR) (1 − SR) Ro L a = Bm (SR)

SANABRIA-WALTER et al.: HIGH-TORQUE-DENSITY HIGH-EFFICIENCY FLUX-SWITCHING PM MACHINE

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Fig. 2. Flux line diagram of a 6/13 FSPM machine with rotor in d-position. TABLE I M ACHINE PARAMETERS OF O PTIMIZED C-C ORE 6/13 FSPM M ACHINE [29] Fig. 3. Flux line directions within one stator piece for an unmodified machine (top) and a machine with a Halbach array for flux compensation (bottom).

The main conclusion of this section is that it is desirable to have the same magnetic flux excitation at a higher SR. This way a higher torque could be achieved for a certain machine volume, since a higher torque arm and greater airgap area in relation to the total machine radius could be realized. III. F LUX C OMPENSATION P RINCIPLE Taking the conclusion of the previous section, the objective now is to analyze how the magnetic flux enters the stator iron in FSPM machines and how the magnet material can be placed so as to obtain the same flux linkage. As a first step a baseline design is defined; a 3-phase C-core FSPM machine is proposed and optimized to deliver the maximum possible torque, and its phase back EMF, cogging torque, and torque ripple are analyzed and compared with those of a conventional FSPM machine. The main machine parameters are listed in Table I. As a second step, taking this optimized C-core FSPM machine as a reference where the SR is 65%, a hypothetical machine with 75% split ratio is proposed while keeping the rotor radius unchanged. This means the airgap geometry remains unchanged and stator tooth width, back-iron thickness, and magnet height retain the optimum values found during optimization in [29]. If the rotor outer radius is kept constant, the overall volume of the machine is reduced by 25% when the stator outer radius is reduced to 39 mm (Fig. 4). This previous measure reduces the machine’s output torque by 15% since, despite having the same electric excitation of 10 Arms , the

amount of magnet material has been reduced by 38% yielding consequently a lower toque. In Fig. 3, observing how the magnets are arranged around the stator, it can be seen that due to the alternating polarity of every magnet, for every C-shaped piece of stator material the magnet flux lines are either all going into or coming out of the magnet material attached to it. This way, the “magnetic polarity” of each C-shaped stator piece changes from one stator piece to the next one. In addition, the area on the back side of the stator is not used for any purpose and exhibits the same lamination profile as the one shown to the permanent magnets embedded between the stator pieces. These two features open the possibility for compensating the magnetic flux lost, when the size of the magnets was reduced, using Halbach arrays placed on the back side of the stator. On this location, magnet material can be placed and protected between the stator and motor housing. Alternatively, banding using epoxy, carbon, or glass fiber is also possible for magnet protection, if no housing is available, without major mechanical requirements, as the array magnets are not subject to stresses or centrifugal forces. This way, a fixture not any more complicated than that required for the rotor magnets of surface permanent magnet machines can suffice. These arrays inject flux of the corresponding polarity into each stator piece (Fig. 3). In this manner, a percentage of the flux comes from the remaining magnet mass in the stator while the rest comes from the Halbach arrays. The total flux then equals the optimum value of the initial configuration ϕpm,opt = ϕpm,new + ϕarray

(4)

where ϕpm,opt is the magnet flux from the initial optimized configuration with 65% SR, ϕpm,new is the magnet flux from the new configuration with 75% SR, and ϕarray is the flux compensated by the Halbach array. Adding Halbach arrays to the back of the structure changes the SR of the machine, as now the overall radius has to be corrected by the thickness of the array. The new machine can be seen compared with the initial design at the same scale in Fig. 4, where the Halbach array is added using the same type of material as for the embedded magnets. The new corrected

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Fig. 4. Unmodified (left) and modified (right) C-core 6/13 FSPM machines. Fig. 6. One Halbach pole compensating the flux of one stator tooth: field profile along machine periphery (d coordinate) on strong-side (top) and physical arrangement (bottom).

Fig. 5. One Halbach pole compensating the flux of two stator teeth: field profile along machine periphery (d coordinate) on strong-side (top) and physical arrangement (bottom).

SR needs can be defined as SRcorrected =

Ri Ro,new + D

(5)

where Ro,new is the reduced outer stator radius. Initially, the wavelength of the Halbach array is set equal to one third of the outer perimeter (the sum of the outer perimeter of two C-shaped stator pieces plus twice the magnet height, as seen in Fig. 5). It will be shown below that this profile does not result in the most efficient use of the magnet material. The peak magnetic flux density at the strong side of a Halbach array can be described through 2π y sin(π/N) − 2π D − 1 − e λ1 e λ1 (6) Bpeak,1 = Br π/N where λ1 the Halbach array’s wavelength. With the analytical definition of the flux density, an analytical expression for the compensation flux/stator tooth coming from the array can be found by integrating (6) over a quarter of a wavelength λ1 /4

It is important to mention that the Halbach array flux equation is only 100% accurate if there are no end effects. However, the equation used here in (8) to make clear the theoretical advantage of this configuration. The impact due to end effects is quantified with FEM calculations. Through replacing (9) in (8) and then grouping all other terms in a factor K (which is equal in both cases) and setting y = 0, the amount of flux in both cases takes the following form: (10) ϕarray,1 = K 1 − e−z −2z ϕarray,2 = K 1 − e (11) with z=

2π D λ1

(12a)

and K =

1 sin(π/N) L a λ1 Br . 2π π/N

(12b)

From (10) and (11), the difference in flux between both cases, all other variables being equal, depends solely on the exponential term. With the same array thickness D in both cases, and dividing (11) by (10) and solving for z, the following condition can be defined: 1.66ϕarray,1 ≤ ϕarray,2 ≤ 2ϕarray,1

(13a)

for 0
15 Arms ), where the torque improvement can be of 30%. Regarding the phase back EMF, its peak value also increased by 30%. This is due to the larger tooth width area, allowing for a higher flux linkage. Revisiting the parameters listed for both machines in Table II, the slot area/phase was reduced by 43%. From [19], it is known that the optimized C-core FSPM machine uses round copper wire with 1.5 mm in diameter with an effective conductor area of 1.13 mm2 for the 42 turns/ coil, yielding a copper fill factor of 60%. Using the same winding technique for the C-core HalbachFSPM machine would yield a fill factor of 93%, which is extremely high and not realistic [21]. Copper fill factor values of 65% using preformed windings with round wire have been reported [21]. Using copper strip wire in preformed windings a higher fill factor of 70% can be readily achieved with the same technique: 1.5 × 0.9-mm strip wire with an effective conductor area of 0.98 mm2 . By doing this, the decrease in effective copper area is 13%. From Fig. 12, a representation of the winding arrangement makes clear that although the fill factor


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Fig. 12. Winding arrangement using rectangular copper strip wire on a C-core 6/13 Halbach-FSPM machine.

is higher than average, the design is feasible. The drawback of a reduced slot area can be overcome by a compromise between fill factor and conductor current density. The change in this last parameter will obviously have an impact on the loss concentration and the copper losses of the machine which will be discussed in Section VI. Additional parameters such as cogging torque, back-EMF harmonic content, and inductivity, although not analyzed here, are expected to have similar values and profiles as those of a normal C-core FSPM. Since the flux linkage principle, torque production mechanism and airgap geometry remain unchanged, no significant differences should arise in the above mentioned parameters. Regarding the demagnetization characteristics, however, since the PM flux linkage was increased, resulting in an improved torque capability, this should decrease the flux weakening capability which is a common tradeoff when improvement in the constant torque region is made. More importantly, the presence of the Halbach array, has an impact on efficiency since the copper loss and magnet loss distributions and value will be different. This aspect is addressed in Section VI. Concluding, the proposed methodology has the potential to boost the output power and torque of the machine for a given current by as much as 30%, when using the principle of flux compensation. While achieving this causes a reduction of the slot area, this can be compensated by making better use of the remaining area with rectangular strip wire in a preformed winding and allowing a slightly higher current density. The results presented, although promising and a good step toward the validation of the topology’s performance, remain theoretical. Since no experimental results are available for this paper, the uncertainty characteristic of numerical methods is still present. An aspect that nonetheless helps increase the confidence in the results is the correct numerical validation of preceding results for C-core FSPM machines. This validation was done using FEM modeling principles that in turn served as cornerstone for modeling the Halbach-FSPM topology. V. E FFICIENCY This section makes a thorough loss and efficiency analysis of the C-core Halbach-FSPM machine and makes a comparison to the known C-core FSPM. Values for copper (ac and dc), iron, and magnet eddy current loss are calculated and discussed in this section.

Fig. 13. Final magnet segmentation for an optimized C-core FSPM (top left) and a C-core Halbach-FSPM machines (top right), plus total magnet eddy current loss contributions for both cases (bottom).

To find the remaining loss quantities and to establish a framework for comparison the following aspects were defined: 1) new alternative base speed of 4000 rpm, 867-Hz electrical frequency (no flux weakening), this is more representative for actuation applications and where conductor ac and iron losses are more pronounced; 2) the control scheme remains Id = 0 control; 3) copper was defined with an electrical conductivity at 80 °C of 4.7 × 107(m)−1 ; 4) magnet material was defined with an electrical conductivity of 6.25 × 105 (m)−1; 5) the average end-winding length is defined as Ro − h c + Ri 4 kCu πkCu Le = + 1− (2bt + h c ) 3 2Ns 2 (14) 6) the laminations used correspond to M250–35A steel; 7) additional parameters are defined in Table II. A short investigation on magnet eddy current loss based on the radial segmentation principle explored in [31] yields the following relationship for C-core type machines: Pmag, p,C−Core 2 ≈ Pmag,1,C−Core 3p

for p > 1.

(15)

A magnet segmentation using four pieces is found to be optimal for this type of machine allowing an 85% loss reduction. In addition, a circumferential partition of the segment nearest to the airgap yields the lowest loss concentration (Fig. 13). Using this configuration, the magnet eddy current loss is calculated for both machines, where the advantage of the Halbach-FSPM is of 20% less loss than the normal machine. Most of the loss (60%) is concentrated in the first segment, nearest to the airgap, where the strongest flux

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TABLE III C OMPARISON OF L OSS AND E FFICIENCY FOR O PTIMIZED C-C ORE 6/13 FSPM AND C-C ORE 6/13 H ALBACH -FSPM M ACHINES

excursions occur. Contrary to that, the contribution from the Halbach array is very low despite the amount of material it uses (11% of total loss). In the same manner, the iron and copper losses were calculated and the results presented in Table III. Both machines present practically the same iron loss, since the total iron mass is roughly equal. The distribution, nevertheless, is different: higher rotor losses for the C-core Halbach-FSPM since it has a higher rotor mass while the stator loss is lower in comparison with the normal C-core FSPM. Finally, the most relevant and considerable difference lies in the copper loss. Due to the smaller slot area consequence of the high SR the following phenomena take place: 1) Regarding ac copper loss: For the C-core HalbachFSPM, there is a higher penetration of magnetic flux from the airgap into a slot with a more tightly packed winding. This causes current density increases in more conductors due to proximity effect than for a normal C-core FSPM and as a consequence, the conductor eddy current loss increases also, although by 8% only. 2) Regarding dc copper loss: Since the copper fill factor was increased by 10%, not enough to compensate for the reduced slot area, the current density must be higher for the C-core Halbach-FSPM. Due to the higher effective conductor area possible with strip wire, the current density must be increased no more than 15%. Therefore, the loss/unit volume J 2 ρ increases by 32% from 1.67 to 2.2 mW/mm3 . On the other hand, since the conductor material amount was reduced by 15%, the final copper loss increase was of 15%. After having determined all the relevant loss quantities and ignoring the mechanical loss, the efficiency of both machines was calculated (Table III). Although the total loss for the C-core Halbach-FSPM machine is higher by 7.5%, its efficiency is still half a percentage point higher than for the normal C-core FSPM machine. This advantage of the new topology helps to compensate for the main disadvantage of a decreased slot area that causes higher copper losses.

Fig. 14. Magnet material used (top) and ratio of torque produced to magnet cost normalized to the conventional 12/10 machine values and for different q-axis currents (bottom) for conventional, C-core, and Halbach-FSPM machines.

VI. M AGNET U SAGE AND C OST This section will discuss the aspect that perhaps stands out the most about this design: added magnet material and efficiency of its use. This aspect acquires relevance since one of the main design drivers currently is the minimization of magnet use due to its high prices. Therefore, the goal here is to highlight the tradeoffs of power/torque density and cost that have become more important over the years. The magnet volume was calculated and normalized to the magnet volume used in the conventional 12/10 machine. This way a direct comparison can be made as well as a measurement of the utilization of the magnet material for torque production. The measures for the reference 12/10 FSPM machine can also be found in [29] and [32]. The top bar diagram in Fig. 14 shows that for the normal C-core 6/13 machine the magnet volume, and therefore mass, is less than half, 47.5% to be exact, than for the conventional 12/10 machine. On the other hand, the Halbach-FSPM machine has even less embedded material (34.5%), but the Halbach array drives the needed magnet mass up to 94% of the material used in the 12/10 machine, which is exactly twice that of the normal C-core machine. In terms of raw magnet material cost, the implementation of a Halbach-FSPM machine is therefore not any more expensive than implementing a conventional FSPM machine, although twice as expensive as a normal C-core machine. On the other hand, consulting with magnet system manufacturers, the magnet mounting cost for the embedded magnets is estimated to be approximately four times higher than for the Halbach array material. Taking this into account it is estimated that


the total cost of magnet material and its mounting for a Halbach-FSPM machine is as high as for a conventional FSPM machine, and 50% higher than for a normal C-core machine. To concisely analyze the utilization of the magnet material the following normalized quantity F can be calculated for each machine Torque/Cost Mag F= (16) Torque/Cost Mag Conv,12/10 where CostMag is the total magnet cost according to the cost structure explanation above. This quantity is then calculated for 1.5 and 3 times the nominal phase current as well as for the nominal value itself (Fig. 14). For all cases, F has a value of one for the conventional FSPM machine since this is the reference. As can be expected, the normal C-core 6/13 machine produces 2.8 torque/magnet cost units in comparison with 1.75 for the C-core Halbach-FSPM machine when operating at nominal current, i.e., 10 Arms . Clearly the normal C-core machine has an advantage here which is expected, since the amount of magnet mass is drastically lower than for the other two cases. Nevertheless, two points have to be highlighted as follows: 1) Despite the additional mass of the Halbach-FSPM machine, its absolute performance has the potential to be better for all phase current values than for the normal C-core machine. For the C-core machine a performance limit has been achieved, but with the help of the modification proposed using Halbach arrays, the advantages of the C-core topology can be extended to higher performance levels. 2) From Fig. 14, it can be seen that as the phase current is increased, the amount of torque produced per unit cost of magnet material for the normal C-core FSPM decreases at a higher rate than for the C-core HalbachFSPM, since it saturates faster. This means the HalbachFSPM operates at lower saturation points than the normal C-core machine allowing some additional overloading. This extends the application spectrum of the C-core architecture since the limit imposed by saturation becomes less of an issue. As a conclusion of this analysis, it can be stated that while a normal C-core machine constitutes a very good performance/low-cost solution, saturation limits its application range in terms of torque. The C-core Halbach-FSPM machine on the other hand, while requiring a higher initial investment in terms of magnet material, has an increased versatility and overloading capability, and as such can be considered a high-performance machine suitable for aerospace applications. Nevertheless, as already seen, the total cost of the magnet material is not higher than that of a conventional FSPM machine. This makes it possible to consider its use in applications where the initial investment is a more sensitive issue. VII. C ONCLUSION This paper has presented a new FSPM machine topology based on the use of Halbach arrays on the back of the machine.

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The topology has been named Halbach-FSPM and it uses a flux compensation principle presented in this paper to achieve higher output power than all other topologies. The highest potential of the flux compensation technique is identified in the C-core and E-core architectures, since these present the largest slot area. Furthermore, potentially suitable for all FSPM topologies as well adopting a multiphase/multiple pole design offers an interesting application scenario. Since multiphase/multipole designs tend to have a high optimum SR [30], a greater slot area is therefore provided as a result of a thinner back iron. In addition, a through efficiency analysis was made. Despite increased copper loss, the new topology can offer a higher efficiency with very little magnet eddy current loss in the Halbach array. Finally, the cost related to the magnet material needed was discussed and compared with the conventional and C-core FSPM machines. R EFERENCES [1] S. R. Durkee and A. Muetze, “Conceptual design of an electric helicopter powertrain,” in Proc. PEMD, Apr. 2010, pp. 1–6. [2] R. Glassock, J. Y. Hung, L. F. Gonzalez, and R. A. Walker, “Multimodal hybrid powerplant for unmanned aerial systems (UAS) robotics,” in Proc. Austral. Conf. Robot. Autom., Dec. 2008, pp. 1–5. [3] F. G. Harmon, A. A. Frank, and J. Chattot, “Conceptual design and simulation of a small hybrid-electric unmanned aerial vehicle,” J. Aircraft, vol. 43, no. 5, pp. 1490–1498, Sep./Oct. 2006. [4] P. Jänker, F. Hofmann, V. Kloeppel, and J. Stuhlberger, “Helicopter hybridization—The key for drastic reductions of fuel burn and emissions,” in Proc. 67th Annu. Forum Amer. Helicopter Soc., May 2011, pp. 1–13. [5] S. E. Rauch and L. J. Johnson, “Design principles of flux-switching alternators,” Power App. Syst. III. Trans. Amer. Inst. Electr. Eng., vol. 74, no. 3, pp. 1261–1268, Jan. 1955. [6] A. M. El-Refaie, “Fault-tolerant permanent magnet machines: A review,” IET Electr. Power Appl., vol. 5, no. 1, pp. 59–74, Jan. 2011. [7] A. S. Thomas, Z. Q. Zhu, and G. W. Jewell, “Comparison of flux switching and surface mounted permanent magnet generators for highspeed applications,” IET Electr. Syst. Transp., vol. 1, no. 3, pp. 111–116, Sep. 2011. [8] Z. Q. Zhu, “Switched flux permanent magnet machines—Innovation continues,” in Proc. ICEMS, Aug. 2011, pp. 1–10. [9] M. Cheng, W. Hua, J. Zhang, and W. Zhao, “Overview of statorpermanent magnet brushless machines,” IEEE Trans. Ind. Electron., vol. 58, no. 11, pp. 5087–5101, Nov. 2011. [10] A. Zulu, B. Mecrow, and M. Armstrong, “Permanent-magnet fluxswitching synchronous motor employing a segmental rotor,” IEEE Trans. Ind. Appl., vol. 48, no. 6, pp. 2259–2267, Nov./Dec. 2012. [11] Z. Q. Zhu and J. T. Chen, “Advanced flux-switching permanent magnet brushless machines,” IEEE Trans. Magn., vol. 46, no. 6, pp. 1447–1453, Jun. 2010. [12] T. Raminosoa and C. Gerada, “Design considerations for a fault-tolerant flux-switching permanent-magnet machine,” IEEE Trans. Ind. Electron., vol. 58, no. 7, pp. 2818–2825, Jul. 2011. [13] T. Raminosoa and C. Gerada, “A comparative study of permanent magnet—Synchronous and permanent magnet—Flux-switching machines for fault tolerant drive systems,” in Proc. ECCE, Sep. 2010, pp. 2471–2478. [14] J. T. Chen and Z. Q. Zhu, “Comparison of all-and alternate-poleswound-flux-switching PM machines having different stator and rotor pole numbers,” IEEE Trans. Ind. Appl., vol. 46, no. 4, pp. 1406–1415, Jul./Aug. 2010. [15] Z. Q. Zhu, Y. Pang, D. Howe, S. Iwasaki, R. Deodhar, and A. Pride, “Analysis of electromagnetic performance of flux-switching permanent magnet machines by nonlinear adaptive lumped parameter magnetic circuit model,” IEEE Trans. Magn., vol. 41, no. 11, pp. 4277–4287, Nov. 2005. [16] W. Hua, M. Cheng, Z. Q. Zhu, and D. Howe, “Analysis and optimization of back-EMF waveform of a flux-switching permanent magnet motor,” IEEE Trans. Energy Convers., vol. 23, no. 3, pp. 727–733, Sep. 2008.

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Christian Sanabria-Walter received the B.Sc. and M.Sc. degrees in electrical engineering from Los Andes University, Bogotá, Colombia, and the Hamburg University of Technology, Hamburg, Germany, in 2004 and 2007, respectively. He is currently pursuing the Ph.D. degree in cooperation with the Delft University of Technology, Delft, The Netherlands. He has been involved in research on conventional and alternative propulsion concepts for the aerospace industry, first for MTU Aero Engines and then for EADS Innovation Works, since 2007. His current research interests include the design and implementation of high torque variable speed synchronous drives for aerospace propulsion.

Henk Polinder (M’97) received the M.Sc. and Ph.D. degrees in electrical engineering from the Delft University of Technology, Delft, The Netherlands, in 1992 and 1998, respectively. He was an Assistant Professor with the Delft University of Technology, from 1996 to 2003, where he has been an Associate Professor in electrical machines and drives since 2003. His current research interests include the design aspects of electrical machines for renewable energy and mechatronic applications.

Jan Abraham Ferreira (M’88–SM’01–F’05) was born in Pretoria, South Africa. He received the B.Sc.Eng. (cum laude), M.Sc.Eng. (cum laude), and Ph.D. degrees in electrical engineering from Rand Afrikaans University, Johannesburg, South Africa. He was with the Faculty of Engineering, Rand Afrikaans University, from 1986 to 1997, where he held the Carl and Emily Fuchs Chair of Power Electronics. In 1998, he became a Professor of power electronics and electrical machines with the Delft University of Technology, Delft, The Netherlands.