Hysteresis Compensation Based on Controlled Current Pulses for ...

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Hysteresis Compensation Based on Controlled Current Pulses for Magnetoresistive Sensors Fei Xie, Roland Weiss, and Robert Weigel, Fellow, IEEE  Abstract—This paper presents a novel hysteresis compensation method for increasing the measurement accuracy of magnetoresistive (MR) sensors such as AMR (anisotropic magnetoresistance), GMR (giant magnetoresistance) and TMR (tunnel magnetoresistance) sensors. A coil consisting of one winding is processed on top of the sensor element. This configuration allows short current pulses (positive and negative) to generate a defined magnet field, which should be much stronger than the field to be measured. In this case the MR sensors can always be kept in the same magnetic loop (major loop) during the measurement cycle. By demodulating or averaging the output signal of the sensor, the influence of the sensors hysteresis can be largely reduced. A mixed signal circuit consisting of a FPGA, analog to digital converters and analog switches is used to generate the pulses. AMR, GMR and TMR current sensors are chosen as examples in the experiment. Current measurements with and without controlled magnetic field pulse are compared. A hysteresis reduction to nearly 20% of the original value, by using the novel hysteresis compensation method, is reached. Index Terms—Anisotropic magnetoresistance (AMR); giant magnetoresistance (GMR); tunnel magnetoresistance (TMR); hysteresis compensation; current measurement; major loop; digital signal conditioning; measurement accuracy.

I. INTRODUCTION

M

AGNETORESISTIVE effect based sensors have showed their widespread applications in the fields of current, voltage, field, position, angle, velocity and rotational speed measurement [1]-[10]. However, nonlinearity caused by the hysteresis effect always limits the accuracy of the MR sensors. A technology improvement on the magnetic material itself usually takes a long time and is mostly very expensive. Therefore many developments on hysteresis reduction take place in measuring technique. Yang et al. [11] for example Manuscript received November 21, 2014; revised April 17, 2015; accepted June 25, 2015. Copyright © 2015 IEEE. Personal use of this material is permitted. However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to [email protected] This work is partly supported by ENIAC project No. 296108-2. F. Xie is currently pursuing the Ph.D. at the Friedrich-AlexanderUniversity Erlangen Nuremberg in cooperation with Siemens Corporate Technology, 91058 Erlangen, Germany (e-mail: [email protected]). R. Weiss is with Sensor System Integration, Corporate Technology, Siemens AG, 91058 Erlangen Germany (e-mail: [email protected]). R. Weigel is with the Institute for Electronics Engineering, FriedrichAlexander-University Erlangen Nuremberg, 91058 Erlangen, Germany (email: [email protected]).

used a bias field to shift the working point of the sensor away from the strongest hysteresis point (zero point). Another common method is the so called “closed loop” operation of sensors [12] [13]. For “closed loop” operation a magnetic core, a compensation coil and a control circuit are used to compensate the magnetic field at the sensing element. As a result, beside other nonlinearities and temperature drift also the hysteresis is very small. Further, a flipping field based compensation method for AMR sensor is introduced by Hauser et al [14] and Caruso et al [15]. With the build-in coils positive and negative current pulses can be used to flip the magnetization of the free magnetized AMR film by 180°. In this way, the offset of the AMR Wheatstone bridge can be completely compensated. Furthermore due to the deep saturation of the sensors by the high current pulse generated magnetic field pulse, the hysteresis effect of the AMR sensor was reduced significantly. A mathematical model for linearization of the output of a GMR sensor is proposed by Jedlicska et al [16] [17] and improved by Han et al [18]. The principle is to keep track of the sensor output on a virtual model with a given start point, therefore the hysteresis can be calculated and compensated. However, all the techniques above have assignable weakness beyond their achievements. With a bias field the measurement range is reduced significantly. The closed-loop principle needs always bulky magnetic components and expensive power consuming analog electronics. The method with the flipping field suits only for AMR sensor. And the mathematical model in [16] - [18] is still without temperature compensation and thus can only work at a certain temperature. In this paper a novel hysteresis compensation method with simple system construction and low power consumption, suitable for all kind of pinned spin valve sensors, independent from the technology (AMR, GMR and TMR) is presented. A coil system e.g. consisting of only one winding is integrated on top of the magnetic sensor. With the build-in coil, very short positive and negative current pulses can be used as control pulses, to keep the magnetization of the pinned sensing layer of the MR sensor in a defined state, by using a minimum of electrical energy. Due to the deep magnetic saturation of the sensor layers during the current pulse, the sensor is always in the rising or falling magnetic major loop after the current pulse. In this way a fixed magnetic loop with fixed hysteresis is defined. Therefore the average value of the sensor output signal after a positive pulse and a negative pulse should be nearly constant. As a result, the hysteresis effect could be


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largely reduced. Analog switches controlled by a FPGA are used to generate very short positive and negative current pulses (850mA; 1µs). In this paper AMR, GMR and TMR current sensors are chosen as examples, this method can also be used in other applications, especially for MR sensor based voltage measurement [10]. The measurement principle is described in detail in Section II. Section III describes the sensor system structure. The results of the hysteresis improvement are shown in Section IV. Section V concludes the paper and gives an outlook.

voltage of the sensor are driven in negative saturation (point 4), and after the control pulse the output returns on the ascending major loop (point 5). To achieve a defined relation, which is nearly independent from hysteresis, the values of the sensor output signal after the positive and negative current pulses are averaged as shown in Fig. 2.

UBridge

US

2

3

II. MEASURE PRINCIPLE Fig.1 shows a typical magnetic hysteresis behavior of a magnetoresistive Wheatstone bridge used as current sensor. The principal inaccuracy of the nonlinear behavior is strongly related to the history of the input current and the varying magnetic “minor loops” caused by changing input current. Therefore after a certain while the relation between input and output of the sensor is difficult to predict.

1 IM

0

I

5

UBridge US

4

-US

Fig. 2. Magnetic hysteresis behavior with a fixed loop. (IM: measure current, US: bridge output saturation, red: major loop, blue: average of the major loop.)

0

I

H

control pulse measure field

HS1

HS HM

-US

tS1

t1

T

HS2

-HS tS2

Fig. 1. Typical magnetic hysteresis behavior of a magneto resistive based current sensor.

The main difficulty is not about the largeness of the hysteresis but its undefined behavior. If the relation between the input current and the output voltage can only move on one fixed loop (Fig.2), the compensation of this nonlinearity is much easier. The most stable loop is the major loop (outmost loop) and it could be achieved by driving the sensor in both positive and negative saturation. Without current pulse, the sensor bridge output can be any point on the yellow dashed line (Fig.2), which depends on its history. For instance, if the U Bridge starts at point 1, during the positive control pulse on the integrated coil, the magnetization and the output voltage of the sensor are in positive saturation (point 2), and the sensor output returns after the control pulse on the descending major loop of the shown characteristic in Fig. 2. So for a certain time after the positive control pulse on the integrated coil, the sensor output signal is equal to the value defined by the current to be measured and the descending major loop (point 3). For the negative control pulse it is exactly vice versa. The magnetization and the output

saturation field

t2

a UBridge H control pulse US 1

HS1

3

UM

5

UM1

tS1 -US

bridge output

2

t1

UM2

saturation bridge output

t2

4 tS2

T

HS2

b Fig. 3. Timing diagram control pulse based measurement. a) Timing diagram of magnetic field caused by control pulse and field to be measured (HS: saturation field; HS1, HS2: magnetic field caused by control pulse; HM: field to be measured; tS1, tS2: pulse length; t1, t2: pulse pause length). b) Timing diagram of control pulse and sensor (Wheatstone bridge) output voltage (US: saturation bridge output; UM: initial bridge output without pulse; UM1: bridge output after positive control pulse; UM2: bridge output after negative control pulse).

In Fig. 3 the timing diagrams of the magnetic field caused by the current pulse on the integrated coil, the field to be

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS measured and the sensor (Wheatstone bridge) output voltage are shown. The field to be measured HM, which is generated by the current to be measured IM, does not change between the control (current) pulses (Fig.3.a). However, it is important to note, that even without a change in the field to be measured, the sensor output voltage alters after a positive and negative pulse due to the hysteresis (Fig.3.b). As in Fig.2 described, the bridge output starts from an initial value U M, during the positive control pulse the bridge output reaches its positive saturation US. After the positive control pulse the bridge output remains on a level slightly higher (UM1) than its initial value UM due to magnetic hysteresis of the MR-sensor. For the negative control pulse it is exactly vice versa. The bridge output is driven in negative saturation -US, and after the control pulse the bridge output returns to a level slightly lower (UM2) than its initial value UM. During the pulse pause (t1 and t2) the bridge output UM1 and UM2 are stored and averaged to gain a defined and stable sensor output. Due to the magnetic feature of the MR sensor, the magnetization direction of the free layer could be changed in a few nanoseconds. This allows the current pulse to be very short, which reduces the power consumption and avoids the thermal problem, and at the same time improves the frequency response.

3

RXMR1 = RXMR - RΔ

U VCC

RXMR4 = RXMR + RΔ

controlled current pulse direction

UBr

RXMR4 = RXMR - RΔ

RXMR3 = RXMR + RΔ

measure current direction Fig. 5. Top view of the current sensor system with integrated one winding coil on a U-shaped conductor (green: MR-elements; blue: control pulse direction; yellow: one winding coil; gold: U-shaped conductor; white: measure current direction; purple: pinned magnetization orientation of (hard) layer; red: (free) layer magnetization orientation based on measure current. Magnetization orientations only suit for GMR and TMR, for AMR there is no pinned hard layer due to the barber pole structure).

coil for current pulse

control field

III. SENSOR SYSTEM STRUCTURE As mentioned a coil with one winding (yellow) is integrated in the sensor chip directly above the MR sensor elements (green) (Fig.4). Due to the small distance of about 10µm, a relative small current of about 850mA could drive the sensor into deep magnetic saturation.

measure field

MR-element

substrate

Fig. 6. Front view of the sensor system with integrated one winding coil (green: MR-element, yellow: one winding coil, blue: control field, orange: measure field).

control pulse direction

sensor supply

bridge output UBr

control pulse I/O

MR-element Fig. 4. Sensor layout with integrated coil (green: MR-element, yellow: one winding coil and bond pads, blue: control pulse current direction).

As can be seen from Fig.4 and Fig.5 the MR elements of the sensor are configured in a Wheatstone bridge. For current measurement the sensor chip is fixed on a PCB with a Ushaped conductor [8]. Both current flows in the U-turn shaped conductor and in the chip integrated coil can cause a quite similar magnetic field as indicated in Fig. 6. This typical magnetic field is causing a specific output voltage of the magnetoresistive sensor (UBr) by increasing the resistance value of the MR elements at one side and decreasing them at the other side.

IV. EXPERIMENTAL SETUP Due to resistive behavior of the integrated coil, the current pulse on the integrated coil is generated by a voltage pulse. As can be seen in Fig. 7 a dual analog switch (ADG1636 from Analog Devices) is used to connect the integrated coil to a voltage source in different directions. The ADG1636 has a transition time of about 200ns and can take a peak current pulse up to 850mA. The control inputs IN1 and IN2 of the analog switch are connected to a FPGA (Fig. 7). The FPGA is used to set the switches (S1 and S2) status between supply voltage (VS) and ground (GND). In the time slot between the control pulses (pulse pause), both switches are set to the ground (GND) and thus no current flows in the coil. For generating the positive and negative control pulses, the switch S1 and the switch S2 would be set shortly to the supply voltage (VS) consecutively. The important parameters, such as the pulse width, the switching frequency and the pulse to pause ratio, of the control pulse can be operated by the FPGA. The magnitude of the control pulse is controlled by the value of the voltage source.


VS GND measure trigger

S1A

S1 S1B

sensor

IN1

FPGA

IN2

VS GND

S2A

S2 S2B

Fig. 7. Control pulse generator circuit based on analog switch and FPGA.

A measure station for MR based current measurement is designed based on [10] [17] [19] to apply this novel hysteresis compensation method for current sensing (Fig.8). With the help of a U-formed copper conductor, the current change can be detected with the GMR sensor (Fig.5). The LabVIEW program controls the current source Stabizet D2425 via DAQ card (PCI-6052E) and controls the oven Vötsch VT7004 via RS232. The measure station allows current measurement from -150A to 150A in a temperature range of -40°C to 120°C. A LEM IT 600 fluxgate transducer with an accuracy of 0.0002% is used as the reference. For each current to be measured I M, five positive pulses and five negative pulses are generated as shown in Fig.3. oven Vötsch U-shaped VT7004 conducter PCB sensor

4 GMR and TMR sensors by using the described hysteresis compensation method. The supply voltage for the MR Wheatstone bridges is 5V. In order to generate the largest hysteresis of the MR Wheatstone bridges used as current sensor, all measurement operations were run from the minimum to the maximum current values and back. Fig.9 compares the digitized AMR sensor hysteresis with and without control pulse at different temperatures. The measurements are taken in a current range from -60A to 60A in 2A steps and a temperature range from -40°C to 100°C in 10°C steps. The hysteresis value in this case is the difference of the two measurements at 0A at each temperature. One digit corresponds in the measurement to about 10µV. In comparison to the measurement without control pulse, the hysteresis is significantly smaller and more stable with the temperature change.

pulse generator

LEM IT600

Stabizet D2425

filter and amplifier

Fig. 9. Hysteresis comparison of an AMR measurement over temperature with and without control pulse (pulses 400mA, 500ns long, pulse-pause ratio 1:200,000)

LabVIEW DAQ NI PCI-6052E

FPGA board

Fig. 8. Block diagram of the measurement station for the MR current sensor. (red: analog measurement with analog filter and amplifier; green: digital measurement with analog to digital converter PCB).

For analog measurement the sensor bridge output is filtered by a 4th order Bessel low pass filter with 20kHz cut off frequency and amplified before it is sent to the DAQ card. The average values after positive pulses and negative pulses are calculated in LabVIEW and at last stored in the PC. For digital measurement a print circuit board (PCB) is designed to convert the analog sensor bridge output and the temperature signal into digital form. Both the digital sensor signal and the temperature signal are sent directly to FPGA. After the filter process in FPGA the signals are sent to PC for the average process. V. RESULTS Analog and digital measurements are taken with AMR,

Fig.10 shows the comparison of a GMR Wheatstone bridge output characteristic with and without control pulse. The measurement is taken at 30°C from -80A to 80A in 10A steps and from -20A to 20A in 1A steps. The hysteresis without control pulse was about 4 times larger compared to the hysteresis with control pulse. Beside the improvement achieved with the control pulse, a small oscillation on the output characteristic is also observed.


5 VI. CONCLUSION AND OUTLOOK

small oscillation

Fig. 10. Total UI-characteristic of a GMR sensor bridge (top), part of the sensor characteristic (bottom) with and without control pulse (pulses 850mA, pulse-pause ratio 1:200,000).

Fig.11 compares a TMR Wheatstone bridge output characteristic with and without control pulse. The measurement is taken at 30°C from -120A to 120A in 10A steps and from -20A to 20A in 1A steps. A nearly 80% improvement of the hysteresis behavior could be clearly observed, however, a residual hysteresis remains.

By applying a novel control pulse based hysteresis compensation method to magnetoresistive sensors, the hysteresis effects of the sensors under test are significantly reduced: the hysteresis effect of the AMR sensor was reduced by more than 60% at room temperature and the residual hysteresis is stable with the changing temperature. In comparison, for MR spin valve sensors with a much larger hysteresis effect, like the shown GMR sensor and TMR sensor, a much larger improvement up to 80% was reached. Compared with the commercial methods for hysteresis reduction mentioned in section I, the control pulse based method is very compact and easy to integrate. Besides, it has very low power dissipation due to its extremely low pulse to pause ratio of about 1:200,000. Moreover, the measurement range is not affected and the method is suitable for all kinds of magnetoresistive sensors. Additionally, due to the pulse procedure no degaussing process is needed after an over current generated by a short circuit. These benefits enable a compact, low cost, and robust sensor system implementation with high accuracy and sensibility. However, the GMR and TMR measurements are still under further research due to the oscillation and the residual hysteresis. Besides the sensor hysteresis improvement at technology side, for much more sensitive GMR and TMR sensors in comparison with AMR sensors, the 850mA control pulse may not be large enough to drive them deep in saturation during the measurement. A multi winding coil design with 2 to 4 windings or a pulse generator with the ability to deliver larger current pulses may give a further improvement of the hysteresis reduction. Nevertheless, the current coil design due to its fine structure (Fig.4) does not allow a control pulse with much larger current. MR sensors with better performance and better integrated coil design are now under development. REFERENCES [1] [2]

[3]

[4]

[5]

[6]

[7]

Fig. 11. Total UI-characteristic of a TMR sensor bridge (top), part of the sensor characteristic (bottom) with and without control pulse (pulses 850mA, pulse-pause ratio 1:200,000).

[8]

A 2008 The present and the future of spintronics Thin Solid Films 517 2–5. Lenssen, K.-M.H; Adelerhof, D.J; Gassen, H.J; Kuiper, A.E.T; Somers, G.H.J; van Zon, J.B.A.D, "Robust giant magnetoresistance sensors," Sensors & Actuators: A. Physical, vol. 85, 2000, issue 1-3, p. 1-8. Medrano-Marques, N.J.; Zatorre-Navarro, G.; Celma-Pueyo, S., "A Tunable Analog Conditioning Circuit Applied to Magnetoresistive Sensors," IEEE Trans. Ind. Electron., vol.55, no.2, pp.966,969, Feb. 2008. Pelegrí, J.; Ejea, J.B.; Ramírez, D.; Freitas, P.P., "Spin-valve current sensor for industrial applications," Sensors & Actuators: A. Physical, vol. 105, 2003, issue 2, p. 132-136. Kataoka, Yasuhiro; Murayama, Shuhei; Wakiwaka, Hiroyuki; Shinoura, Osamu, "Application of GMR line sensor to detect the magnetic flux distribution for nondestructive testing.," International Journal of Applied Electromagnetics & Mechanics, vol. 15, (2001/2002), issue 1-4, p. 47. op lens ichal ip a a el ub , Jan; Tondra, Mark, "Improved GMR sensor biasing design," Sensors & Actuators: A. Physical, vol. 110, 2004, issue 1-3 (Selected Papers from Eurosensors XVI Prague, Czech Republic), p. 254-258. Reig, Càndid; Cubells-Beltrán, María-Dolores; Ramírez^Muñoz, Diego, "Magnetic Field Sensors Based on Giant Magnetoresistance (GMR) Technology: Applications in Electrical Current Sensing.," Sensors (14248220), 2009, vol. 9, issue 10, p. 7919-7942. . Weiss . attheis G. eiss. “Ad anced giant magnetoresistance technology for measurement applications.” easurement Science and Technology 24 (8), 082001, 2013.

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Lenz, J.; Edelstein, Alan S., "Magnetic sensors and their applications," IEEE Sensors J., vol.6, no.3, pp.631,649, June 2006. Xie, Fei; Weiss, Roland; Weigel, Robert, "Giant magnetoresistive based galvanically isolated voltage measurement," in Proc. IEEE Int. Workshop Appl. Meas. Power Syst. (AMPS), Sep. 2014, pp. 1–5. Xiaoguang Yang; Cunfu Xie; Yuanyuan Wang; Youhua Wang; Wenrong Yang; Guoya Dong, "Optimization Design of a Giant Magneto Resistive Effect Based Current Sensor With a Magnetic Shielding," IEEE Trans. Appl. Supercond., vol.24, no.3, pp.1,4, June 2014. Grandi, G.; Landini, M., "Magnetic-field transducer based on closedloop operation of magnetic sensors," IEEE Trans. Ind. Electron., vol.53, no.3, pp.880,885, June 2006. Xiaoguang Yang; Hang Liu; Yuanyuan Wang; Youhua Wang; Guoya Dong; Zhenghan Zhao, "A Giant Magneto Resistive (GMR) Effect Based Current Sensor With a Toroidal Magnetic Core as Flux Concentrator and Closed-Loop Configuration," IEEE Trans. Appl. Supercond., vol.24, no.3, pp.1,5, June 2014 Hauser, H.; Fulmek, P.L.; Haumer, P.; Vopalensky, M.; Ripka, P., "Flipping field and stability in anisotropic magnetoresistive sensors," Sensors & Actuators: A. Physical, vol. 106 issue 1-3 (Proceedings of the 4th European Magnetic Sensors and Actuators Conference), p. 121-125. Caruso, M.J.; Bratland, T.; Smith, C.H.; Schneider, R.; A new perspective on magnetic field sensing. Sensors (1998) vol.15, no.12, p.34-46. 22 refs. I. Jedlics a . Weiss and . Weigel “Increasing the measurement accuracy of GMR current sensors through hysteresis modeling,” in Proc. IEEE Int. Symp. Ind. Electron. (ISIE), Jun./Jul. 2008, pp. 884–889. Jedlicska, I.; Weiss, R.; Weigel, R., "Linearizing the Output Characteristic of GMR Current Sensors Through Hysteresis Modeling," IEEE Trans. Ind. Electron., vol.57, no.5, pp. 1728–1734, May 2010. Jinchi Han; Jun Hu; Yong Ouyang; Wang, S.X.; Jinliang He, "Hysteretic Modeling of Output Characteristics of Giant Magnetoresistive Current Sensors," IEEE Trans. Ind. Electron., vol.62, no.1, pp.516–524, Jan. 2015. Bluemm, C.; Weiss, R.; Weigel, R.; Brenk, D., "Correcting nonlinearity and temperature influence of sensors through B-spline modeling," in Proc. IEEE Int. Symp. Ind. Electron. (ISIE), Jul. 2010, pp. 3356–3361.

Fei Xie was born in Wuhan, China, in 1987. He received the B.S. and M.S. degrees from the University Duisburg Essen, Duisburg, Germany, in 2010 and 2012, respectively. He is currently working toward the Ph.D. degree jointly at Sensor System Integration, Corporate Technology, Siemens AG, Erlangen, Germany, and in the Institute for Electronics Engineering, FriedrichAlexander-University of Erlangen-Nuremberg, Erlangen. His research interests include magnetoresistance based current and voltage sensor systems, hysteresis compensation and calibration method for industrial mass production of the MR sensors. Mr. Xie has held presentations on the aforementioned topics at several international conferences and workshops.

Roland Weiss received his Dipl. Ing. degree from Friedrich Alexander University Erlangen, Germany, in electrical engineering in 1999. Afterwards he joined the Fraunhofer-Gesellschaft (FhG) and started his PhD work on power electronic devices on silicon carbide (SiC) in cooperation with Infineon Technologies. In 2004 he joined Siemens Corporate Research, working on current sensing and powermetering issues in power electronic systems, as well as on topics related to the application of nanometer scaled magnetoelectronic effects like GMR or TMR-Sensors and magnetic

6 nano particles. He has gained several national and international patents for inventions in his different fields of research. Currently he is working for Siemens as Senior Scientist and is responsible for research in the field of magnetic sensors.

Robert Weigel (S’88–M’89–SM’95–F’02) was born in Ebermannstadt, Germany, in 1956. He received the Dr.-Ing. and the Dr.-Ing.habil. degrees, both in electrical engineering and computer science, from the Munich University of Technology in Germany where he respectively was a Research Engineer, a Senior Research Engineer, and a Professor for RF Circuits and Systems until 1996. During 1994 to 1995 he was a Guest Professor for SAW Technology at Vienna University of Technology in Austria. From 1996 to 2002, he has been Director of the Institute for Communications and Information Engineering at the University of Linz, Austria where, in August 1999,th he co-founded the company DICE, meanwhile split into an Infineon Technologies (DICE) and an Intel (DMCE) company which are devoted to the design of RFICs and MMICs. In 2000, he has been appointed a Professor for RF Engineering at the Tongji University in Shanghai, China. Since 2002 he is Head of the Institute for Electronics Engineering at the University of Erlangen-Nuremberg, Germany. There, respectively in 2009 and in 2012, he co-founded the companies eesy-id and eesy-ic. Dr. Weigel has been engaged in research and development of microwave theory and techniques, electronic circuits and systems, and communication and sensing systems. In these fields, he has published more than 800 papers. For his work in microwave acoustics, he received the 2007 IEEE Microwave Applications Award. Dr. Weigel is a Fellow of the IEEE, an Elected Member of the German National Academy of Science and Engineering (acatech), and an Elected Member of the Senate of the German Research Foundation (DFG). He is and has been serving on numerous advisory boards of government bodies, research institutes and companies in Europe and Asia as well as on various editorial boards such as that of the Proceedings of the IEEE, and he has been editor of the Proceedings of the European Microwave Association. He has been member of numerous conference steering and technical program committees and was Technical Program Chair of several conferences such as the 2002 IEEE International Ultrasonics Symposium in Munich, Germany as well as General Chair of several conferences such as the 2013 European Microwave Week in Nuremberg, Germany. He served in many roles for the IEEE MTT-S and UFFC-S. He has been Founding Chair of the Austrian COM/MTT Joint Chapter, Region 8 MTT-S Coordinator, Distinguished Microwave Lecturer, MTT-S AdCom Member, and the 2014 MTT-S President.