An Interior Permanent Magnet Motor-Based Isolated On-Board Integrated Battery Charger for Electric Vehicles Ayman S. Abdel-Khalik1*, Ahmed Massoud2, and Shehab Ahmed3 1
Department of Electrical Engineering, Faculty of Engineering, Alexandria University, Alexandria 21544, Egypt Department of Electrical Engineering, Qatar University, Doha, Qatar 3 Department of Electrical and Computer Engineering and Texas A&M University at Qatar 23874, Doha, Qatar *
[email protected] 2
Abstract: This paper proposes an isolated on-board integrated battery charger using an Interior Permanent Magnet (IPM) machine with a 9-slot/8-pole combination or its multiples, and equipped with a non-overlapped Fractional Slot Concentrated Winding (FSCW). The proposed winding layout comprises three three-phase winding sets that are connected in such a way as to provide six motor terminals. Hence, a six-phase or two three-phase converters will be required for propulsion. Under motoring mode, the machine can be effectively regarded as a six-phase machine, which provides a high fault tolerant capability, and allows for a “limp home” mode of operation. Additionally, all MMF sub harmonics are eliminated, which significantly reduces the induced rotor eddy current losses, when compared with a conventional three-phase motor having the same slot/pole combination. In battery charging mode, the winding is reconfigured so that the machine is considered as a three-phase to six-phase rotating transformer. A 40kW IPM machine is designed and simulated under different modes of operation using 2D Finite Element Analysis (FEA) to validate the proposed concept. A small-scale prototype machine is also used for experimental validation.
1. Introduction A recent trend in plug-in Electric Vehicles (EVs) is to exploit the propulsion circuit components for battery charging instead of having a separate charging circuit with bulky inductors [1]. This can potentially save EV cost and weight. This technology is technically recognized as onboard integrated battery chargers, where the motor winding is used to replace the grid interfacing inductance, while the propulsion converter is employed as an ac/dc converter [1]. Although conventional three-phase stators can be used as an interfacing inductor during charging mode by simply opening the star point of the stator winding and connecting the phase terminals to the grid, the charging current flow yields an undesirable average torque production under this operational mode. This increases mechanical wear and introduces audible noise [2]. Different topologies for both isolated and non-isolated onboard integrated chargers have been proposed in literature based on either single-, three-, or multi-phase windings [3]– [8]. Although the former is technically preferable, as it offers a complete galvanic isolation from the grid, most provided solutions in the available literature are so far non-isolated. The common mode voltage, which appears between the battery and the ground in non-isolated topologies, may cause problems in the battery management system. Moreover, the parasitic capacitance between the EV chassis and the battery may yield significant displacement currents, which flow through the protective earth connection [3]. The simplest way to provide a galvanic isolation at a low cost without having a separate isolation transformer is to use the propulsion machine as a transformer during charging mode [1]. In the available literature, a main requirement for any non-isolated charger proposal is to avoid the electromagnetic torque production during charging mode. This is why the simplest proposed topologies have therefore been based on
single-phase charging [4]-[6]. Due to this technical challenge, some recent fast on-board integrated battery chargers have alternatively employed multiphase induction motors with a number of phases higher than three. These promising topologies not only ensure a zero-torque production under charging mode at high charging capacity, but also simplify the required system reconfiguration to switch between different operational modes. A promising topology that is based on an asymmetrical nine-phase stator winding was shown effective to facilitate both propulsion and charging modes without any hardware reconfiguration [7], [8]. Besides, increasing the number of phases also concurrently improves the machine torque density and fault tolerant capability, highly required features in EV applications, while allowing for semiconductor switches with lower power ratings since power is split among more phases [7]. Most of the available solutions provided in the state-ofthe-art for both isolated and non-isolated topologies employ motors with conventional distributed windings that are characterized by their relatively low leakage inductance. This introduces serious challenges when this technology is employed, and an additional front-end passive filter will be necessary to limit the charging current ripple magnitude to an acceptable level. Recently, Permanent Magnet (PM) machines with Fractional Slot Concentrated Windings (FSCW) have been gaining interest in practical motor designs employed in EVs [9]. One of the salient merits of FSCW is its relatively high per phase inductance [9], which improves the charging current waveform and reduces possible circulating currents. Nevertheless, this relative increase in per phase inductance is mainly due to the corresponding increase in the winding leakage inductance due to the high distortion of the MMF distribution produced by such a winding layout [10]. Although the effect of these space harmonics on torque ripple production could be minimized by the proper selection of
1
slot/pole combination [11], the resulting induced rotor eddy current losses may be intolerable. Different techniques have been proposed in the available literature to tackle this problem when FSCW is adopted [12]. This paper proposes a new winding layout suitable for a PM machine with a 9-slot/8-pole combination or its multiples [13] and equipped with a non-overlapped FSCW. Each coil around each stator slot forms a separate phase, with a total of nine phases or three three-phase winding sets. These winding sets are then connected in such a way as to provide six stator terminals. Hence, the machine can be regarded as a six-phase or dual three-phase machine. A similar connection was first introduced for high power multiphase induction machines in [14] to obtain a unity winding factor and to allow for a singlelayer winding layout. The same concept will be used in the motoring mode by connecting the six stator terminals to the six-phase propulsion inverter. In charging mode, the same winding is simply reconfigured as a three-phase to six-phase rotating transformer. Compared to the provided solutions in the literature [1], [2], [15], the proposed winding topology can offer the following advantages: • It can be regarded as an isolated on-board battery charger with simple winding reconfiguration to activate different modes. • Since an FSCW is employed, the corresponding merits offered by this winding layout will be gained, especially its high per phase inductance compared to a conventional distributed winding. • Since the machine is inherently an asymmetrical ninephase machine, all MMF sub harmonics will be cancelled [16]. This highly reduces the induced rotor eddy current losses when compared to a conventional three-phase motor having the same slot/pole combination. • The high phase order improves the machine fault tolerant capability under both propulsion and charging modes, and allows for a limp home mode. 2. Proposed Winding Configuration The proposed winding layout, using a 9-slot stator, is shown in Fig. 1(a). The main concept is to equip the 9-slot stator with an asymmetrical 9-phase winding or three threephase winding sets; where different phases are connected as shown in Fig. 1(b) to allow for six terminals only [14]. During propulsion mode, and unlike conventional sixphase stators, the current phase shift between the two threephase current sets (𝑎𝑏𝑐1 ) and (𝑎𝑏𝑐2 ) should be 400 to be similar to an asymmetrical nine-phase machine. By connecting the different stator phases as shown in Fig. 1(b), the currents of the winding set (𝑎𝑏𝑐3 ) will simply be the summation of the currents of the other winding sets. Consequently, the relation between currents in phase a, for example, for the three three-phase sets will be given by 𝐼𝑎3 = 𝐼𝑎1 + 𝐼𝑎2 = 𝐼𝑚 ∠00 + 𝐼𝑚 ∠ − 400 = 1.88𝐼𝑚 ∠ − 200
(1)
If the same number of turns per coil is used for all phases, the middle set (𝑎𝑏𝑐3 ) will produce a different ampere-turn magnitude. In order to maintain the same ampere-turn production from all phases, it has been shown in [14] that the number of turns of the winding group (𝑎𝑏𝑐3 ) should be 0.53
times the number of turns of the other winding sets. This ratio, however, can be approximated to 0.5, without a significant difference to provide a practical value for EVs motor designs, where the number of turns per coil is typically low. The rated phase voltage of (𝑎𝑏𝑐3 ) winding set will also be 0.5 times the rated voltage of the other winding sets, while the current magnitude will be double the rated line current. The required inverter phase voltage will be 1.5 times the rated phase voltages of the winding sets (𝑎𝑏𝑐1 ) or (𝑎𝑏𝑐2 ) [14]. During charging mode, the winding is reconfigured as shown in Fig. 1(c). In this case, the machine can be viewed as a three-phase to six-phase rotating transformer, with the winding set ( 𝑎𝑏𝑐3 ) considered as a three-phase primary, while ( 𝑎𝑏𝑐1 ) and ( 𝑎𝑏𝑐2 ) represent the transformer secondaries. Similar to the concept introduced in [15], the proposed system may operate at standstill, or the rotor may rotate synchronously with the grid. The former option corresponds to a high magnetizing current, hence a lower efficiency. While, the latter requires opening the clutch between the electric machine and the mechanical transmission [15]. For the latter option, the machine is rotated unloaded; hence, the developed mechanical power, which adds to the system losses in this case, can be neglected. The rotor magnet flux will induce two three-phase voltage sets across the windings (𝑎𝑏𝑐1 ) and (𝑎𝑏𝑐2 ) with a time phase shift of 400. Based on the winding turns ratio, 0.5, the magnitude of this induced secondary voltage will be 2 times the applied voltage. For the same rated winding current and frequency, the charging power in this case will be limited to one-third the propulsion power to avoid winding overloading. In practice, the battery power (correspondingly the bidirectional dc-dc converter power) of the commercial EVs is typically less than half the motor peak power [17]. Therefore, the limitation on charging power of the proposed connection may be seen as an acceptable practical constraint. To highlight the expected privileges of the proposed winding over conventional three-phase windings under propulsion mode, the MMF distributions of both winding layouts and their spectra are plotted and shown in Figs. 2(a) and 2(b). Since this is an 8-pole machine, the torque producing flux component will be the fourth harmonic order. The fifth harmonic represents the main slot harmonic, while the 1st and 2nd harmonics represent the MMF sub harmonics, which have a significant contribution to the induced rotor eddy current losses [10], [11]. It is clear from Fig. 2(b) that the sub harmonics (1st and 2nd) are approximately cancelled and the MMF distribution is generally improved. This will directly reduce the induced rotor eddy current losses and eventually improve efficiency as well be shown in the simulation results. As far as the charging mode is concerned, the MMF distribution and its spectrum under this mode are shown in Fig. 2(c). Since the machine is acting as a rotating transformer under this mode, the currents in the winding group (𝑎𝑏𝑐3 ) are reversed and can be approximately assumed to have the same magnitude as the propulsion mode. Clearly, the main torque producing component, 4th, is reduced by 35%, while the two subharmonics (1st and 2nd) are alternatively increased. The effect of the increase of these harmonics on the induced core losses and the magnet demagnetization will be considered in the following sections. 2
3. Machine Design To validate the proposed concept, a 40kW 9-slot/8-pole machine is designed with the proposed winding layout. The design is based on similar specifications of the 2004 Toyota Prius drive motor [18], [19] which employs an 8-pole IPM motor with a speed range up to 1500 rpm for base speed/maximum torque and a constant power range extending from 1500 to 6000 rpm. The line current frequency is then 100 Hz at 1500 rpm and 400 Hz at 6000 rpm. The required specifications are shown in Table 1 [20]. The machine is first designed as a basic three-phase machine using JMAG Express software to optimize the machine
dimensions and magnet volume. This initial design is then modified to the proposed design by splitting the three adjacent coils comprising each phase of the conventional three phase winding into separate phases forming three threephase sets, as shown in Fig. 1(a). The winding layout given in Fig. 1(a) can be simply converted to a conventional threephase machine by connecting adjacent coils in series (for example a1, a2, and a3 are connected to form phase a). The number of turns of the winding set (𝒂𝒃𝒄𝟑 ) is reduced to half, while its conductor cross section area is doubled. Hence, the same copper volume will be maintained. The machine dimensions and winding specifications are given in Table 2.
a b Fig. 1. Proposed 9-slot/8-pole machine. (a) Winding layout. (b) Propulsion mode. (c) Charging mode. MMF harmonic amp., AT
MMF distribution, AT
60 40 20 0 -20 -40 -60
c
0
60
120 180 240 Peripheral angle, deg
300
360
30
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10
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0
5
10 15 20 Harmonic order
25
30
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5
10 15 20 Harmonic order
25
30
0
5
10
25
30
a MMF harmonic amp., AT
MMF distribution, AT
60 40 20 0 -20 -40 -60
0
60
120 180 240 Peripheral angle, deg
300
360
30
20
10
0
b 30
20 0 -20 -40
0
60
120 180 240 Peripheral angle, deg
300
360
MMF harmonic amp, AT
MMF distribution, AT
40
20
10
0
15
20
c Fig. 2. Stator MMF distributions for a 9-slot stator and their spectra. (a) Three-phase winding propulsion mode. (b) Nine-phase six-terminal winding propulsion mode. (c) Nine-phase six-terminal winding charging mode.
3
Table 1 Design specifications for the traction motor Base speed 1500 rpm Maximum speed 6000 rpm Continuous power 40 kW Peak Power @ 1500 rpm 70 kW Continuous torque below rated speed 250 Nm Peak torque below rated speed 437 Nm Constant power speed ratio 4:1 Rated current 110 Arms Maximum current limit 190 Arms DC link Voltage 500V Table 2 Machine Dimensions and Winding Specifications. Parameter Value Number of slots 9 Stator outer diameter (mm) 300 Stack length (mm) 90 Air gap length (mm) 1 Rotor diameter (mm) 150 Teeth width (mm) 30 Stator back iron height (mm) 20 Magnet thickness (mm) 6 Magnet width (mm) 33 (NMX-43SH) Magnet material Br = 1.142 T, Hc= 564 kA/m at 150 0C Steel material B50A1000 No. of turns per coil 40 Wire size 17 AWG Number of strands 10 Winding inductances under Ld = 3.806 mH and motoring mode Lq= 4.631 mH Per phase winding resistance 0.016 Ω Equivalent machine inductance 3.6 mH under charging mode 4. Simulation Results In this section, the designed machine is simulated under different operational modes to validate the presented concept. The simulation is carried out using 2D FEA in JMAGStudio10.0. Under propulsion mode, rated conditions given in Table 1 are simulated. Under rated line current and from (1), the RMS current magnitude in the winding group (𝑎𝑏𝑐3 ) will be 1.88 × 110A = 207A , approximately double the rated line value (110A). The rated stator frequency will be 100Hz. To switch to charging mode, the machine is reconfigured as shown in Fig. 1(c), and the winding group ( 𝑎𝑏𝑐3 ) is connected to the 60Hz grid. Hence, the machine synchronous speed for this line frequency will be 900rpm. Since the machine is rotated mechanically unloaded, the corresponding mechanical power can be then neglected. Hence, the charging power delivered through winding groups (𝑎𝑏𝑐1 ) and (𝑎𝑏𝑐2 ) will represent the difference between the input grid power to (𝑎𝑏𝑐3 ) and the total machine losses. In the following subsections, the machine is simulated under both propulsion and charging modes. This is done for the healthy case and single VSC fault mode. In the latter mode, one of the three-phase VSC is assumed open due to a fault in one or more converter legs. In all simulated cases, the maximum load angle is assumed under propulsion mode,
while a zero-load angle, i.e. neglected mechanical power, is assumed under charging mode. In the FE simulations, the sixphase current controlled inverter is emulated with two threephase current sources connected to the winding terminals (𝑎𝑏𝑐1 ) and (𝑎𝑏𝑐2 ), where the currents are assumed perfectly sinusoidal. In charging mode, the current angles are adjusted to ensure unity power factor at the primary side. While the phase shift angle between the two current sets (𝐼𝑎𝑏𝑐1 and 𝐼𝑎𝑏𝑐2 ) will be always 400 at both operational modes. 4.1. Propulsion Mode Table 3 summarizes the main conclusions obtained from the FE simulation results during propulsion. For the propulsion case, results for a conventional three-phase stator are also added to show the main advantages obtained when the proposed 9-phase 6-terminal winding is adopted under this mode. Table 3. Simulation Results under Propulsion Mode at 1500 rpm 9-ph 3-ph 9-ph Mode (Single (Healthy) (Healthy) VSC) 𝑇𝑎𝑣 (Nm) 241.8 248.7 150.2 𝑇𝑝−𝑝 (Nm) 17 11 9 ∆𝑇⁄𝑇𝑎𝑣 (%) 3.5 2.25 3 Copper loss (W) 1729 1673 725 Stator core loss 1100 818 847 (W) Rotor loss (W) 653 329 503 Total loss (W) 3482 2820 2075 Output power (kW) 37.98 39.07 23.56 Input power (kW) 41.46 41.89 25.64 Efficiency (%) 91.6 93.3 91.9 From Table 3, the proposed winding offers an approximate 3% gain in the average torque for the same line current magnitude when compared with a conventional 9slot/8-pole machine with a three-phase winding. Moreover, the torque ripple percentage is reduced from 3.5% down to 2.25%. The stator copper loss is also reduced by approximately 3.3%. This reduction is due to the fact that the actual phase current of the winding group (𝑎𝑏𝑐3 ) is 1.88 times rather than double the line current, while the conductor cross sectional area was selected as the double of the other winding sets. The proposed winding also corresponds to a significant reduction in stator and rotor core losses. The reduction of the rotor core loss is even more pronounced. This notable loss reduction corresponds to a 1.7% improvement in the overall machine efficiency at rated conditions. This conclusion can be further assisted by plotting the efficiency maps calculated using the estimated machine parameters and losses for both winding layouts as shown in Figs. 3(a) and 3(b), respectively. It is clear that the proposed winding layout corresponds to a better efficiency in the whole speed range than the conventional three-phase case. As far as fault tolerance is concerned, the simulation results for a single VSC fault mode show that the machine can develop up to 60% of its rated torque for the same line current magnitude with a small reduction in the overall 4
efficiency. However, a notable increase in the induced rotor losses will take place due to the generated space harmonics when one winding set is completely disabled. Nevertheless, the induced rotor losses are still below the conventional threephase case. The corresponding developed torque waveforms for different cases are shown in Fig. 3(c). Clearly, the torque ripple is improved compared with conventional three-phase winding while the average torque is higher by approximately 3%. Under single VSC fault mode, the torque ripple magnitude can still be considered low and acceptable. 4.2. Charging Mode a
Table 4. Simulation Results Under Charging Mode at 900 rpm 9-ph 9-ph Mode (Healthy) (Single VSC) 𝑇𝑎𝑣 (Nm) 0 0 𝑇𝑝−𝑝 (Nm) 29 16 Copper loss (W) 1757 761 Stator core loss (W) 642 313 Rotor loss (W) 327 119 Total loss (kW) 2.72 1.19 Input grid power 12.82 6.5 (kW) Charging battery 10.1 5.31 power (kW) Efficiency (%) 79 81.6
b 260
Torque, Nm
240 220
9-ph. 9-ph. single VSC 3-ph.
200 180 160 140
0
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0.004 0.006 Tims, s
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c Fig. 3. Simulation results to propulsion modes. (a) Efficiency maps for a conventional three-phase stator. (b) Efficiency maps for a nine-phase six-terminal stator. (c) Torque profile for different cases.
40 9-ph. 9-ph. Single VSC 20
Torque, Nm
Table 4 summarizes the main conclusions obtained from the FE simulation results under this mode of operation, while the simulation results are shown in Fig. 4. It is clear from Fig. 4(a) that the average developed torque under charging mode is zero. This case, of course, neglects the required developed torque to overcome the mechanical loss component since zero load angle is assumed. On the other hand, the torque ripple is higher than the propulsion mode, approximately 6%, which is affected by the corresponding space harmonics appearing in the air gap under this mode. A further reduction in the torque ripple under this mode requires a special rotor optimization study, which is postponed to a future study. Interestingly, the machine can operate in charging mode using a single VSC, however, the charging level is limited to 50% of the rated charging power. The torque ripple is reduced due to the reduction in the MMF magnitude with the loss of one of the rotating transformer secondaries. The charging grid current is shown in Fig. 4(b) and 4(c) for the healthy and single VSC cases, respectively. Under charging mode, the secondary currents are controlled from the converter side to ensure a unity power factor on the primary side. Although the secondary currents are assumed sinusoidal, the grid current experiences some distortion, especially under the single VSC fault case due to core saturation and different machine asymmetries. This can be easily compensated from the secondary side by proper harmonic current compensation techniques [21], which is out of scope of this paper.
0
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4.3 Effect of Charging mode on PM Demagnetization
-150
Ig -40
2) Core loss reduction: by addressing different techniques proposed in literature to reduce the stator as well as rotor core losses [22]. Among different techniques, the stator shifting concept represents a promising recent technique to significantly suppress the effect of slot harmonics, the main contributor to eddy current losses [11]. It was also shown effective to enhance the reluctance torque component in interior PM rotor types [22]. The concept is based on increasing the coil pitch from one, in single tooth winding, to two, which dictates doubling the number of stator slots. This coil pitch represents a compromise between the single tooth winding, which offers minimum end turn volume, and distributed winding, which offers a better flux. The stator shifting is also expected to reduce the induced torque ripple under both operational modes due to the significant reduction in different space harmonics.
-300 0.03
c Fig. 4. Simulation results for charging mode. (a) Developed torque in charging mode. (b) Grid phase voltage and current under healthy charging mode (c) Grid phase voltage and current under charging mode and single VSC. It is fair to acknowledge that the efficiency of the designed machine under charging mode is relatively lower than conventional on-board chargers. This problem occurs because EV motors are commonly designed at a high current density (up to 20A/mm2 under peak output torque) to maximize the machine torque density [19], which yields a relatively high copper loss at maximum torque. Therefore, EV motors have a mandatory water and oil cooling to limit the temperature rise within acceptable limits. The rated current density of the designed machine is approximately 10.5A/ mm2. Although the efficiency under normal motoring mode was acceptable, the estimated efficiency under charging mode for the same phase current magnitude, and hence the same copper loss, was only 79% due to the fact that the maximum charging power will be only one third the machine rated power. The other main contributor to the total loss is the core loss due to the unavoidable space harmonics [22]. In order to improve the charging efficiency, the following solutions may be considered: 1) Copper loss reduction: by designing the machine at a lower maximum current density, which slightly increases the machine volume. However, since this system generally saves the required space and volume needed for the additional charging circuit, the motor volume increase will be justified. This requires solving an iterative design problem to come up with an optimum machine design, which enhances the efficiency, while minimizing the total volume of the on-board system.
This section investigates the effect of charging mode on PM demagnetization. As a rule of thumb in designing PM machines, the magnet volume is selected such as the operating point will maximize the product of flux density B (T) and magnetic field H (A/m), which is commonly called the maximum energy product BHmax [23], and is represented by the knee point of the magnet BH curve. As long as the operating point is above this point, the magnet will be able to resist the demagnetizing effect of external fields due to armature winding. Since the MMF distribution of the armature field is different under both operation modes, as depicted in Fig. 2, the effect of the charging current on PM demagnetization should be carefully checked and compared with the propulsion mode. To this endeavor, the flux density (B) and the magnetic field (H) distributions are plotted in Fig. 5 for both operational modes under rated current and worst expected thermal conditions in traction motor application, where temperature may reach 150 0C [24]. For the employed NMX43SH magnet, the residual flux density (Br) and the coercive force (Hc) at 150 0C are 1.142T and 546kA/m, respectively. By investigating the MMF distribution due to armature winding shown in Fig. 2 under charging mode, the torque producing MMF component, 4 th, which is also the main demagnetizing component, is reduced to only 35% of its value under motoring mode. Since controlling the magnitude and the angle of this MMF component will control the fundamental sequence dq current component, this significant reduction will ensure that the expected demagnetization, due to the effect of the armature reaction, is avoided. This can be supported by comparing the flux density and magnetic field distributions under both modes shown in Fig. 5. Under propulsion mode, clearly, the minimum value of the flux density (B) and the maximum value of the magnetic field (H) over the whole magnet area are above 0.35T and below 546kA/m, respectively, which is the knee point for this magnet under 150 0C. Under charging mode, the minimum value for magnet flux density increases to 0.6T, which indicates a stable operation under this mode.
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4.4 Grid Compliance of the Proposed Integrated On-board Charger As far as the high-power charging is concerned, the extent to which the charger performance will fulfill the grid regulation should be therefore investigated. Typically, there are four charging modes of electric vehicles as defined by the IEC 61851-1 standard (Mode 1 (M1), Mode 2 (M2), Mode 3 (M3), and Mode 4 (M4)) [25], [26]. Each charging mode has its own standard for the charging voltage and current requirements. M3 is concerned with high-power battery chargers, which is the main concern in the proposed system. M3 utilizes a high-power ac charging, i.e. electric vehicles are charged from an ac high power source. In M3, the IEC 610003-12 standard for a current between 16 A and 75 A per phase, and the IEC 61000-3-2 standard for a current lower than 16 A per phase can be utilized [27], [28]. The IEC 61000-3-12 current harmonic limitations and the IEC 61000-3-2 current harmonic limitations are shown in Tables 5 and 6, respectively. In Table 5, the harmonic current limitations depend on the short circuit level. Therefore, if the short circuit level is not identified, the ratio in the same table will be considered. An ac passive filter may be designed (either first order or higher) to meet the relevant standard. In the proposed system, the grid harmonic current limitations in Table 5 are adopted. Therefore, investigating the harmonic spectrum of the grid current yields, according to Table 7, a compliance with the IEC 61000-3-12 standard. As far as the ripple current may be concerned, in conventional system an efficient EMI filter is a mandatory element. However, for the designed machine with FSCW employed, the per-phase inductance is sufficiently high (3.6mH as calculated in Table 2) and the bulky filter can be then dispensed with. Another important aspect that should be carefully checked is the magnitude of the induced torque ripple under charging mode. Fig. 6 shows the torque ripple spectrum under both healthy and single VSC cases. Clearly, the fundamental torque ripple component is approximately 6.3%. Although for the current design, the torque ripple magnitude is relatively accepted, the machine design optimization under both modes of operation will help to further reduce the magnitude of the torque ripple under both modes concurrently.
4.5 Advantages and Challenges of the Proposed Integrated On-Board Charger The on-board battery charger (represented by the sixphase converter and machine windings (acting as a passive filter)) has the following pros and cons: • Weight and volume: The non-overlapped FSCW has a higher leakage inductance, which saves in the size, cost and weight of the required passive filter in charging mode, when compared with distributed winding. Using off-the-shelf three-phase VSC as the main building block enhances the power density and improves system modularity. • Controllability: The control of the VSC in the proposed system is typically achieved in a similar fashion to the control of other conventional on-board chargers. Yet, in charging mode, the six-terminal nine-phase winding needs to be reconfigured to a three-phase to six-phase rotating transformer. If operating at standstill, high magnetizing current, hence a lower efficiency, is introduced. Otherwise, the rotor may rotate synchronously with the grid (unloaded). Nonetheless, opening the clutch between the electric machine and the mechanical transmission is a necessity. • Simplicity: Extra components (mostly relays) are needed for reconfiguration of the machine, in charging mode, to 3-phase to 6-phase rotating transformer. • Reliability: A single VSC operating mode is possible in the proposed system (with a faulty VSC) which enhances the system reliability and fault ride through capability. • Limitations: The charging power is limited to one-third the propulsion power to avoid winding overloading. In charging mode, the torque ripple is slightly increased over propulsion mode. This is due to the space harmonics appearing in the air gap in the rotating transformer. Also, in a single VSC mode of operation, the machine can operate in charging mode, yet with a limited charging level (50% of the rated charging power). In charging mode, the grid current experiences some distortion, especially under the single VSC fault case due to core saturation and different machine asymmetries.
Table 5. IEC 61000-3-12 current harmonic limitations Short circuit ratio
Admissible individual harmonic current (In/I1 %)
Admissible harmonic current distortion factors THD%* PWHD%** 23 23
I3 I5 I7 I9 I11 I13 21.6 10.7 7.2 3.8 3.1 2 * THD is computed considering the first 40 current harmonics. ** Partial Weighted Harmonic Distortion (PWHD). This factor is employed to ensure that the current harmonics of order higher than 13 are reduced sufficiently 33
I3 2.3
I5 1.14 I2 1.08
Table 6. IEC 61000-3-2 current harmonic limitations Admissible individual harmonic current [A] Odd harmonics I7 I9 I11 0.77 0.4 0.33 Even harmonics I4 I6 0.43 0.3
I13 0.21
In (13
