Integrated current-time characteristic measurement

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Integrated current–time characteristic measurement system for multichannel positive temperature coefficient thermistors

This article has been downloaded from IOPscience. Please scroll down to see the full text article. 2011 Meas. Sci. Technol. 22 045903 (http://iopscience.iop.org/0957-0233/22/4/045903) View the table of contents for this issue, or go to the journal homepage for more

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IOP PUBLISHING

MEASUREMENT SCIENCE AND TECHNOLOGY

Meas. Sci. Technol. 22 (2011) 045903 (8pp)

doi:10.1088/0957-0233/22/4/045903

Integrated current–time characteristic measurement system for multichannel positive temperature coefficient thermistors Binxin Hu, Buyin Li and Ming Liu Department of Electronic Science and Technology, Huazhong University of Science and Technology, Wuhan 430074, People’s Republic of China E-mail: [email protected]

Received 11 December 2010, in final form 30 January 2011 Published 15 March 2011 Online at stacks.iop.org/MST/22/045903 Abstract The current–time characteristic plays a crucial role in assessing the actual working performance of a positive temperature coefficient (PTC) thermistor. However, the existing measurement instruments have serious limitations and hardly adapt to the actual production. This paper presents a state-of-the-art measurement system for the current–time characteristic of the PTC thermistor. Multiple updated measurement methods, including pulsed dry circuit testing, RMS digital computation, oversampling and decimation, are proposed to design this system. Several advanced techniques, specifically CAN bus, mixed signal SoC and VI, are introduced to implement this system. As a result, high accuracy class 0.1 is achieved. The current dynamic range is extended to 104 . The efficiency is increased by more than 12 times. In short, this highly integrated and cost-effective system reaches commercial grade for actual production. The practical application demonstrates that this system is quite suitable for batch production. Keywords: PTC thermistor, current–time characteristic, integrated measurement system,

fieldbus control system (Some figures in this article are in colour only in the electronic version)

current–time characteristic, which is one of the basic electrical characteristics of the PTC thermistor. Not only can it reflect the actual working performance of the PTC thermistor, but also it has some advantages such as short test period and intuitive results. It is particularly popular in actual production [4, 5]. An appropriate measurement system is vital to obtaining information about the current–time characteristic. However, it is difficult to implement an all-in-one measurement system due to multiparameters as well as multichannels. Over the years, numerous manufacturers have resorted to the conventional test scheme, where a variety of unifunctional measurement instruments, such as stabilized voltage supply, electric load, storage oscilloscope, voltmeter, ammeter, ohmmeter and stopwatch, are operated manually and they work together inefficiently and insecurely; besides, high accuracy is hard

1. Introduction The device made of donor-doped barium titanate ceramics usually exhibits a very low resistance until the device reaches a critical temperature that is referred to as its Curie point. As this point is approached, the device begins to exhibit a rising, positive temperature coefficient (PTC) of resistance as well as a large increase in resistance. The resistance change can be as much as several orders of magnitude within a temperature span of a few degrees [1, 2]. This device is known as a PTC thermistor, which is widely used in the fields such as home appliance, power facility, electronic equipment and automobile industry [3]. Accordingly, a certain amount of power applied to the PTC thermistor will result in a change of current over time. This process is called 0957-0233/11/045903+08$33.00

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to ensure. In recent years, with the industrial scale expansion, a few researchers have begun to focus on this field and some multifunction measurement instruments have been reported [6–9]. In general, most of them are based on the computer control system (CCS), where a central computer is used to program and control all functional units through one of its communication interfaces such as RS-232, EPP and PCI bus. Although they have simple structure and are easy to implement, they have serious limitations, such as normal accuracy, single range, complicated connection, weak reliability, low efficiency and high cost, thus hardly adapt to actual production. Nowadays, with the development of fieldbus technology, microelectronics, computer hardware and software, it is possible to develop an all-in-one, multichannel, high-performance, low-cost, practicable measurement system for the current–time characteristic of the PTC thermistor. This system aims at obtaining the four most critical parameters related to the current–time characteristic, namely rated zero-power resistance, action time, dissipation power and recovery time. To achieve this goal accurately, reliably, efficiently and economically, the fieldbus control system (FCS) is followed. For one thing, multiple updated measurement methods, including pulsed dry circuit testing, RMS digital computation, oversampling and decimation, are proposed to design this system. For another, this system provides several special features such as communication network based on CAN bus, intelligent node based on mixed signal SoC and display console based on a virtual instrument (VI). Thus, this system with good performance is highly integrated and costeffective. The efficiency is greatly improved. The complete system is calibrated and tested. The results show that high accuracy is achieved while the current dynamic range is greatly extended. In short, the complete system reaches commercial grade for actual production. This system has been adopted by a well-known PTC thermistor manufacturer in China. The practical application demonstrates that this system is quite suitable for batch production. Although this system is designed for the current– time (I–t) characteristic, it can be applicable to the current– voltage (I–V) characteristic by changing the VI front panel and replacing the voltage source. This is particularly beneficial for measuring some photovoltaic devices such as LEDs and solar cells.

Figure 1. Overall test scheme.

Figure 2. Pulsed dry circuit testing.

2.1. Rated zero-power resistance Firstly, each sample is placed on its fixture. Then all doublepole single-throw (DPST) switches from K1 to K10 are closed simultaneously. Thus, the rated zero-power resistance of each sample can be measured properly. To minimize the effects of device heating, test current should not be more than 10 mA. To eliminate the voltage effect, the test voltage should be limited to 2 V or less [11]. Therefore, the pulsed dry circuit testing is proposed to measure the rated zero-power resistance. Figure 2 shows its configuration. With this configuration, a precision shunt resistor (RSH ) is connected across the source terminals to clamp or limit the voltage to the sample. The dry circuit testing is operated in a pulsed current mode (as opposed to a continuous current mode) to minimize device heating during test, shorten test time and maximize throughput. To do so, a narrow current pulse generator made up of VREF and RREF is selected instead of the constant current, which is usually used in the conventional four-wire method [12]. The current pulse is forced through the sample (R) via one set of test leads called source leads, while the voltage across the sample is measured via a second set of leads called sense leads. The resistance value can be calculated as VM R= (1) VREF /RREF − VSH /RSH

2. Measurement principle and procedure

where VM is the sense voltage, and VSH is the voltage across the clamping resistor.

Figure 1 depicts the overall test scheme, where an ac voltage source is considered as a 50 Hz undistorted sine wave. All test items should be performed at a room temperature of 25 ± 2 ◦ C and the samples should operate under natural convection conditions. If the sample is tested repeatedly, the repetition rate should be sufficient to avoid thermal accumulation [10]. Take a motor starter PTC thermistor for example; the test procedure and the corresponding measurement principle can be summarized as follows.

2.2. Action time Next, the voltage source and the current limiter are configured properly and the trunk switch S0 is closed soon after. Then each branch switch between S1 and S10 is individually closed in succession. Thus, the I–t curve of each sample is gathered and the action time (ta ) is subsequently calculated, that is, the period during which each sample heats up rapidly and the 2

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Figure 3. Action time obtained from the I–t curve. Figure 4. Recovery time obtained from the ln(R)–t curve.

current falls to half of the maximum value (0.5Im ). Figure 3 shows that the action time can be obtained from the I–t curve. When power is applied to the PTC thermistor, soon after, there is a steep decline in the current near the Curie point. The current as a function of time is I = I0 e−kt

The most obvious method of computing the RMS value is to perform the functions of squaring, averaging and square rooting in a straightforward manner [14]. Nowadays, with the development of an advanced SoC MCU including the highperformance ADC and processor, it is convenient to implement RMS digital computation. Some advantages of RMS digital computation over conventional RMS convertors are fewer components, greater dynamic range, shorter response time, and generally lower cost, especially when measuring a lowfrequency signal. The algorithm of RMS digital computation can be expressed as follows:   N 1  V2 (6) Vrms =  N i=1 i

(2)

where I0 is the current at the Curie point and k is the attenuation coefficient. Since the data points of the I–t curve are discrete, the lower limit cannot be exactly located. To calculate the action time accurately, the curve fitting method is used and the attenuation coefficient (k) is obtained easily. The action time is then calculated using the following equation: 1 (3) ta = [ln I0 − ln(0.5Im )] + t0 k where Im is the maximum current, I0 is the current at the Curie point and t0 is the time at the Curie point.

where N is the number of samples and Vi is the sampled data at time i.

2.3. Dissipation power

2.4. Recovery time

Thirdly, each branch switch between S1 and S10 is closed successively, and the corresponding sample quickly reaches high resistance. Then the voltage source is set to its maximum value (Vm ) and this process lasts for 10 min. The current (Id ) flowing through each sample is measured properly. Thus, the dissipation power (W ) of each sample can be calculated as W = Vm · Id .

At the end, the channel switches from S0 to S10 are opened. Soon after, the DPST switches from K1 to K10 are closed simultaneously. The timers of each channel are started together, and the recovery time is obtained after a limited period, during which each sample cools down gradually and its resistance value falls to the double-rated zero-power resistance (2Rn ). Figure 4 shows that the recovery time can be obtained from the ln(R)–t curve. When power is removed from the PTC thermistor, the body temperature gradually decreases with the lapse of time. The body temperature (T) as a function of time (t) is

(4)

The current and the voltage are expressed as root mean square (RMS). In contrast to measuring the average value, RMS measurement is a universal language among waveforms, allowing the magnitudes of all types of voltage (or current) waveforms to be compared to one another and to dc [13]. The RMS value of the voltage is defined as   1 T [V (t)]2 dt (5) Vrms = T 0

T = (T1 − T0 ) e− τ + T0 t

(7)

where T1 is the equilibrium temperature, T0 is the room temperature and τ is the time constant. The resistance (R) is calculated as a function of the body temperature (T) as follows:

where T is the integration period and V(t) is the instantaneous voltage at the time t.

R = R0 eBT 3

(8)

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Figure 5. Configuration of the measurement system.

where R0 is the resistance at the Curie point and B is the tangent slope. From equations (7) and (8), the recovery time can accurately be calculated using the following equation: tr = τ [ln A0 − ln(ln(2Rn − Y0 ))]

Figure 6. Scheme of trunk node.

structure of the dual core processor is used to implement trunk node. The primary MCU takes the voltage measurement and the secondary MCU takes the current measurement. Both MCUs perform similar functions such as RMS digital calculation, auto range and abnormal alarm. Since trunk node works in a strong interference environment with high voltage, large current and frequent action, the two MCUs and their peripheral circuits are isolated from the test loop to strengthen the stability and the reliability. For one thing, the test voltage is attenuated to the desired range by a precision voltage divider. The divider ratio, either 0.01 or 1, is set by controlling a single-pole doublethrow (SPDT) relay. This resulting difference signal passes through an In-amp, an ISO-amp, a low pass filter (LPF), and eventually enters the secondary SoC MCU. An overload protection module, including fast Schottky diodes and large input resistors, is added to adequately protect In-amp inputs from high voltage transients while autoranging. For another, the test current flows through a shunt resistance of 0.6 . The resulting voltage drop is amplified to the desired range by a PGA, which is operated at gains of 10 and 1000. Then the amplified signal passes through an ISO-amp, a LPF, and eventually enters the primary SoC MCU. Taking the primary SoC MCU as an illustration, its builtin HVDA conditions the desired signal range to be suitable for input to on-chip 12-bit 100 ksps ADC. The output voltage of the HVDA is calculated as follows:

(9)

where A0 , Y0 and τ are constants obtained through curve fitting.

3. Implementation of the system As shown in figure 5, the system consists of the following functional units: PC node, trunk node, branch node, switch node, current limiter node, voltage source node and CAN network. All intelligent nodes communicate through the CAN network, which has distinct advantages such as real-time capability, strong load capacity, high reliability, convenient connection, multimaster protocol and low maintenance cost [15]. Considering integration and usability, C8051F040 is chosen as a core processor of trunk node and all branch nodes, which is a fully integrated mixed-signal SoC MCU with an integrated CAN 2.0 B controller. Besides, it has a high voltage difference amplifier (HVDA), which can be used to measure high differential voltages up to 60 V peak-to-peak and allow measurement of signals outside the specified ADC input range using on-chip circuitry. Switch node, current limiter node and voltage source node are classified as a relay switch. They just accept control commands. STC11F04 is chosen as a core controller, which is an enhanced 8051 MCU with powerful anti-interference performance and perfect cost performance. Without a built-in CAN controller, the UART to CAN module is used to connect each node with the CAN network. PC node serves as the display console, where the VI front panel runs. PC node is linked to the CAN network through the USB to the CAN intelligent adapter. The display console is based on VI, where a computer and functional hardware modules are integrated by a customized application program. Users can operate the computer on the friendly VI front panel to accomplish the tasks acting on the measured signal, such as data acquisition, analysis, display and storage, as if all the work was finished on a user-defined instrument.

VO = [(HVAIN+) − (HVAIN−)] · Gain + HVREF

(10)

where HVAIN+ and HVAIN– serve as the differential inputs to the HVDA and HVREF is used to provide a common mode reference for input to the ADC. The HVDA channel is configured as difference mode. HVREF is connected to 1.25 V reference voltage and HVDA Gain is set as 0.125. Thus, the ADC allows a ±1.25 V ac analog signal to be converted to a ±1024 digital signal. The noise due to quantization error can harm the signal-tonoise ratio (SNR) of A/D conversions. Because quantization error depends on the effective number of bits (ENOB) of the ADC, the best case SNR is calculated as a function of ENOB of a data conversion as follows [16]: SNR(dB) = (6.02 · ENOB) + 1.76.

(11)

To increase the ENOB, the signal is oversampled. For each additional bit of resolution, the signal must be oversampled by a factor of 4 [17]:

3.1. Trunk node Figure 6 shows the scheme of trunk node. To achieve the real-time performance of digital signal processing, a hardware

fos = 4w · fs 4

(12)

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Figure 7. Beneficial effect of oversampling.

Figure 8. Linearity error for RMS digital computation.

where w is the number of additional bits of resolution desired, fs is the original sampling frequency requirement and f os is the oversampling frequency. Without oversampling, the ENOB is only 10-bit with a range of 0–1023 through a digital rectifier, and the best SNR is 62 dB. Moreover, due to offset voltage inherent in the isolated amplifier and the HVDA, the rectified waveform is not symmetrical and the resulting signal contains serious odd harmonics, which further lowers the SNR. Considering the trade-off between resolution and throughput, the oversampling ratio is 64 and the oversampling frequency is 64 kHz. Thus, the ENOB is increased to 13-bit with a range of 0–8191, and the best SNR is increased to 80 dB. To do so, 64 consecutive ADC samples are accumulated, and the total is shifted right by 3 bits. Such an operation is commonly referred to as decimation. This results in 13 bits of useful data. This process is performed in an ADC end-of-conversion interrupt service routine. Oversampling and decimation do not affect the signal power [18]. Thus, the SNR can be improved because oversampling lowers noise power and does not affect signal power. Figure 7 shows that the low current measurement can benefit from oversampling and decimation, where the test current is 1.0 mA while the gain is 1000. The current measurement is taken using RMS digital computation. To minimize errors due to odd harmonics inherent in the rectified waveform, a digital filter is necessary to obtain the effective RMS data. To do so, an array containing 30 words acts as a data window. Once 20 consecutive useful data are collected by oversampling and decimation, they are transferred to the data window and occupy 20 low-order words. Before this process, ten low-order words are first moved to ten high-order words. Thus, the effective RMS data are calculated as follows:  +20 9    1 j 1  Erms = V2 (13) 10 j =0 21 i=j i

Figure 9. Additional error for the different crest factor.

found well within a ±0.1% error, which is maintained for different input waveforms including an undistorted sine wave, an undistorted triangle wave and a symmetrical square wave. The crest factor (CF) is defined as the ratio of the peak amplitude to the average amplitude, which usually denotes the type of waveform. As shown in figure 9, RMS digital computation provides an accurate RMS reading for input waveforms with CFs of 1–1.73. However, an average responding solution depends on the type of waveform being measured. For example, if an average responding solution is calibrated to measure the RMS value of the undistorted sine wave (CF = 1.414) and then is used to measure the symmetrical square wave (CF = 1), the average responding solution has an additional error of 11%. Likewise, if it is used to measure the undistorted triangle wave (CF = 1.73), it has an additional error of −3.8%. It is obvious that RMS digital computation can provide higher accuracy when the type of waveform changes due to voltage source fluctuation or load variation.

where Vi is the useful data at item i in the data window. As shown in figure 8, the linearity error for RMS digital computation is measured over a basic range of 100 and 5

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Table 1. Communication protocol. Node

Direction

Address

Command

Description

Trunk

Out In In Out In/out In/out Out In/out In/out In/out Out In In

0001

000 001 010 011 100 101 000 001 010 011 100 000 001

Alarm I–t start I–t stop I–t data Power Temperature Alarm Current Resistance R–t data Recovery time Alarm Control

Branch Figure 10. Scheme of branch node. Switch

3.2. Branch node Figure 10 shows the scheme of branch node, which mainly measures such parameters as residual current, resistance and recovery time. The current measurement is taken similar to that of trunk node. The gain of the In-amp is fixed at 1001 by using a 0.1%, 25 ppm, 50 , surface-mount resistor. It provides better than 13-bit accuracy over a 10 ◦ C temperature range. When the resistance measurement is performed, all test switches have been shut, and each branch node is absolutely isolated from the test loop. Thus, the resulting signal of resistance directly enters the SoC MCU. The resistance measurement is taken using pulsed dry circuit testing. While in this mode, the voltage-controlled current source (VCCS) will apply only a single, brief current pulse to the sample during the measurement cycle, thereby minimizing errors caused by device heating. The VCCS is initiated by an on-chip 12-bit DAC. The pulse width is 100 ms and pulse amplitude is 1.2 V. The source current is set as 10 mA. Thus the reference resistance (RREF ) is 120 . As shown in figure 2, the precision shunt resistor (RSH ) is also set as 120 . Then, the resistance value can be computed from the sense voltage (VM ), the voltage across clamping resistor (VSH ), the known value of RSH and the source current (using equation (1)). When measuring recovery time, the timer is started, and recovery resistance is measured once every second. To avoid faulty action, the recovery process is not considered terminated until there are three successive cases where the recovery resistance is less than the double-rated zero-power resistance.

0010–1011

1100–1110

initialize the CAN controller, the processor has to set up the bit timing register and each message object. The TxRqst bit in the transmission request register is set to start the message transmission. The TxOk bit in the status register is set to indicate a successful transmission. This bit will be reset in the interrupt service routine. Likewise, the RxOk bit in the status register is set to indicate that the new data have been received. This bit will be reset in the interrupt service routine. The processor reads the message object and obtains useful information such as all arbitration bits, the data length code and the data bytes. The bit rate is programmed to 1 Mbps. The frame format is standard frame and the frame type is data frame. The length of IDENTIFIER is 11 bits. These bits are transmitted in the order from ID.10 to ID.0. All nodes configure the acceptance filtering. Only if the byte including bits from ID.10 to ID.7 is the same as the local address is the message regarded as valid. Considering the number and stability of the node, structure of data frame can be defined as special meanings including information needed. The data frame consists of destination address (ID.10–ID.7), source address (ID.6–ID.3), command word (ID.2–ID.0), data length (DLC3–0), and data field (Data0–7). The sender is the source address and the receiver is the destination address. The IDENTIFIER defines a static message priority during bus access. The mechanism of arbitration guarantees that neither information nor time is lost. Table 1 gives an overview of the made-to-order communication protocol. PC node addressed 0000 takes priority over any other node. It can perform all commands shown in table 1, among which the alarm command has the highest priority.

3.3. CAN network The CAN network consists of a CAN interface, connecting cable and communication protocol. The SoC MCU has an integrated CAN controller; it is easy to construct the CAN interface adding an isolated CAN transceiver. The ADμm1201 are used to substitute traditional optoelectronic isolators with a few discrete components, which are dual-channel, digital iCoupler isolators. Thus, the design of the isolation circuit is simplified and the data communication gets more reliable and stable [19]. The CAN driver includes CAN initialization, CAN message transmission and CAN message reception [20]. To

3.4. PC node PC node serves as the display console, where the VI front panel runs. Since CAN communication between PC node and any other node is frequent but not large bulk, the USB to the CAN intelligent adapter is used to link them, which provides powerful interface function library. It is easy to accomplish further development of the VI front panel. Programming the VI front panel is based on MFC characterized by message-based, event-driven and multithreading. When the application is started, the 6

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Table 2. Results of static calibration. Item Resistance

Range

2 k 1 m 20  100 m

AC Voltage

6V 600 V

AC Current

100 mA 10 A

Table 3. Results of the dynamic test.

Resolution Accuracy Uncertainty (k = 2) 0.04% 0.03%

0.06%

1 mV 100 mV

0.06% 0.08%

0.10%

10 μA 1 mA

0.09% 0.06%

0.13%

Voltage Initial Action Maximum Recovery Item source (V) current (A) time (s) voltage (V) time (s) 1 2 3 4 5 6 7 8

initialization routine will be executed such as the user interface, CAN interface, system variables. Then the thread of CAN communication is enabled, and in the CAN message handling function, some important task will be performed such as data reception, data processing and data flag setting. When login succeeds, the configuration will be performed, which mainly consists of sample type, test item, test channel and test condition. When the configuration is confirmed, the test will be started. Then the thread of OnTimer is enabled and in the OnTimer message handling function, control command will be sent to other nodes in the manner of state machine. According to the test flag, corresponding useful results, such as measurement results, I–t curve and test information, will be displayed at the man–machine interface.

4.662 4.704 4.705 4.760 4.712 4.753 4.712 4.701

0.60 0.59 0.59 0.58 0.59 0.59 0.59 0.59

347.2 350.9 349.0 350.2 350.6 349.4 347.7 348.3

91.0 91.5 91.0 91.5 91.5 91.5 91.0 91.0

follow Gaussian distributions, k is 2 at 95% confidence level. u(X) and u(N) are uncertainty components. They are from the repeatability error and the calibrator error, respectively. Since u(N) is much smaller than u(X), Uref mainly depends on u(X), which can be obtained by ten times repeated measurements and subsequent computation of the experimental standard deviation. 4.2. Dynamic test The dynamic test is performed to examine the repeatability of measuring dynamic parameters such as action time and recovery time. The QP2–15 motor starter PTC thermistor is chosen as a sample. The test is performed every other hour and eight times in all. The test conditions include the room temperature of 25 ◦ C, current limiter of 30 , test voltage of 220 V, maximum voltage of 350 V and test time of 10 min. Table 3 shows the results of the dynamic test. It can be seen from table 3 that the deviation of action time is caused by fluctuant initial current partly due to voltage source drift. The action time gets longer with the initial current decreasing. On the other hand, the small difference of maximum voltage slightly influences the deviation of the recovery time. Although these deviations are within the accepted error range, it is suggested that the voltage source should be kept stable enough to minimize the deviations. The repeatability is expressed as the experimental standard deviation or the Bessel formula. In this way, the repeatability of the action time measurement is 0.005 s, while the desired accuracy is ±0.02 s. Likewise, the repeatability of the recovery time measurement is 0.1 s, while the desired accuracy is ±1.0 s.

4. Results 4.1. Static calibration The Fluke 5700A/5720A multifunction calibrator is used to implement the static calibration. The ambient temperature is within 25 ± 2 ◦ C. The calibrated parameters include resistance, ac voltage and ac current. Table 2 shows the results of static calibration. The accuracy is expressed as the fiducial error [21]. It is obvious that the accuracy class of any functional unit reaches 0.1 while the current dynamic range is extended to 104 . The measurement uncertainties of all items are also contained in table 2. They are expressed as the related expanded uncertainty [22]. To evaluate measurement uncertainties of all items, several typical values are selected from the basic range of these items. They are 10 , 2 V (50 Hz) and 1 A (50 Hz). The direct comparison method is put into practice. The Fluke 5700A/5720A still serves as the calibrator. The general mathematical model is shown as follows: =X−N

218.3 220.1 219.3 220.3 219.7 220.1 219.2 219.0

4.3. Practical application

(14)

This system has been adopted by Xiamen Hollyland Electronics Ltd, which is a well-known PTC thermistor manufacturer in China. The practical application demonstrates that this system is particularly suitable for batch production. Take the motor starter PTC thermistor, for example, under normal conditions; it takes about 13 min to test one sample while using a single channel. If the test continues for 8 h one day, there are not more than 30 samples finished. However, due to saving time of replacing samples and frequent starting, it takes about 12 min to test a batch of samples while using ten

where  is the measurement error, X is the measured value and N is the standard value. The measurement uncertainty of each item depends on the uncertainties of X and N. They are independent of one another. The related expanded uncertainty (Uref ) is defined as  k · [c1 u(X)]2 + [c2 u(N )]2 Uref = (15) N where c1 and c2 are the sensitivity factors. They are equal to 1 and −1 respectively. k is the coverage factor. Since the results 7

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channels. If the test continues for 8 h one day, there are not less than 400 samples finished. Thus, the efficiency has been increased by more than 12 times.

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5. Conclusions A state-of-the-art measurement system for the current–time characteristic of the PTC thermistor has been introduced. In contrast to reported multifunction measurement instruments, this system has better accuracy, larger dynamic range, easier connection, stronger reliability, higher efficiency and lower cost. Pulsed dry circuit testing is proposed to ensure the zeropower condition of the resistance measurement, which is difficult for the conventional four-wire method. Oversampling and decimation are done to lower the in-band noise by 18 dB, and increase the resolution of the measurement by 3 bits. RMS digital computation is put into practice. The linearity error is measured within ±0.1% over a basic range of 100. The additional error is confirmed within ±1% for input waveforms with crest factors of 1–1.73. Compared with the complex dedicated IC solution, the desired results are obtained at a lower cost. This system derives from the FCS, where CAN bus is introduced to construct the communication network, thereby extending ten branch nodes. Mixed signal SoC is used to build the intelligent node and all-in-one functional units come true. The display console is based on VI, providing a friendly man–machine interaction for users to control the instrument. The results show that high accuracy class 0.1 is achieved while the current dynamic range is extended to 104 . The efficiency is increased by more than 12 times. Moreover, if more instrumentation is connected through CAN bus hub, more channels can be extended and efficiency will be doubled and redoubled. In short, this highly integrated and cost-effective system reaches commercial grade for actual production. Although this system is applied to the PTC thermistor, it can be extended to some photovoltaic devices such as LEDs and solar cells by changing the VI front panel and replacing the voltage source. Furthermore, the proposed methods can also be applied to any other measurement instruments with multiple intelligent nodes.

Acknowledgments This work was partly supported by the National Natural Science Foundation of China under grant no 60871017/F010612. The authors thank H Wang and M Zhang, Hubei Optoelectronics Test Center, who provided reliable calibration service.

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