Microstrip-to-Waveguide Transition using Waveguide ... - IEEE Xplore

Proceedings of 2010 IEEE International Conference on Ultra-Wideband (ICUWB2010)

Microstrip-to-Waveguide Transition using Waveguide with Large Broad-Wall in Millimeter-Wave Band Kazuyuki Seo

Kunio Sakakibara

Nippon Piller Packing Co., Ltd. 541-1,Utuba,Shimouchigami Sanda, 669-1333, Japan Abstract- Broad-wall connected microstrip-to-waveguide transition using a waveguide with large broad-walls is proposed. Maximum width of the waveguide where higher order mode dose not propagate is used. The dimension of the rectangular patch element is examined to have optimum width for wideband. Moreover, the distance from the edge of the broad-wall of the waveguide to via holes is also examined to have optimum length for wideband. A numerical investigation of a transition shows appropriate parameters. Proposed transition has insertion loss less than 0.28 dB from 77 GHz to 81 GHz, and a bandwidth of 18.2 % (14.4GHz) for reflection coefficient below −15 dB.

I.

INTRODUCTION

Many kinds of millimeter-wave automotive radars have been developed [1],[2]. Microstrip antennas become good candidates when radar sensors are being widely used in vehicle due to its advantages of low cost and low profile. Generally, microstrip antennas are placed on the front surface of a radar sensor and are connected to millimeter-wave circuits inside of the sensor via waveguides. Therefore, transitions from waveguide to microstrip line are required. Various types of millimeter-wave transitions from waveguide to microstrip line have been proposed. The ridge waveguide type [3], quasi-Yagi type [4], and planar waveguide type [5] have been studied as longitudinal connection of waveguide with microstrip line. With regard to perpendicular transitions, a conventional type of probe feeding has a wideband characteristic [6]-[8], but it needs a metal short block with a quarter-wavelength on the substrate. The replacement of the metal short block is a patch element in the waveguide to achieve sufficient coupling between waveguide and microstrip line. The slot coupling type [9] achieves coupling between the microstrip line and the patch element in the waveguide by means of a slot. It is composed of two dielectric substrates without a metal short block. The proximity coupling type [10] has been developed more recently. It can be composed of a single dielectric substrate attached to the waveguide end. A rectangular patch element on the lower plane of the dielectric substrate couples with a microstrip line on the upper plane of the dielectric substrate. It is suitable for mass production. The proximity coupling type has been further developed for wideband [11]. For 79 GHz UWB automotive radar applications, 4 GHz bandwidth is required. The proximity coupling type [11] has

978-1-4244-5306-1/10/$26.00 ©2010 IEEE

Nobuyoshi Kikuma

Nagoya Institute of Technology Gokiso-cho, Shouwa-ku Nagoya, 466-8555, Japan Microstrip line

Probe

Waveguide short

Via holes

Substrate with metal pattern

Substrate Rectangular patch element Surrounding ground

Waveguide with large broad-wall

Fig. 1. Configuration of the proposed transition.

bandwidth of 6.9 % (5.29 GHz) for the reflection coefficient below −15 dB. Considering the tolerance for the manufacturing accuracy, much wider bandwidth is required. In this paper, a microstrip-to-waveguide transition using a waveguide with a large broad-wall is proposed to extend a bandwidth. The transition can be composed of a single dielectric substrate attached to the waveguide end. The configuration of the transition is presented in next Section. II. MICROSTRIP-TO-WAVEGUIDE TRANSITION USING WAVEGUIDE WITH LARGE BROAD-WALL Configuration of the transition is shown in Fig. 1 and 2. A microstrip line, a probe and a waveguide short are located on the upper plane of the dielectric substrate. A rectangular patch element and a surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes surround the aperture of the waveguide on the lower plane of the dielectric substrate to connect the surrounding ground and the waveguide short electrically. For 79GHz UWB automotive radar applications, the required operation bandwidth is from

A

Waveguide short

φ

S

Via holes

Description

Probe

Width of patch element Length of patch element Broad wall length of waveguide

Wp

εr

Wl Port #2 A’ (a) Upper pattern

x

z

B

Vx

Waveguide with large broad-wall (lower pattern)

Rectangular patch element

W Vy b

L

a Surrounding ground

B’ (b) Lower pattern

Waveguide with large broad-wall

ρ A’

A

B’

B T z

Narrow wall length of waveguide

b

1.27 mm

Width of microstrip line

Wl

0.3 mm

Width of probe Overlap length of probe with rectangular patch element Width of gap

Wp

ρ

0.35mm 0.32 mm

G

0.1 mm

Thickness of substrate

T

0.127 mm

Relative permittivity

εr φ

2.2

Space between via holes

S

0.4 mm

Distance from broad wall to via hole

Vy

0.46 mm

Distance from narrow wall to via hole

Vx

0.4 mm

Diameter of via hole

0.2 mm

with the width Wp = 0.35 mm is inserted into the waveguide and overlaps on the rectangular patch element by the length ρ = 0.32 mm of the overlap length of the probe with the rectangular patch element. The distance Vy from the edge of the broad-wall of the waveguide to via holes on the broad-wall side is 0.46 mm. The distance Vx from the edge of the narrowwall of the waveguide to via holes on the narrow-wall side is 0.4 mm. The thickness of the dielectric substrate T is 0.127 mm with relative permittivity εｒ is 2.2. The parameters of the transition are presented in TABLE I. Mode conversion from the waveguide to the microstrip line is achieved by resonance of the rectangular patch element. The dominant TE10 mode of the waveguide is converted to the quasi-TEM mode of the microstrip line. III. NUMERICAL INVESTIGATION

b x

Value 2.26 mm 0.98 mm 3.1 mm

y

Port #1

Waveguide with large broad-wall

(c) Sectional view in yz-plane Fig. 2. Detailed configuration of the transition.

77 GHz to 81 GHz. In terms of the bandwidth, it becomes wider as broad-wall length a of the waveguide increases, and narrow-wall length b of the waveguide decreases [11]. The broad-wall length a of the waveguide is determined to be 3.1 mm which is the maximum width where higher order mode in the waveguide dose not propagate. The narrow-wall length b is determined to be 1.27 mm which is the same as the narrow-wall length of WR-10. A rectangular patch element with the width W = 2.26 mm and the length L = 0.98 mm is located on the lower plane of the dielectric substrate at the center of the waveguide. The width Wl of the microstrip line is 0.3 mm corresponding to approximately 56 ohm of characteristic impedance. The probe

The proposed transition is investigated numerically by using electromagnetic simulator based on the finite-element method. In this calculation, loss tangent tanδ = 0.001 and conductivity σ = 5.8 × 107 S/m are used as loss factors. The required operation bandwidth is from 77 GHz to 81 GHz. 0

0

-5

-1

-10

-2

-15

-3

-20

-4

-25

-5

-30

-6 60

65

70 80 85 75 Frequency [GHz] |S11 | [dB]

90

95

|S 21| [dB]

Fig. 3. Reflection characteristic |S11| and insertion loss |S21|.

|S21| [dB]

y

G

Name W L a

|S11 | [dB]

Microstrip line

TABLE I PARAMETERS OF TRANSITION

[V/m]

0

1.0e+5

-5

L = 0.98 mm

|S11| [dB]

1.0e+4

Intensity

-10

L = 0.99 mm -20 L = 0.97 mm

-25

y

z

-15

x

-30

1.5e-2

(a) Electric field intensity at 76.4 GHz

64

69

74

1.0e+5

84

89

L = 0.98 mm

L = 0.97 mm

94

L = 0.99 mm

Fig. 5. |S11| vs length L of the patch element. (Lower resonant frequency control.)

Intensity

1.0e+4

79 Frequency [GHz]

[V/m]

0 Vy = 0.48 mm -5 Vy = 0.47 mm -10

|S11| [dB]

1.5e-2

(b) Electric field intensity at 79 GHz [V/m] 1.0e+5

Vy = 0.46 mm

-15 -20 Vy = 0.45 mm

1.0e+4

Intensity

-25 -30 64

69

74

79

84

89

94

Frequency [GHz] Vy = 0.45 mm

(c) Electric field intensity at 85.5 GHz

1.5e-2

Vy = 0.46 mm

Vy = 0.47 mm

Vy = 0.48 mm

Fig. 6. |S11| vs distance Vy from the edge of broad-wall of the waveguide to via holes. (Higher resonant frequency control.)

Fig. 4. Electric field intensity distribution in xy-plane. 0 -5 ρ = 0.28 mm

ρ = 0.36 mm

-10 ρ = 0.32 mm

|S11| [dB]

Reflection characteristic |S11| and insertion loss |S21| of the transition with parameters in TABLE I are presented in Fig. 3. It can be seen from the simulation results that the bandwidth for the reflection coefficient |S11| below −15 dB is 14.4 GHz, and insertion loss |S21| is less than 0.28 dB from 77 GHz to 81 GHz. In this case, two different resonances are observed. Lower resonant frequency is 76.4 GHz and higher resonant frequency is 85.5 GHz. Figure 4 shows calculated electric field distributions in the xy-plane including BB'-line at 76.4, 79 and 85.5 GHz. It is observed that fundamental TM01 mode is excited at 76.4 and 79 GHz in Fig. 4(a) and (b). On the other hand, a higher order mode is observed at 85.5 GHz as shown in Fig. 4(c). Length L of the rectangular patch element affects lower resonant frequency as shown in Fig. 5. Lower resonant frequency can be controlled by the length L of the rectangular patch element. Distance Vy from the edge of broad-wall of the waveguide to via holes affects higher resonant frequency as shown in Fig. 6. Higher resonant frequency can be controlled by the distance Vy from the edge of broad-wall of the waveguide to via holes.

-15 -20 -25 -30 64

69 ρ = 0.28 mm

74

79 Frequency [GHz]

84

ρ = 0.32 mm

89

94

ρ = 0.36 mm

Fig. 7. |S11| vs length ρ of inserted probe .

Overlap length ρ of a probe with a rectangular patch element is most effective for impedance matching to the waveguide as shown in Fig. 7. At the same time, the width Wp of probe affects impedance matching as shown in Fig. 8.

In the next step, the fabrication and measurement will be done to confirm the performance of proposed transition and to compare simulation results with measured results.

0 -5

Wp = 0.40mm

|S11| [dB]

-10

REFERENCES

Wp = 0.30 mm -15

[1]

-20 -25

Wp = 0.35 mm

-30 64

69

74

79 Frequency [GHz]

Wp = 0.30 mm

84

89

Wp = 0.35 mm

94 Wp = 0.40mm

Fig. 8. |S11| vs width Wp of probe . 0

Bandwidth = 14.4 GHz

-5

|S11 | [dB]

-10

Wp = 0.3 mm ρ = 0.32 mm

-15 -20 Wp = 0.35 mm ρ = 0.32 mm

Wp = 0.4 mm ρ = 0.3 mm

-25 -30 64

69

74 79 Frequency [GHz]

Wp = 0.3 mm ρ = 0.32 mm Wp = 0.4 mm ρ = 0.3 mm

84

89

94

Wp = 0.35 mm ρ = 0.33 mm

Fig. 9. Wideband impedance matching . (Simultaneous optimization of ρ and Wp)

For wideband impedance matching, both of the length ρ of inserted probe and the width Wp of probe are optimized simultaneously as shown in Fig. 9. It can be seen from the simulation results that both of ρ and Wp affect wideband impedance matching. Least insertion loss and wideband characteristic is obtained at ρ =0.32 mm and Wp = 0.35 mm shown in Fig. 3. IV. CONCLUSION A transition to connect a microstrip line to a waveguide is examined in the millimeter-wave band. A waveguide with large broad-walls is effective for wideband characteristics. Consequently, wideband characteristic for 79 GHz UWB systems is obtained by controlling the distance Vy from the edge of broad-wall of the waveguide to via holes on broadwall side. A numerical investigation of a transition shows appropriate parameters. Proposed transition has insertion loss of 0.28 dB from 77 GHz to 81 GHz, and a bandwidth of 18.2 % (14.4GHz) for the reflection coefficient below −15 dB.

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