MILLIMETER-WAVE FIBER-OPTIC COMMUNICATION SYSTEM ...

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Millimeter-wave fiber-optic communication system using InP HEMTs Author(s): Orzati, Andrea Publication Date: 2003 Permanent Link: https://doi.org/10.3929/ethz-a-004619835

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MILLIMETER-WAVE FIBER-OPTIC COMMUNICATION SYSTEM USING INP HEMTS

Andrea Orzati DISS. ETH No. 15102

DISS. ETH No. 15102

MILLIMETER-WAVE FIBER-OPTIC COMMUNICATION SYSTEM USING INP HEMTS

A dissertation submitted to the SWISS FEDERAL INSTITUTE OF TECHNOLOGY ZURICH for the degree of Doctor of Technical Sciences presented by ANDREA ORZATI Laurea in Ing. Elettronica, Universit`a degli Studi di Cagliari Born July 11, 1973 in Cagliari (Sardinia, Italy) accepted on the recommendation of Prof. Dr. W. B¨achtold, examiner Prof. Dr. H. J¨ackel, Prof. M. Vanzi, coexaminers 2003

You see, wire telegraph is a kind of a very, very long cat. You pull his tail in New York and his head is meowing in Los Angeles. Do you understand this? And radio operates exactly the same way: you send signals here, they receive them there. The only difference is that there is no cat. A. Einstein

Acknowledgments I would like to acknowledge all the people who, in different ways, supported me during my dissertation. In the first place, I would like to thank my “Doktorvater” Prof. Werner B¨achtold for giving me the opportunity to work in such a valid research group. I would also like to express my gratitude to Prof. Heinz J¨ackel and Prof. Massimo Vanzi who accepted to be my coexaminers. I would like to acknowledge Otte J. Homan for his (not only) scientific support, Hanspeter Meier for his patient and indispensable work in circuit fabrication, and Hansruedi Benedickter for revealing me the fine art of high-frequency characterization. My gratitude goes also to Bruno Graf from Avalon Photonics for helping us in the dielectric layer deposition, to Iwan Schnyder, for bearing with me the “slings and arrows” of the project, and to Dominique Schreurs for sharing her great experience in device modeling. A special thank goes to all my friends and colleagues at the institute, for the cheerful working atmosphere they managed to create. Thanks to Franck Robin for convincing me to wake up early to go skiing, climbing, or hiking, to Thomas Brauner for his patience with my poor sailing skills, to Esteban “Abuelo” Moreno for being always willing to try my latest brownies recipe, and to Federico Beffa for showing me a delicious alternative to the ETH Mensa. Un ringraziamento speciale va anche ai miei genitori, per il loro inesauribile sostegno, alla mia psicologa personale e a tutte le persone che mi sono state vicine in questi anni passati a “dissertare”.

Contents Acknowledgments

v

Abstract

xi

Riassunto

xiii

1 Introduction 1.1 Motivations 1.2 Project description 1.3 Outline of the work

1 1 4 4

2 The MFOCS System 2.1 System Design 2.1.1 Important Terminology 2.1.2 Guidelines for the Design of mm-wave fiber-optic communication system 2.1.3 Different System Concepts 2.1.4 Proposed Approach 2.2 Hub Design 2.2.1 Analysis of the Fiber-Optic Data Link 17 2.2.2 Analysis of the Radio Link 2.2.3 Hub Architecture 2.3 Conclusions

7 7 7

3 Process Technology 3.1 The High Electron Mobility Transistor 3.1.1 From the two-dimensional electron gas to InP HEMTs

8 11 15 17 19 23 25 27 27 27

viii

Contents

3.1.2 Physical principles of heterostructures 28 3.1.3 HEMT Material Systems 3.1.4 Device structure 3.2 Process Description 3.3 Components 3.3.1 InP HEMTs 3.3.2 Transmission Lines 3.3.3 Line Discontinuities 3.3.4 Thin Film Resistors 3.3.5 MIM Capacitors 3.3.6 Spiral Inductors

29 31 32 35 35 37 40 44 44 44

4 InP HEMT Large Signal Model 4.1 Large-Signal Modeling of High-Frequency FETs 4.2 Small-Signal and Large-Signal Equivalent Circuit 4.3 Model Extraction 4.4 Model Validation 4.4.1 Small-signal verification 4.4.2 Large-signal verification 4.4.3 Large-Signal Validation through MMIC Design 4.5 Limitations of Look-up Table Based LargeSignal Models 4.6 Conclusions

47

5 MFOCS Circuits 5.1 0-20 GHz Traveling Wave Amplifier 5.1.1 Introduction 5.1.2 Circuit Design 5.1.3 Measurement Results 5.2 16 GHz Up-converting Image-Rejection Resistive Mixer 5.2.1 Introduction

71 71 71 72 75

48 50 55 58 58 61 64 68 69

77 77

Contents

ix

5.2.2 Circuit Design 5.2.3 Measurement Results 5.3 16-48 GHz Active Frequency Triplers 5.3.1 Introduction 5.3.2 Theoretical Analysis 5.3.3 Circuit Design 5.3.4 Measurement Results 5.4 V-band Up-Converting Active Mixer 5.4.1 Introduction 5.4.2 Circuit Design 5.4.3 Measurement Results

77 79 83 83 83 89 92 97 97 97 99

6 Flip-Chip Bonding 6.1 General Aspects 6.2 Why Flip-Chip 6.3 Process description 6.4 Optimization for V-band applications 6.5 Application to InP MMICs 6.6 Encountered Problems 6.7 Conclusions

103 104 104 106 107 113 116 116

7 Summary, Conclusions, and Outlook 7.1 System Design 7.2 InP HEMT Large-Signal Modeling 7.3 Circuit Design and Fabrication 7.4 Flip-Chip optimization for V-band Applications 7.5 Conclusions and Future Work

119 119 120 120

Bibliography

125

121 122

Abstract Goal of the work presented in this dissertation is to contribute to the technology, the design, and the characterization of a 60 GHz transceiver for a fiber-optic millimeter-wave wireless LAN. This was done in the frame of the MFOCS (Millimeter-wave Fiber-Optic Communication System) project, funded by the the Swiss Federal Institute of Technology. Through an analysis of the characteristics of both the optical link and the radio link, a feasible system architecture and realistic hub specifications can be derived. Remote generation of the mm-wave local oscillator through frequency multiplication in the hub is proposed as the best way to generate a stable high-frequency LO signal and contemporaneously overcome the problems due to chromatic dispersion in the fiber. A two-step signal up-conversion in the hub is suggested instead of a direct up-conversion in order to relax the mixer specifications in terms of LO- and image-rejection and allow the frequency conversion circuits to be fabricated on the InP HEMT process of the Microwave Electronics Laboratory. In order to design and fabricate integrated circuits that meet the required specifications, precise models for both passive and active components are indispensable. Transmission line elements were modeled by means of an electromagnetic simulator. For InP HEMTs, a look-up table-based large-signal model was developed which takes impact ionization into account. S-parameter measurements demonstrate that the linear performance of the model is excellent up to 110 GHz. The non-linear performance is validated in the impact ionization region as well as up to 64 GHz by non-linear measurements at different bias points. Using the developed model, different circuits were designed and

xii

Abstract

fabricated. They include a 0 − 20 GHz optical receiver amplifier, a K-band up-converting image-rejection resistive mixer, a V-band up-converting active mixer, and a 16 − 48 GHz frequency tripler. The measured performances of the fabricated circuits are in line with the required system specifications. In order to assemble the system, a simple low-cost gold-ball flip-chip bonding process was optimized for V-band applications. For the optimized CPW-CPW flip-chip transition, return losses lower than −20 dB up to 80 GHz were measured. The application of the optimized bonding technique to microwave amplifiers was also demonstrated.

Riassunto Il lavoro di ricerca confluito nella presente dissertazione vuole essere un contributo allo sviluppo della tecnologia, delle tecniche di progettazione e delle tecniche di caratterizzazione di un ricetrasmettitore per reti locali che utilizzi sia fibre ottiche che segnali radio ad onde millimetriche. Per la realizzazione del sistema si `e proposto l’uso di circuiti integrati in fosfuro d’indio basati su transistor ad alta mobilit`a elettronica (HEMT), realizzati usando il processo sviluppato dal Laboratorio di Alta Frequenza (IFH). La ricerca `e stata finanziata dal Politecnico Federale di Zurigo, nell’ambito del progetto MFOCS (Millimeterwave Fiber-Optic Communication System). L’architettura e le specifiche del sistema sono state derivate da attente analisi del collegamento su fibra ottica e del canale radio. Nel ricetrasmettitore, la generazione del segnale dell’oscillatore locale a onde millimetriche avviene tramite la moltiplicazione di un segnale di riferimento a bassa frequenza ricevuto attraverso la fibra ottica. In questo modo vengono contemporaneamente risolti i problemi della dispersione cromatica nella fibra ottica e della generazione di un segnale di riferimento ad onde millimetriche con i giusti requisiti di stabilit`a. Nel ricetrasmettitore, la conversione di frequenza viene effettuata in due stadi; questa soluzione ha l’effetto di rilassare le specifiche dei circuiti che operano la miscelazione del segnale, sia dal punto di vista della reiezione delle componenti spurie dovute sia all’oscillatore locale che alla frequenza immagine. Per poter progettare e, quindi, realizzare circuiti integrati che rientrino nelle specifiche richieste dal sistema, `e indispensabile disporre di modelli circuitali precisi, sia per i componenti passivi che per gli HEMT. Le discontinuit`a delle linee di trasmis-

xiv

Riassunto

sione sono state modellate con successo usando un simulatore elettromagnetico. Per gli HEMT, `e stato realizzato un modello a largo segnale basato su look-up table che tiene conto degli effetti della ionizzazione da impatto. Misure dei parametri di dispersione hanno dimostrato come la capacit`a di predizione del comportamento lineare del transistor sia ottima fino a 110 GHz. Il comportamento a largo segnale del modello `e stato verificato, per diversi punti di lavoro, sia a basse frequenze, dove `e osservabile l’effetto della ionizzazione da impatto, sia a frequenze delle onde millimetriche (64 GHz). Con l’ausilio dei modelli sviluppati, i diversi componenti del sistema sono stati progettati e realizzati nella camera bianca dell’istituto. Questi circuiti sono: un amplificatore distribuito per il foto ricevitore con una banda passate da 200 MHz a 20 GHz, un up-converting mixer resistivo in banda K a reiezione della frequenza immagine, un up-converting mixer attivo in banda V ed un moltiplicatore di frequenza da 16 a 48 GHz. Le caratteristiche di tutti i circuiti misurati rientrano nelle specifiche dettate dal sistema. Allo scopo di facilitare l’assemblaggio del sistema, `e stato sviluppato e ottimizzato per applicazioni in banda V un processo di montaggio a chip rovesciato (flip-chip). In questo modo `e stato possibile ottenere transizioni tra il chip e la scheda di montaggio con coefficienti di riflessione misurati inferiori ai −20 dB per frequenze fino ad 80 GHz. Questa tecnica `e stata usata con successo per il montaggio su scheda di amplificatori integrati a microonde.

1 Introduction 1.1

Motivations

Wireless systems represent a significant part of today telecommunication business. This is mainly due to the extraordinary technical and commercial success experienced by mobile phones in the last ten years. These days, thanks to the increasing popularity of portable personal computers and data entry terminals, academic and corporate interest has gradually shifted to wireless data systems and wireless local area networks (WLANs). Second generation WLANs offering data rates between 10 and 25 Mb/s are in a phase of advanced development (wireless ATM), if not already on the market (wireless Ethernet). Nevertheless, there is still a multitude of multimedia applications requiring wireless transmission over a short range which can not be satisfied by existing communication systems [1, 2]. Some of the most significant short-range wireless applications are summarized in Table 1.1. The frequency band around 60 GHz has been indicated as the most convenient for short-distance wireless communications [1, 3]. As shown in Fig. 1.1, in a 8 GHz band centered around 60 GHz an attenuation peak (10 − 15 dB/km) due to atmospheric oxygen limits communication distance to less than one kilometer. As a result, this portion of the spectrum allows heavy frequency reuse, particularly in an indoor environment, where the signal propagation is de facto heavily conditioned by the geometry of the building [4]. These are all reasons why in Europe, in the United States, as well as in Japan, huge band-

2

Introduction Table 1.1 Some of the most significant short-range wireless applications.

Application

Capacity per user (Mb/s)

Wireless LAN bridge Virtual reality allowing free body movements Wireless IEEE 1394 Wireless High-resolution recording camera Trading terminal having multiple video channels that can be viewed simultaneously Wireless News tablet Internet download of lengthy files High-quality video-conference Ad hoc communication, i.e., direct communication between notebooks, between notebook and nearby printer, etc.

100 − 1000 450 100, 200, 300 150 − 270 50 − 100 50 − 100 10 − 100 10 − 100 0.1 − 100

widths have been allocated around the 60 GHz band, which can be therefore proficiently exploited for very high data rates transmissions [5, 6]. Next generation WLANs will be found in offices as well as at home and will consist of various pico-cells, each one covering a single room. In a typical installation of a future WLAN, there will be a large number of fixed access points connected to some base stations through a fixed cable infrastructure. Optical fibers are the natural candidate to link access points with base stations because of their excellent characteristics in terms of bandwidth and propagation losses [7]. It is, therefore, not controversial to expect that fiber-optic-fed millimeter-wave wireless systems will be a central element of future networking infrastructures. The data will be distributed through a building by means of optical fibers and received by hubs responsible for the optical-electrical conversion and for the signal transmission, at millimeter-wave frequencies, to the final users. An example of

Motivations

3

3

Atmospheric Attenuation (dB/km)

10

2

10

O

2

1

10

0

10

−1

10

60 GHz −2

10

1

10

2

10

3

10

Frequency (GHz)

Fig. 1.1 Atmospheric attenuation of microwaves as a function of frequency.

a possible future office environment is illustrated in Fig. 1.2. High electron mobility transistors (HEMT) based on the InP material system have have proved to be of high interest for millimeter-wave applications. In the V-band, compared to GaAs pseudomorphic HEMTs (PHEMTs), they have shown higher gain, lower noise figure, and, in spite of their lower breakdown voltage, better power added efficiency [8–10]. For these rea-

Fig. 1.2 A 60 GHz wireless LAN system in an office environment.

4

Introduction

sons, in the last fifteen years this field of research has been the object of constant attention at the Swiss Federal Institute of Technology in Zurich (ETHZ). Significant advantages are expected from employing InP HEMTs technology in millimeterwave wireless communication systems. Goal of the project described in this work was to contribute to the technology, the design, and the characterization of a 60 GHz transceiver for a fiber-optic millimeter-wave wireless LAN. This was done without the pretense of delivering a commercially attractive product, but with the intention of contributing, by showing its advantages, to the establishment of a mature InP HEMT circuit technology at ETHZ. The project has been carried out using the in-house InP HEMT process of the Laboratory for Electromagnetic Fields and Microwave Electronics (IFH). The following chapters will present the author’s efforts and achievements during the last four years. 1.2 Project description The research work upon which this dissertation is based was carried out within the MFOCS (Millimeter-wave Fiber-Optic Communication System) project, directly funded by the the Swiss Federal Institute of Technology. Goal of the project was to contribute in the technology, design, and characterization of an integrated transceiver for a fiber-optic millimeter-wave wireless LAN. The project was undertaken in collaboration with the Electronics Laboratory (IfE) of the Swiss Federal Institute of Technology in Zurich, which runs an InP HBT process, and was aimed to support InP technology at ETHZ. IFH was responsible for the transceiver design which used InP HEMTs, while IfE attended to the generation and transmission of the optical signal through the fiber, which was implemented on an InP HBT process. 1.3 Outline of the work The present dissertation is structured as follows. In Chapter 2 the system design constrains are described and the chosen

Outline of the work

5

system architecture is presented and motivated. Chapter 3 describes the in-house InP HEMT process employed for the MMICs’ fabrication, its features, and its limitations. Chapter 4 presents the InP HEMT large-signal model developed during the project and used for circuit design; the model takes impact ionization into account and was validated up to the Vband by linear and non-linear measurements. All the MMICs that have been designed, fabricated, and characterized are presented in Chapter 5; their design issues are discussed together with their measured performance. Chapter 6, finally, presents a flip-chip mounting technique optimized for V-band applications and its application to millimeter-wave coplanar circuits.

2 The MFOCS System The present chapter describes the design flow that, starting from unrefined considerations, led to the development of the system architecture and, from there, to concrete circuit specifications. This chapter is structured as follows. In the first section, the architecture of the complete system is presented. After defining some key terms often used in the rest of the chapter, the practical aspects limiting the designer freedom in the choice of the system architecture are discussed. Finally, after a review of the most significant approaches found in the literature, the proposed system is presented. In the second part of the chapter the attention is focused on the transceiver, whose development was the actual goal of the work. In the description of how the transceiver specifications are derived, also typical engineering issues are discussed: the analysis of the fiber-optic data link and the radiation constraint given by the current legislation. At the end of the chapter, the transceiver architecture is presented and a description of the needed circuits is given. 2.1

System Design

2.1.1 Important Terminology At the beginning of this section it is needful to univocally define some terms that will be often used later in the chapter. The section, which has not the pretension to be systematic or exhaustive, contains a list of important terms, associated with

8

The MFOCS System

their meaning in the context of this work. This is necessary because of the lack of harmonization in the terminology found in the current literature and will provide an easier understanding of the following sections. • Base station. It is the part of the system responsible for distributing the information coming from a data source (e.g. a server) to a number of remote hubs and vice versa. The base station is wired to the different hubs through , e.g., optical fibers (Fig. 2.1). • Hub. It is the interface between the final user (e.g. a mobile computer) and the communication system (Fig. 2.1). It provides a bidirectional link with both the base station and the final users. In a communication system there are several hubs connected with the same base station. As sometimes found in literature, this part of the system can also be called transceiver or transponder or access point. • Mobile receiver. It is the final user of the WLAN (Fig. 2.1). • Picocell. It is the portion of space served by a single hub (Fig. 2.1). In a 60 GHz WLAN the picocell usually coincides with a single room, due to the propagation characteristics of millimeter-waves at these frequencies. • Down-link. It is the unidirectional radio link (in case of a WLAN) that connects the hub with the final user (Fig. 2.1). • Up-link. Analogous to the down-link, but directed from the final user to the hub (Fig. 2.1). 2.1.2

Guidelines for the Design of mm-wave fiberoptic communication system

In the design of a millimeter-wave fiber-optic communication system there are various practical issues that have to be taken into account and that constrain the choice of the system designer. These factors are of disparate nature and will be illustrated in the following sections.

System Design

9

Fig. 2.1 Schematic representation of the different components of a WLAN.

Current Regulation on Frequency Allocation Frequency allocation in the mm-wave region changes depending on the country. While in Japan the Multimedia Mobile Access Communication (MMAC) committee is still deciding how to exploit the advantages of the 60 GHz band, in Europe and in the United States the European Commission for Posts and Telecommunications (CEPT) and the Federal Communications Commission (FCC), respectively, already proposed an ad hoc regulation on this matter [5,6]. In Europe, the 62−63 GHz and the 65 − 66 GHz bands are dedicated to WLAN down-link and up-link, respectively. In the United States, the 59 − 64 GHz frequency band was set aside for general unlicensed applications. This is the largest contiguous block of radio spectrum ever allocated. In this work, the recommendations of the European Commission for Posts and Telecommunications are taken as reference. Limitations due to Chromatic Dispersion The term chromatic dispersion indicates the wavelength dependence of the optical-signal propagation velocity on the fiber material. Since every optical signal has a finite spectral width,

10

The MFOCS System

this dispersion results in a spreading of the signal spectrum. In intensity-modulated direct-conversion links operating above 20 GHz, chromatic dispersion significantly limits the transmission distance by introducing a carrier to noise penalty in the transmitted signal [11]. Due to chromatic dispersion, an intensitymodulated 60 GHz signal would be completely extinct after one kilometer. In [12] the transmission penalty due to chromatic dispersion in a single-mode fiber (SMF) is calculated as a function of the signal frequency and of the fiber length. It is shown that for a low-penalty transmission over 10 km SMF the modulation frequency can not be higher then 20 GHz. Numerous methods have been studied to overcome the effects of chromatic dispersion. Smith and Novak, using an approach based on optical single-sideband modulation techniques [13, 14] demonstrated a 51.8 Mb/s digital transmission at 12 GHz over 80 km of single-mode fiber. Other methods based on optical heterodyning [11, 15] were also successful. Generation of a Stable High-Frequency Local Oscillator Signal A crucial point in the design and implementation of mm-wave fiber-optic communication systems is the generation of millimeterwaves in the hub, since their direct transmission from the base station over the fiber encounters chromatic dispersion. A possible way to skirt the issue would be to have a high-frequency local-oscillator (LO) in the hub and use it to mix the data signal up to 60 GHz. This solution is, for technological reasons, not as practicable as it seems in a first moment. In fact, millimeter-wave oscillators with the reproducibility and stability characteristics required by the application are not feasible. Possible alternatives are the use of frequency multiplication chain together with a stable low-frequency LO source or of voltage controlled oscillators with a control loop. Both solutions drastically increase the hub complexity and, therefore, its cost and can not considered valid. It is clear that the generation of a stable high-frequency local oscillator can not be performed in the hub but other solutions have to be found for it.

System Design

11

2.1.3 Different System Concepts In this section, an overview of different approaches to the design of millimeter-wave fiber-optic system is presented. The attention is focused on the down-link implementation. For almost all the described systems, in fact, the up-link and the down-link have analogous implementations, and a description of both of them would just cause redundancy. In case up- and down-link show significant differences, it will be brought to the readers attention. Millimeter-waves over the fiber This approach was proposed by Imai and Kawamura in 1995 [16, 17]. The idea is to modulate the data signal directly in the base station using a 43.75 GHz LO and to send the modulated signal over the fiber. In the hub, the optical signal is received, amplified, and transmitted via antenna to the mobile receiver without the need for any frequency conversion. This method has the advantage of keeping the hub architecture at minimum complexity, but has also some significant disadvantages. In the first place, the fiber length, i.e. the distance between the base station and the hub is strongly limited by chromatic dispersion, which is already very strong at 43.75 GHz. Additionaly, very high frequency photodetectors, lasers, and laser-modulators are needed, not to mention the fact that it would be very difficult to extend this approach to the 60 GHz band. Nevertheless, the mentioned authors were able to successfully demonstrate a 118 Mb/s digital transmission over 50 m optical fiber and 4 m wireless link. A schematic representation of the proposed system is depicted in Fig. 2.2(a). Wavelength Division This method was proposed in 1997 by No¨el and is described in [18]. A broad-band (120 Mbyte/s QPSK data and 20 digital TV channels) 60 GHz transmission over a 13 Km optical fiber and a 5 m radio link was demonstrated. The system works as follows: a 59 GHz LO is generated in the base station using a

12

The MFOCS System

master/slave distributed-feedback laser configuration and sent over the fiber. This signal has a very high spectral purity which makes it almost insensitive to chromatic dispersion. The data signal is in the 0.8 − 2 GHz band and is transmitted using a distributed-feedback laser, separated from the LO in the wavelength domain. An electroabsorption modulator (EAM) is used in the hub to receive (and transmit) the data signal over the fiber, while the carrier is detected in the hub by means of a photo diode. The LO signal is used to up-convert the data signal to the 60 GHz band. The generated millimeter-wave signal is then sent to a high-gain antenna. This approach overcomes the effects of chromatic dispersion in the fiber by using a high-purity LO source and by separating data signal and carrier in the wavelength domain. On the other hand, high-cost components such as an electroabsorption modulator, a wavelength division multiplexer, and a 60 GHz photoreceiver have to be used in the hub in order to obtain these, indeed, excellent results. This approach is schematically represented in Fig. 2.2(b). Optical Heterodyning A demonstration of a system based on the optical heterodyning principle was given by Braun in 1998 and is presented in [19,20]. The base station generates two optical signals with a frequency spacing equal to the frequency of desired millimeterwave signal. If one of these laser is amplitude modulated with the data signal and if coherent detection in the hub is guaranteed, then the beating of the two optical signal generates a millimeter-wave carrier amplitude-modulated by the data signal. There are different ways of generating the two beating optical signals, both with single optical sources (modulation sideband techniques, mode locked laser, dual-mode laser, FMmodulated laser in conjuntion with fiber dispersion) and with multiple optical sources. This method guarantees a simple and effective down-link architecture and is very well suited for applications where only unidirectional transmission is required, but

System Design

Fig. 2.2 Schematic representation of different system concepts.

13

14

The MFOCS System

still presents open issues when it comes to bidirectional transmission. It is possible using optical sideband injection locking to generate a high-frequency LO signal in the hub, but it still remains complicated to make an effective use of it. A simplified version of the system is given in Fig. 2.2(c). Subharmonic Up-conversion This approach was implemented by Smith and Novak in 1998 and its detailed description can be found in [21]. They demonstrated a full-duplex transmission over over 40 km of standard single mode fiber and 5 m radio link. A schematic representation of the system can be found in Fig 2.2(d). In the base station, a data signal in the 2 − 3 GHz band is combined with a 17 GHz LO and the resulting signal is then transmitted over the fiber. In the hub the signal is detected and the LO is separated from the data by means of a diplexer. Then, the LO signal is used in a sub-harmonically pumped image-rejection mixer to convert the data up to the 40 GHz band. This approach solves the problem of chromatic dispersion by limiting the frequency of the signal sent over the fiber to 17 GHz and has the advantage of presenting a simple hub architecture. However, this method can arduously be extended to the 60 GHz band just by using sub-harmonically pumped mixers. Frequency Multiplication in the Hub In 1998 a novel system design and transmission experiments were presented by Kojucharow [22,23], where the digital transmission of 50 Mb/s over a 4.5 m radio link was successfully demonstrated. A schematic representation of the proposed approach is depicted in Fig. 2.2(d). The base station design is analogous to the one proposed by Smith in [8] and described in the previous section. A data signal modulated around 2.223 GHz is combined with a 4.8 GHz local oscillator by means of an EAM. The resulting signal is then sent over the fiber and detected by a photo diode at the hub. After reception the LO component of the signal is separated from the data using a

System Design

15

microstrip diplexer. The LO is them multiplied in frequency by a factor of twelve and used to up-convert the data signal to the V-band. This approach has several advantages; effects of chromatic dispersion are minimized by limiting to 4.8 GHz the highest frequency transmitted over the optical fiber, the hub architecture is kept quite simple, and no high-performance optical component is required. 2.1.4 Proposed Approach A schematic representation of the proposed system is show in Fig. 2.3. This design is an extension of the approach suggested by Kojucharow and described in [22, 23]. Some modifications were introduced in order to take the current European regulations on frequency allocation into account and to further simplify the hub architecture. In the base station, a 155 Mb/s NRZ data signal modulates the phase of a 1.5 GHz carrier. The modulated signal is then combined with a 16 GHz local oscillator, and the resulting signal is given as input to a laser driver and sent through the optical fiber. Since the maximum modulation frequency of the optical signal is 16 GHz, no significant penalty due to chromatic dispersion is expected, if the fiber length is kept below 10 km. Using 16 GHz as reference frequency in the base station instead of 4.8 GHz, facilitates the generation of a high-frequency local oscillator in the hub. Since no frequency multiplication chain is needed, the result is a simpler architecture. Furthermore, a 10 km transmission distance over the fiber is more than what can be expected for this kind of application. In the hub, the optical signal is detected, amplified, and separated by a diplexer into its data and LO components. A power splitter divides the 16 GHz LO signal in two; a part is used to mix the data up to 14.5 GHz, the other is converted by a frequency tripler into a 48 GHz reference signal, which is then used to up-convert the data signal up to 62.5 GHz, i.e. in the middle of the frequency band allocated for the down-link. In contrast with the approaches presented in the previous sections, the up-

16

The MFOCS System

Fig. 2.3 Schematic representation the proposed system.

conversion of the data signal is performed in two successive steps. The reason of this choice lies in the extreme difficulty of designing a monolithically integrated V-band mixer able to directly up-convert the 1.5 GHz data signal to 62.5 GHz and ensure at the same time an effective suppression of the 64 GHz LO together with a good image rejection (65.5 GHz). The quality factors required by the filtering networks, unfortunately, do no belong to this world. Additionally, the two-step up-conversion scheme only requires an additional mixer if compared to direct up-conversion. The generated 16 GHz and 48 GHz carriers can be also employed to down-convert the up-link signal.

Hub Design

2.2

17

Hub Design

In the previous sections, an approach to the development of a millimeter-wave fiber-optic communication system was proposed. In this second part of the chapter, the attention is focused on the hub, which was the main topic of the work leading to this dissertation. The hub architecture was already introduced in the previous sections. Nevertheless, to derive correct specifications for the transceiver, a deeper insight of how this part is interfaced with the rest of the system is necessary. The hub is connected to the base station by an optical-fiber link and communicates with the mobile receiver through a radio channel. To lay down a proper set of specifications for the hub, an analysis of these two links is, therefore, essential. 2.2.1

Analysis of the Fiber-Optic Data Link

In this section, a simple analysis of the optical data link is presented, which can be used to derive coarse specifications for the optical power transmitted from the base station to the hub as a function of the receiver noise figure (NF). This is done under the assumption that the data signal is phase modulated using a antipodal phase shift keying (2-PSK) and has a bandwidth B of 1 GHz. The transmission occurs over a 10 km single-mode fiber with an attenuation of 0.25 db/km, for the transmitting laser a relative intensity noise (RIN)of −137 dB/Hz was assumed, and the responsivity ηP D of the photo diode was set to 0.4 A/W. These values are extracted from the data sheets of commercially available components. To simplify the calculations, the optical modulation index is considered to be one and the laser non-linearities, as well as the finite extinction ratio and the dark current are not considered. For an antipodal 2-PSK modulation the bit error probability (BEP) is given by equation 2.1 [24]: 1 BEP = erf c 2

r

 SN R . 2

(2.1)

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The MFOCS System

Fig. 2.4 Noise equivalent circuit of an optical receiver.

According to the International Telecommunication Union - Telecommunication Standardization Sector (ITU-T) recommendations ( [25]), a good transmission quality requires a BEP smaller than 10−10 , which, from equation 2.1, leads to a SNR greater than 13 dB. Figure 2.4 shows the noise equivalent circuit of a optical receiver [26]. The light power Popt coupled into the photo diode generates an electrical current ηP D Popt at the receiver input which leads to a quantum shot noise current: 2 hIsh i = 2qηP D Popt B,

(2.2)

where q is the elementary charge. Another noise source derives from the relative intensity noise of the transmission laser: 2 IRIN = RIN (ηP D Popt )2 B.

(2.3)

The noise contribution of the amplifier can be calculated from its noise figure (NF) and from its equivalent input resistance Ramlpi . The amplifier noise current can be expressed as:   4kT B 2 · NF − 1 , (2.4) In = Rampli where k is the Boltzmann constant (1.38 · 10−23 J/K) and T the temperature in Kelvin. The total noise current becomes: q 2 2 i + hIR2 i + IRIN + In2 , (2.5) In,tot = hIsh and the SNR can be expressed as:

Hub Design

19

50 45

Pt = 10 mW

SNR (dB)

40 Pt = 1 mW

35 30 25 20

Pt = 0.1 mW

15 13 10 0

2

4

6

Amplifier Noise Figure (dB)

8

10

Fig. 2.5 Noise equivalent circuit of an optical receiver.



 Popt ηP D √ SN R = 20log . In,tot B

(2.6)

Figure 2.5 shows the SNR as a function of the amplifier noise figure for a transmission power ranging from 0.1 to 10 mW. It can be deduced that the requirements on the amplifier noise figure are quite relaxed for this kind of application. 2.2.2 Analysis of the Radio Link In this section, the different aspects of the radio link between the hub and the mobile receiver are investigated. This is necessary not only to determine the minimum transmitted power which is required for a given bit-error-rate, but also to set its upper limit according to the existent regulations on electromagnetic emissions. Aspects Related to Current Regulations on Electromagnetic Emissions The Swiss Federal Council, in an ordinance relating to protection from non-ionising radiation [27], set 1 mW/cm2 as maximum allowed power density at 60 GHz. In Fig. 2.6 power den-

20

The MFOCS System 2

Power Density (mW/cm2)

1.8 1.6 25 dBi 1.4 1.2 20 dBi 1 0.8 15 dBi 0.6 0.4

10 dBi

0.2 0 0

5

10

15

20

25

30

Transmitted Power (dBm)

Fig. 2.6 Power Density at 0.5 m distance from the transmitter for four different antenna gains.

15

Power Density (µW/cm2)

25 dBi

20 dBi 10 9 15 dBi

10 dBi 5

0 0

5

10

15

20

25

30

Transmitted Power (dBm)

Fig. 2.7 Power Density at 3 m distance from the transmitter for four different antenna gains.

Hub Design

21

−90

−92

Pmin (dBm)

−94

−96

−98

−100

−102 0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Signal Bandwidth (GHz)

Fig. 2.8 Minimum received power as a function of the signal bandwidth for a 2-PSK modulation.

sity at 0.5 m distance from the transmitter is plotted as a function of transmitted power for four different antenna gains. In order to trespass the limits set in [27] more than 20 dBm transmitted power in combination with a 25 dBi antenna is needed. Rules set by the Federal Commission for Communication In the United States, the Federal Commission for Communication (FCC) allows 10 W of equivalent isotropic radiated power (EIRP) in the 60 GHz frequency band [6]. This corresponds to a maximum power density of 9 µW/cm2 at 3 m distance from the transmitter. Given that similar rules are going to be adopted in Europe as well, this sets another upper limit for the transmitted power. Figure 2.7 shows the power density at 3 m from the antenna for four different antenna gains. It can be noticed that these recommendations are more restrictive than the ones in [27].

22

The MFOCS System −45 −50 −55

PR (dBm)

−60 −65 −70 Pt=15 dBm

−75

P =10 dBm

−80

t

P =5 dBm

−85 −90 1

t

2

3

4

5

6

7

Transmission Distance (m)

8

9

10

Fig. 2.9 Received power as a function of transmission distance.

Link Budget In order to determine the minimum transmitted power which is required for a satisfactory link performance, a two-way link equation was adopted, which has some modifications in order to take additional losses at millimeter-wave frequencies into account [28, 29]. The minimum allowed received power can be expressed as: Pmin = 10log(kT B) + N Frec + SN Rmin ,

(2.7)

where k is the Boltzmann constant, T the room temperature in Kelvin, B the signal bandwidth, N Frec the noise figure of the mobile receiver, and SN Rmin is the minimum allowed SNR. In equation 2.7, 10log(kT B) represents the gaussian noise added by the radio channel. In Fig. 2.8 the minimum received power for a N Frec of 10 dB and a SN Rmin of 13 dBm (2-PSK with 10−10 BER) is shown as a function of the signal bandwidth. According to [28] the received power can be calculated as: Prec =

Pt Gt Gr λ2 Fm . 16π 2 Ra

(2.8)

Hub Design

23

In the 2.8, Pt is the transmitted power, Gt and Gr the antenna gains of the transmitter and of the receiver, respectively, λ is the operating wavelength (0.0045 m), Fm is a link margin introduced to compensate non-ideal effects, R is the transmission distance, and a is a loss exponent that can be experimentally determined. Assuming a total antenna gain (Gt Gr ) of 20 dBi, a link margin of −15 dB, and a loss exponent of 3, we calculated the curves of Fig. 2.9, where the received power is shown as a function of the transmission distance for three different values of the transmitted power. It can be observed that 5 dBm transmitted power are enough to ensure the transmission of a 2-PSK modulated signal with a bandwidth of 1 GHz. The hub transmission power was chosen to be 10 dBm, which lies on the middle between the minimum and the maximum permitted power and represents a feasible goal with the available technology. 2.2.3 Hub Architecture The link equations described in the previous sections were used to develop the fuzzy hub description given in Section 2.1.4 into a precise and feasible design. Figure 2.10 presents the final design of the hub. The marked circuits were developed during the project and will be presented in this dissertation. The circuits

Fig. 2.10 Schematic representation of the proposed hub architecture.

24

The MFOCS System

needed for the realization of the hub are: • 0-20 GHz Traveling Wave Amplifier (TWA). It amplifies the electrical signal generated by the photo diode. Its specifications are quite relaxed both in terms of noise figure and of gain. • Active Diplexer. This component takes the signal coming from the TWA and separates the frequency components in the 0 − 2 GHz band from the 16 GHz LO. It is called active because the LO signal is also amplified. • K-Band Resistive Mixer. It mixes the data signal from the 1 − 2 GHz band up to the 14 − 15 GHz band. It has to provide a sufficient image suppression. • Frequency Tripler. It multiplies the 16 GHz LO frequency by a factor of three up to 48 GHz, thus generating the LO signal needed by the V-band active mixer for the final upconversion. • V-Band Active Mixer. This circuit performs the final frequency conversion that finally brings the data signal in the 62 − 63 GHz transmission band using the 48 GHz LO. Its conversion loss has to be as low as possible. • 16 and 48 GHz Amplifiers. They are necessary to bring the data signal and the LO signal to the needed power levels. • Power Splitter. All the necessary passive components were designed and fabricated. They are, in modern microwave techniques, straightforward to implement and will not be presented. • Power Amplifier (PA). The notation PA indicates the amplifying chain needed to rise the power of the V-band mixer output signal up to 10 dBm necessary for transmission. These circuits were not developed during the project. • Antenna. Antennae were only briefly investigated during the project, therefore no result will be presented.

Conclusions

2.3

25

Conclusions

In this chapter, a feasible system architecture for a millimeterwave fiber-optic communication system was presented and realistic specification for the system hub were derived. Remote generation of the mm-wave local oscillator through frequency multiplication in the hub is proposed as the best way to generate a stable high-frequency LO signal and contemporaneously overcome the problems due to chromatic dispersion. A two-step signal up-conversion in the hub is suggested instead of a direct up-conversion in order to relax the mixer specifications in terms of LO- and image-rejection and allow the frequency conversion circuits to be fabricated on our InP HEMT MMIC process. In order to properly set the hub specification the fiber-optic and the radio links were investigated. For the radio link the limitations on the transmitted power due to the current regulations on electromagnetic emission were taken into account. The analysis of the fiber optic link leads to extremely loose specifications on the hub optical front-end in terms of noise figure. Non-linear effects were, anyway, not considered. The radio link investigation indicates that a transmitted power between 5 and 15 dBm ensures a reliable transmission in a 10 m radius and does not violate law limitations.

3 Process Technology This chapter describes the InP HEMT process technology employed at the Microwave Electronics Laboratory of the Swiss Federal Institute of Technology in Zurich. The first section introduces the physics of the high electron mobility transistor and gives an overview of the different material systems used in HEMT technology. In the second section, the MMICs fabrication process is illustrated, while the last part of the chapter presents the characteristics of the active (HEMTs) and passive (transmission lines, resistors, capacitors, inductors) components available on the process. Details about passive components modeling for circuit simulations are also given in the last section, while InP HEMTs modeling will be specifically addressed in the next chapter. 3.1 The High Electron Mobility Transistor 3.1.1 From the two-dimensional electron gas to InP HEMTs Since 1978, when the two-dimensional electron gas was discovered in GaAs at the n-AlGaAs/GaAs heterointerface [30], and thanks to the development of molecular beam epitaxy (MBE) growing techniques for III-V compound semiconductors, many different heterostructure field-effect transistors with increasing performance have been demonstrated. The first heterostructure FETs were fabricated by research groups in the United States [31,32], in Japan [33], and in Europe [34]. These devices were given various names: high electron mobility transistor

28

Process Technology

(HEMT), selectively doped heterojunction transistor (SDHT), modulation-doped FET (MODFET), two-dimensional electron gas FET (TEGFET). Ketterson [35] was the first to introduce in 1985 a pseudomorphic InGaAs quantum well between the nAlGaAs/GaAs layers. InGaAs has higher carrier mobility and superior saturation velocity than GaAs; this resulted in a device with improved power-gain and noise performance at high frequencies, better 1/f noise and less generation-recombination noise [36]. A couple of years later researchers from different laboratories [37–39], taking advantage of the excellent transport characteristics of the InGaAs layer, started to use the latticematched InGaAs quantum well in the AlInAs/InGaAs structure on an InP substrate. These efforts resulted in one of the most promising high-speed technologies based on III-V compound semiconductors. 3.1.2 Physical principles of heterostructures The basic idea behind HEMTs is creating a 2-DEG in the device channel. In a 2-DEG, free channel electrons are physically separated from the ionized donors, reducing ionized impurity scattering and consequently enhancing electron mobility. A 2DEG can be obtained when two semiconductors with different bandgap energies EG,1 and EG,2 and unequal electron affinities χ1 and χ2 are interfaced. At thermal equilibrium, the Fermi levels of the two materials align. If the difference between the bandgap energies is greater than the difference between the electron affinities (EG,1 − EG,2 > χ1 − χ2 ), a discontinuity in the conduction band appears at the heterointerface and the conduction band of the small-bandgap semiconductor bends below the Fermi level creating a triangular potential well. Electrons are transferred from the material with the highest conduction band energy into the potential well and, as illustrated by Fig. 3.1, a 2-DEG is created. The electron density in the 2DEG can be increased by selectively doping the material with the higher EC , from which the free electrons are donated; this is called doping modulation. In this way a greater separation

The High Electron Mobility Transistor

29

between electrons and donors is achieved, allowing the 2-DEG to suffer less from Coulomb scattering and to enjoy an higher mobility. In order to achieve an even higher electron mobility, the distance between the donors and the 2-DEG can be further increased by means of an undoped spacer layer. This makes the mobility of free channel electrons greater than in bulk materials. The great transport properties of 2-DEG can be exploited to fabricate devices excelling for low-noise and high-frequency performances. 3.1.3

HEMT Material Systems

The availability of good substrate materials and the quality of the epitaxial layers that form the active region of a device are indispensable ingredients for the successful fabrication of heterostructures and, thus, of HEMTs. The increased substrate quality (better orientation mismatch, less defect density) and the advances in Molecular Beam Epitaxy (MBE) techniques (increased uniformity, better material control) are responsible for the enormous progresses of the last decade. High electron mobility transistors can be fabricated using various materials. The first HEMTs used a AlGaAs/GaAs heterojunction on a GaAs substrate, which is easy to fabricate because of

Fig. 3.1 Heterojunction band diagram. ∆EC = χ1 − χ2

30

Process Technology

Fig. 3.2 Bandgap energy versus lattice constant for various compound semiconductors.

the perfect lattice match between AlGaAs and GaAs. Nevertheless, the GaAs channel has a limited electron mobility and of a poor electron confinement. An valid alternative is the Inx Al1−x As/Inx Ga1−x As heterojunction. Both electron mobility and electron confinement in the potential well increase with an increasing Indium content in the channel. However, on a GaAs substrate, the strong lattice mismatch between InGaAs and GaAs entangles the growth of good quality layers. The limitations of GaAs substrates were circumvented with the introduction of metamorphic structures [40, 41] in the late nineties. The idea is to distribute the strains deriving from the lattice mismatch in a thick buffer layer with an increasing lattice constant grown on the GaAs substrate. High-quality heterostructures can be subsequently built on top of the buffer layer. This technique is, unfortunately, still very difficult to domesticate using MBE. Lattice match and high Indium content in the channel can be simultaneously obtained using InP as substrate material. As shown in Fig. 3.2, the Indium content in both InAlAs and InGaAs can be chosen so that the lattice constants

The High Electron Mobility Transistor

31

Fig. 3.3 HEMT Structure

of the two layers is matched to that of InP (In0.52 Al0.48 As, In0.53 Ga0.47 As), thus allowing the growth of lattice matched devices on an InP substrate. In this way, room-temperature mobilities up to 12000 cm2 /Vs and electron densities up to 3.5 × 1012 cm2 can be obtained. 3.1.4 Device Structure In Fig. 3.3 the HEMT layer structure used in this work is depicted. The active layers are grown using molecular beam epitaxy. The layer geometry and composition were optimized for both high frequency and low noise applications [42, 43]. The substrate is Fe-doped semi-insulating InP with a thickness of 600 µm, on top of which an InAlAs buffer layer with a thickness of 300 nm is grown. Its function is to compensate for the surface roughness of the substrate and to allow the growth of high-quality active layers. The heterojunction is formed by a 50 nm thick In0.53 Ga0.47 As channel, followed by a 10 nm thick In0.52 Al0.48 As spacer and by a 15 nm thick Schottky layer of the same material. Between the spacer and the Schottky layer a Silicon δ-doping plane (2 × 1012 cm−2 ) supplies the channel with additional electrons. The top layer is a n+ Silicon-doped In0.53 Ga0.47 As cap layer which protects the Aluminum-rich layers from oxidation and allows the formation of ohmic contacts

32

Process Technology

with the device channel. The next section explains how, starting from this epi-structure, InP HEMT integrated circuits are fabricated. 3.2 Process Description A good knowledge of the technology available to the circuit engineer is indispensable to fully understand the design of a given circuit. This section contains a brief description of the InP HEMT MMIC process developed in the clean room facilities of the Laboratory for Electromagnetic Fields and Microwave Electronics. The complete fabrication process can be summarized in the following steps, as schematically illustrated by Fig. 3.4. 1. The device active layers are grown using molecular beam epitaxy on a semi-insulating Fe-doped InP substrate. The epitaxial wafers used in this work were bought from commercial suppliers [44]. 2. The wafer is diced into several 8.5 × 8.0 mm2 chips. Each chip is processed separately. 3. Ohmic contacts are defined. Ge (17 nm), Au (48 nm), Ni (10 nm), and Au (200 nm) are first evaporated and, after lift off, annealed for two minutes at 300 ◦ C in N2 atmosphere. The ohmic contacts are patterned using an image reversal resist. 4. Isolation between the devices is obtained by mesa etching. The active layers are etched down to the buffer layer using a H3 PO4 :H2 O2 :H2 O 1:1:5 solution. The successful device isolation is verified with I-V on-chip measurements. 5. The gate-recess of the device is etched in order to form a Schottky contact between the gate and the channel. First, the T-shaped gate is patterned with electron beam lithography using a three-layers resist stack. After resist development, the InGaAs cap layer is etched using an highly selective succinic acid solution. The same solution etches from the side the InGaAs channel which remains exposed

Process Description

33

Fig. 3.4 Schematic representation of the used InP HEMT MMIC fabrication process.

34

6.

7.

8.

9.

10.

11. 12.

Process Technology

after device isolation. This prevents contacts between the gate metallization and the channel, thus reducing the gate leakage current [45]. The gate metal (20 nm Ti, 250 nm Au) is formed. Titanium is used to prevent Gold diffusion as well as to improve the gate mechanical stability. The T-shaped gate metal allows the fabrication of very short gates with low parasitic gate resistance. Thin film resistors are first patterned using an image-reversal resist and then fabricated by evaporation of 23 nm Ti. The thickness of the evaporated layer is optimized for 50 Ω/ resistors. The first metal interconnect is defined using an imagereversal resist; the evaporated metallization consists of 20 nm Ti and 250 nm Au. A SiNx dielectric layer for simultaneous device passivation and MIM capacitors fabrication is deposited. This is done at Avalon Photonics [46] using plasma-enhanced chemical vapor deposition and reactive ion etching. The second metal interconnect is defined using an imagereversal resist; the evaporated metallization consists of 20 nm Ti and 250 nm Au. It is noticeable that this second metallization is everywhere coplanar with the first one, except in those areas where SiNx is deposited. The air bridge feet are defined using a 4 µm thick positive resist and 20 nm Ni evaporation. The air bridges are patterned with a 3 µm thick positive resist and then electroplated with Gold. The thickness of the electroplated Gold in 1 µm. Electroplating is also used to increase the metal thickness of passive components such as inductors and coplanar wave-guides.

The fabrication process of a complete MMIC is performed in approximately two weeks. Figure 3.5 shows a detail of a fabricated MMICs, where different circuit components such as

Components

35

Fig. 3.5 Detail of a fabricated MMICs. Various circuit components like HEMTs, capacitors, transmission lines etc. can be identified.

HEMTS, capacitors, transmission lines etc. can be recognized. The characteristics both active and passive components available on this process will be described in the next section. 3.3 Components This part of the chapter describes the different circuit components that can be implemented with the process illustrated in the previous section. Circuit design-oriented modeling of passive components is also presented and comparisons between measurement and simulations are shown. Large-signal modeling of InP HEMTs will be extensively treated in the next chapter. 3.3.1

InP HEMTs

The standard HEMT device used in this work can be seen in Fig. 3.5. It is a two finger transistor with a gate length of 0.2 µm and a total gate width of 2×75 µm. The HEMT has a “butterfly” layout, which presents two important advantages. First, the low parasitic effects simplify high-frequency measurements,

36

Process Technology 400 350

Id (mA/mm)

300 250 200 150 100 50 0 0

0.5

1

V

ds

(V)

1.5

2

2.5

Fig. 3.6 I-V characteristics of the typical HEMT device. Vgs ranges from −1 V to 0.5 V with 50 mV steps.

and, consequently, full characterization at millimeter-wave frequencies. Additionally, the symmetric layout can be easily interfaced with transmission lines in a coplanar environment, especially in those circuits where a common-source configuration is needed. Typical I-V curves for these devices are depicted in Fig. 3.6. The maximum saturated drain current lies around 350 mA/mm, the maximum transconductance is 580 mS/mm, and the typical threshold voltage VT is −0.5 V. The I-V characteristics indicate a perfect pinch-off and a good saturation, with a low DC output conductance. This is an indication of the good electrical qualities of the process. Figures 3.7 and 3.8 show the extraction of the high-frequency figures of merits, i.e. the transit frequency fT and the maximum frequency of oscillation f max. The transit frequency is the frequency at which the current gain h21 is unity. The maximum frequency of oscillation is defined as the frequency where the maximum available gain (MAG) or the Manson’s unilateral power gain (U) become unity. In this work we chose to extract fmax from the Manson’s unilateral power gain, but it

Components

37

can be easily demonstrated that both the MAG and U become unity at the same frequency [47]. Both fT and fmax are extrapolated using a tangent with a −20 dB/decade slope. An fT of 135 GHz and an fmax of 200 GHz are obtained. The extracted values of the high-frequency figures of merit attest the device suitability for V-band application. Another important figure of merit is the minimum noise figure (N Fmin ). Figure 3.9 shows N Fmin of a typical device as a function of frequency. The device is biased in the saturation region and the bias drain current is 10 mA. The low-noise characteristics of these devices are excellent and comparable to the best reported literature results. Cryogenically-cooled low-noise amplifiers with record N Fmin and noise temperature where demonstrated using this process [48]. 3.3.2

Transmission Lines

Transmission lines are vital components of any analog millimeterwave circuit. While at microwave frequencies the possibility of 50 45 40 35

h21 (dB)

30 25 20 15 10 5 0 8 10

9

10

10

10

11

10

Frequency (Hz)

Fig. 3.7 Current gain h21 of a typical device and its associated fT , extracted using a −20 dB/decade tangent.

38

Process Technology 50

Unilateral Power Gain (dB)

45 40 35 30 25 20 15 10 5 0 9 10

10

11

10

10

Frequency (Hz)

Fig. 3.8 Manson’s unilateral power gain U of a typical device and its associated fmax , extracted using a −20 dB/decade tangent.

integrating lumped elements often represents an efficient al1.3 1.2 1.1

F

min

(dB)

1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 2

4

6

8

10

12

14

16

18

20

Frequency (GHz)

Fig. 3.9 Minimum noise figure as a function of frequency for a typical device. The transistor is in saturation region and the bias drain current is 10 mA.

Components

39

ternative to transmission lines, no other option is available at millimeter-wave frequencies. In this work, coplanar waveguides (CPW) are used, which were first proposed by Wen in 1969 [49]. A schematic representation of the cross section of a CPW is shown in Fig. 3.10. This structure supports quasi-TEM propagation up to a certain frequency, which depends on the groundto-ground spacing. A rule of thumb is given by: d < λmax /10,

(3.1)

where λmax is the wavelength along the line of the maximum propagation frequency and d is the ground-to-ground spacing [50]. Coplanar waveguides were preferred to microstrip lines for various reasons. In CPW technology, no wafer thinning, backside metallization, and via holes through the InP substrate are needed. This reduces significantly the needed fabrication steps. Additionally, a via hole to ground introduces a parasitic inductance, which can be avoided with the presence of a coplanar ground on top of the substrate. Another benefit of coplanar waveguides, compared to microstrip lines, is the fact that their dimensions are not univocally defined for a given impedance. This introduces an element of flexibility that can be exploited by the circuit designer. CPWs can be easily flip-chip bonded on a coplanar substrate, which results in better signal transitions between the mounting substrate and the integrated circuit. More details on flip-chip bonding of coplanar circuits will be given in the last chapter. Employing coplanar waveguides presents also some drawbacks. Losses are generally higher than in microstrip lines. If the ground potential on both sides of the center conductor is not equal, asymmetric modes might start to propagate in the CPW. This can happen when the travelling signal reaches a discontinuity like a T-junction, a cross-junction, or a 90◦ bend. In order to avoid asymmetric modes propagation, the ground planes have to be short-circuited both in proximity of a discontinuity and

40

Process Technology

Fig. 3.10 Schematic cross-section of a coplanar waveguide.

along long (> λ/4) transmission lines. The lack of models in commercial circuit simulators represents, from a designer’s point of view, a significant disadvantage. Modelling of CPW discontinuities will be addressed in the next section. In order to guarantee quasi-TEM propagation up to the V-band and further, a ground-to-ground spacing of 90 µm was chosen in this work. By changing the ratio between the center conductor width w and the gap g, the line impedance can be varied in a range between 30 and 70 Ω. In order to reduce propagation losses, the thickness of the center conductor was increased from 0.26 to 0.8 µm using electroplating over the first level of metal. Agilent LineCalc was used to determine the line dimensions for different impedances, and the CPW model implemented in Agilent ADS was used to model straight transmission lines. 3.3.3 Line Discontinuities In order to perform reliable simulations of coplanar circuits at millimeter-wave frequencies, it is mandatory to carefully consider the effects of CPW discontinuities such as T-junctions, cross-junctions, 90◦ bends, short circuits, and open circuits. Figure 3.11 shows some fabricated CPW discontinuities. As already mentioned, the simulation of coplanar circuits is complicated by the absence, in commercial circuit simulators, of reliable models for CPW discontinuities. A possible way to circumvent this problem is the approach presented in this section [42]. With the help of a “3d-planar” [51] electromagnetic simulator the S-parameters of the discontinuities are calculated over a broad range of frequencies. Since the simulated struc-

Components

41

tures have small sizes, the computational effort is limited. The S-parameters of the modeled structures are then imported as data files into the circuit simulator. This method has the advantage of being simple and flexible, but requires some verifications. In order to prove the correctness of the simulations, test structures were fabricated and then measured. Figure 3.12 shows the measured and simulated S-parameters of a 1.3 mm folded CPW with four consecutive 90◦ bends and ending in an open circuit. Both the simulated magnitude and phase of the input return loss are in excellent agreement with the measured data. Another example is illustrated in Fig. 3.13, which contains the measured and simulated S-parameters of a 48 GHz amplifier containing T-junctions, cross-junctions, open circuit stubs, and short circuit stubs in its matching networks. In the simulation, the measured S-parameters of the active device were used. The agreement is excellent, and the frequency difference between the measured and simulated optimum return losses is only 2% (49 GHz instead of 48 GHz).

Fig. 3.11 CPW discontinuities. (a) 90◦ bend, (b) T-junction, (c) cross-junction, (d) air bridge along a line.

42

Process Technology 0

S11 (dB)

−0.5 −1

−1.5 −2 −2.5 0

10

20

30

40

50

60

70

80

60

70

80

Frequency (GHz) Phase S11 (deg)

200 100 0

−100 −200 0

10

20

30

40

50

Frequency (GHz)

Fig. 3.12 Measured (continuous line) and simulated (dashed line) S-parameters of a 1.3 mm long folded CPW presenting four consecutive 90◦ bends and ending in an open circuit.

10 5

S−parameters (dB)

0 −5

−10 −15 −20 −25 −30 30

35

40

45

50

55

60

Frequency (GHz)

Fig. 3.13 Measured (symbols) and simulated (continuous line) S-parameters of a 48 GHz amplifier presenting various T-junctions, cross-junctions, and open circuit stubs.

Components

43

Phase S21 (deg)

200 100 0

−100

S11, S21 (dB)

−200 0

10

20

10

20

30

40

50

60

70

80

30

40

50

60

70

80

Frequency (GHz)

0

−10 −20 −30 −40 0

Frequency (GHz)

Fig. 3.14 Measured (continuous line) and simulated (dashed line) S-parameters of a 1 mm long CPW crossed by 11 air bridges with regular spacing of 100 µm between each other.

A different approach was used for air bridges. With the process described in section 3.2, only air bridges with a maximum length of 50 µm can be reliably fabricated. Since the ground to ground spacing of the typical CPW is 90 µm, i.e. to wide for an air bridge, the first level of metallization is used to short circuit the ground planes of the coplanar waveguides and the air bridge is fabricated on the center conductor, as it can be seen in Fig. 3.11. Air bridge effects at T-junctions, cross-junctions, and 90◦ bends are included in the electromagnetic simulations, but air bridges along a straight line have to be taken separately into account. A capacitor to ground was found to be a suitable model; 3 fF is the typical capacitance value for an air bridge on a 50 Ω transmission line with 90 µm ground to ground spacing. Figure 3.14 shows the measured and simulated S-parameters of a 1 mm long coplanar waveguide crossed by 11 regularly spaced air bridges. The good agreement between measurements and simulations confirms the validity of the modeling approach.

44

Process Technology

3.3.4 Thin Film Resistors Thin film resistors are fabricated by evaporation of a Titanium layer with a thickness of 23 nm. The resistive layer has a resistance of 50 Ω/ with a typical variation of 3% over the same chip and less than 10% tolerance over different chips. It is worth mentioning that, in order to protect the Titanium layer during the RIE step performed after dielectric deposition, the thin film resistors are protected by a SiNx passivation layer. When drawing the layout, the designer must ensure that the contacts between resistors and metal are done using the first metallization layer. If the resistors are covered with passivation, in fact, it is not possible to contact them with the second metallization layer. 3.3.5 MIM Capacitors On this process, metal-insulator-metal (MIM) capacitors are fabricated using the first metallization level as bottom plate, a 120 nm thick SiNx layer as dielectric and the second metallization layer as top plate. In this way, capacitors with a capacitance of 0.5 fF/µm2 can be fabricated. For reasons ranging from mask tolerance to alignment problems and lithography resolution issues, the fabrication of MIM capacitors with capacity values lower than 20 fF is not recommended. Inter-digital coplanar structures are better suited for this purpose. The process tolerance, which depends on the variations of the dielectric thickness and dielectric constant is less than 10% for different chips, and less than 3% on the same chip [52]. 3.3.6 Spiral Inductors At Ku-band frequencies and lower, the large physical dimensions of transmission lines make their employment in the fabrication of passive structures not practical anymore. If a small circuit size is needed, the use of lumped inductors becomes necessary. During this work, spiral inductors with different sizes and geometries were designed, fabricated, and characterized. Tables 3.1 and 3.2 provide an overview of the fabricated inductors with

Components

45

Table 3.1 Square spiral inductors fabricated with the InP HEMT process. Conductor width and conductor spacing is 10 µm. Metallization thickness is 0.8 µm.

Type

Inductance (nH)

DC Resistance (Ω)

Qmax

fsr (GHz)

L250 L600 L1000 L1300 L2000 L3500

0.35 0.75 1.04 1.2 1.9 3.9

3.2 4.2 4.86 4.8 6 10.6

13.3 9.8 9.8 8.76 10.05 7.09

> 40 36 32.4 25.6 21.5 10.7

Table 3.2 Circular spiral inductors fabricated with the InP HEMT process. Conductor width and conductor spacing is 10 µm. Metallization thickness is 0.8 µm.

Type

Inductance (nH)

DC Resistance (Ω)

Qmax

fsr (GHz)

L530 L750 L1010 L1310 L1550 L1860 L2020 L2630 L3200 L3360

0.65 0.9 1.3 1.6 2.1 2.4 2.6 3.6 4 4.3

4.2 4.6 5.4 6.2 7.2 7.7 8.3 7.9 10.7 11.3

10.6 10 9.5 8.8 8.6 8.6 8.5 9.9 7.5 7.6

40 31 24.7 22.3 18.2 16.6 15.6 13 11.7 11.2

their most important parameters, i.e. inductance value, DC resistance, quality factor, and self-resonance frequency fsr . Two inductor families were developed, one with a square spiral geometry and the other with a round spiral geometry. In both cases, the conductor width is 10 µm, the conductor spacing is 10 µm, and the metallization thickness is 0.8 µm. Their inductance ranges from 0.35 to 4.3 nH, and the quality factor Q varies between 7 and 13. The low quality factor is due to the limited metallization thickness, which results in a high series resistance and, thus, in high losses. Nevertheless, the developed inductors

46

Process Technology

were successfully employed in circuit fabrication. No inductor model was developed; the measured S-parameter were directly imported in the circuit simulator as data files.

4 InP HEMT Large Signal Model The development of the millimeter-wave fiber-optic communication system proposed in the second chapter embraces the design and the fabrication of a variety of linear and non-linear circuits with operating frequencies ranging from a few GHz to the V-band. It is, therefore, important to carefully model the linear and the non-linear characteristics of the employed devices both in the lower GHz range and at millimeter-wave frequencies. For InP HEMTs, one of the most important effects at low frequencies is the so called kink effect related to impact ionization [53]. Impact ionization results in high gate leakage current, high output conductance, and low breakdown voltage. Due to the smaller bandgap of the InGaAs channel of InP HEMTs, this phenomenon occurs at lower drain-source voltages compared to GaAs-based devices. Its effects, in fact, can be observed at operating conditions typical for non-linear circuits and should, therefore, be taken into account by large-signal models. In the framework of the MFOCS project, an efficient method to extract a large-signal look-up table based model for InP HEMTs was developed that takes impact ionization into account [54]. The large-signal model has been verified with linear measurements up to the W-band and with non-linear measurements up to the V-band. This chapter is organized as follows; the first section gives

48

InP HEMT Large Signal Model

an overview of the different modeling approaches and briefly presents the state of the art of large-signal FET modeling. In the second section, the used small-signal and large-signal circuit topologies are introduced and discussed. The third section describes the extraction of the small-signal model parameters and their integration in the large-signal model. Then, small-signal and large-signal model validation is addressed and comparisons between measured and simulated large-signal performances of fabricated MMICs are presented. Finally, a discussion on the limitations of look-up table based models concludes the chapter. 4.1

Large-Signal Modeling of High-Frequency FETs

There are three possible approaches to high-frequency modeling of FET devices. Each of them has different advantages and disadvantages, and the choice of a specific approach strongly depends on the application the model is needed for. 1. Physical models describe the device behaviour in terms of carrier transport properties, device geometry, and material characteristics. This kind of model usually provides a deep understanding of the device, but is inherently complex and often requires sophisticated numerical methods to converge to a solution. Physical models can be efficiently used for device optimization, but, at the moment, they are not suited for circuit simulation. 2. Black box models are the conceptual opposite of physical models, since they try to represent a device with an input-output transfer function, without giving any insight on their nature. The mathematical description of the transfer function relies on model parameter which are extracted by fitting the measurement results. Parameter fitting often results in an arduous optimization problem, which makes the extraction of black box models rather complicated. 3. Small-signal equivalent circuit based models are the most used for circuit simulations. In this kind of model, the

Large-Signal Modeling of High-Frequency FETs

49

device behaviour is described by means of lumped circuit elements, which are extracted from DC and S-parameter measurements. If the small-signal equivalent topology is properly chosen, this approach is physically meaningful, accurate over a wide range of frequencies, and easy to implement in a circuit simulator. Since the large-signal model described in this work was needed only for circuit simulations, it was mandatory to choose a smallsignal equivalent circuit based model. Among the many possible implementations of this kind of model, two main families can be recognized, which differentiate themselves depending on how the constitutive relations of the large-signal model are expressed. In empirical models, the large-signal behavior of the device is described by means of analytical functions of the terminal voltages. These functions must fulfill various conditions and are usually fitted on the small-signal parameters extracted from the measured S-parameters. Empirical models need a small amount of measurements for their extraction 1 , but require an effort in choosing the most appropriate analytical functions and in fitting them with the small signal parameters. Empirical models have the advantage of allowing reliable device simulations also outside the measurement space, but show a limited adaptability to changes in the device due, for example, to technology variations. In a look-up table based model the constitutive relations are tabulated as a function of the terminal voltages. During nonlinear simulation the table files are accessed and their values interpolated. The large-signal constitutive relations are generated by numeric integration of the small-signal circuit elements extracted from S-parameter measurements. In order to guarantee a good model accuracy, the interpolation errors must be kept as low as possible. This means that the number of S-parameter 1 Normally, for the extraction of an empirical based model, it is sufficient to perform S-parameter measurements at 30 different bias points.

50

InP HEMT Large Signal Model

measurements needed for the extraction of such a model is generally an order of magnitude higher than for empirical models. An advantage of look-up table based models is their ability of being easily adaptable to device variations, given the invariance of the small signal topology. The choice of a table based model is, therefore, the most appropriate for a technology aimed to device optimization and consequently open to process modifications. Many works on high-frequency FET device modeling have already been reported. Root presented one of the the first lookup table based GaAs FET large-signal models in 1991 [55], but limited its validation to microwave frequencies. In 1999, as a refinement of Root’s work, Wei presented a table-based model with better S-parameters prediction capability [56]; this model was also verified only up to microwave frequencies. A MODFET small-signal equivalent circuit validated up to 120 GHz was presented by Tasker in 1995 [57], while Wood and Root published a bias-dependent scalable model validated up to the same frequency in 2000 [58]. Small-signal and noise behavior of impact ionization were modeled by Reuter in 1997, but were not given a large-signal representation. A large-signal model validated up to the V-band with non-linear measurements and up to 118 GHz with linear measurements was presented by FernandezBarciela in 2000, but this model addresses GaAs FETs and, therefore, does not take impact ionization into account [59]. 4.2 Small-Signal and Large-Signal Equivalent Circuit The devices used for model extraction are two-fingers InP HEMTs with 0.2 µm gate length, 75 µm gate-finger width, 135 GHz fT , and 200 GHz fmax . The fabrication process and the most important characteristics of these devices were presented in the third chapter. The modeling procedure is an extension of the method presented in [60], developed at the K.U.Leuven, and successfully used for GaAs PHEMTs. In [60], where only experimental results up to microwave frequencies are shown, a

Small-Signal and Large-Signal Equivalent Circuit

51

non-quasi-static small-signal model and its corresponding largesignal model are presented, where in the small-signal equivalent circuit the feedback elements between the gate and the drain terminals are eliminated and replaced with transelements. This has the advantage of allowing the small-signal topology to be consistent with the large-signal equivalent circuit, which helps to respect charge conservation. For mm-wave effects and impact ionization to be included into the model, some modifications have to be made both to the small-signal and to the largesignal equivalent schemes. The small-signal equivalent circuit is depicted in Fig. 4.1. At mm-wave frequencies the electrical length of gate and drain metallizations becomes relevant and transmission-line effects need to be included. This is achieved by splitting the extrinsic gate and drain capacitances Cpg and Cpd in two [57], [58]. Because of the device layout, distributed effects on the source metallization can be ignored. Since Cpg and Cpd are bias-independent extrinsic elements, this splitting does not influence the large-signal equivalent scheme. In order to model impact ionization effects, an extra network is added at the drain side of the small-signal equivalent circuit [61]. This network consists of a resistance Rim in series with the parallel connection of a capacitance Cim and a transconductance gim controlled by the intrinsic drain-gate voltage Vdgi . The used large-signal intrinsic representation of the transistor is shown in Fig. 4.2. It consists of the parallel connection of a charge source and a current source at the drain side and of a current source in parallel with the series connection of a charge source and a resistor at the gate side. This large-signal equivalent circuit is consistent with the intrinsic small-signal equivalent scheme already introduced. To evaluate the effects of impact ionization in non-linear circuits, it is necessary to find a large-signal representation of the small-signal network introduced to model it. From experimental results, it was found that impact ionization can be best modeled by means of a first order dispersive network. This is somehow similar to the modeling

52

-

gate Lg

Rg

+

Cpg /2

V’gsi

+ Cgi

source

drain gimVdgi

jwCdmVdsi

-

+

Vdsi

Cds

(Gm+jwCm)Vgsi

Rs

Rd

Ld

Cim R ds

Rgsf Ri

Ls

+

Rgdf

Vgsi Cpg /2

Vdgi

Rim

-

Cpd /2

Cpd /2

InP HEMT Large Signal Model

Fig. 4.1 Small-signal equivalent circuit.

intrinsic

drain

Cgi(Vgsi,Vdsi) dVgsi/dt Cdm(Vgsi,Vdsi) dVdsi/dt Vgsi Igs(V’gsi,Vdsi) V’gsi

Cm(Vgsi,Vdsi) dVgsi/dt + Cds(Vgsi,Vdsi) dVdsi/dt

IdsDC(V’gsi,Vdsi)/(1+jwtLF)+ jwtLF/(1+jwtLF)(IdsLF(Vgsi,Vdsi)/(1+jwtio)+ jwtio/(1+jwtio)IdsHF(Vgsi,Vdsi)) Vdsi

Ri(Vgsi,Vdsi)

source

Small-Signal and Large-Signal Equivalent Circuit

Fig. 4.2 Large-signal equivalent circuit.

gate

53

54

InP HEMT Large Signal Model

approach of the dispersive behavior of Gm and Gds in the kHzMHz range [62]. The bias-dependent values of Gm and Gds are extracted twice, once at frequencies lower than those of interest for impact ionization (250 MHz-0.5 GHz), the second time at frequencies where impact ionization does not influence the S-parameters anymore (7-110 GHz). It must be noticed that at 250 MHz, which is the lowest measurement frequency, the low-frequency output dispersion is no longer effective; the only dispersion effect is the one originated by impact ionization. The low frequency drain-source current IdsLF and the high frequency drain current IdsHF are obtained from the integration of Gm and Gds towards their intrinsic terminal voltages. Another point worth noticing is that the charges are not obtained from the numerical integration of the small-signal capacitances, but are expressed as analytical functions of the bias-dependent capacitances themselves, as already successfully demonstrated in [63, 64]. At the gate side this is mathematically equivalent since terminal charge conservation is guaranteed, nevertheless one should be aware of possible peculiarities. The average capacitive current, in fact, might not be zero, leading to non conservation of energy [65]. At the drain side terminal charge conservation is not necessarily guaranteed since two capacitances with two different physical meanings are used; Cds is a spacecharge capacitance, Cm is related to the intrinsic time delay τ . It is important to underline that terminal charge conservation is different from physical charge conservation, which is always valid and expressed in circuits theory by Kirchhoff’s Current Law. Terminal charge conservation is related to the bias dependence of different capacitance functions attached to a single node, and implies that all the capacitances attached to a given terminal can be uniquely described by the partial derivatives of a single charge function. The choice of a large-signal model formulation which respects terminal charge conservation is simply a modeling choice, which is needed to avoid possible consequences, such as a DC current spectral component under

Model Extraction

55

large-signal conditions [66]. Kirchhoff’s laws are always valid in a large-signal model, charge conservation not necessarily. The large-signal equivalent circuit is complete after adding another first order transfer function which regulates the transition between the measured DC current IdsDC and the low frequency current IdsLF [62], thus modeling the low frequency output dispersion. In the next section it will be shown how the smallsignal and the large-signal model parameters are extracted from the measurements. 4.3

Model Extraction

The first step of the modeling procedure is the extraction of the bias-independent extrinsic parameters. This is done from cold (Vds = 0) measurements using the method described in [67]. The main difference between this method and the classic Dambrine approach [68] lies in the extrinsic resistances extraction. The maximum positive value of the gate-source voltage Vgs is set to be just below the turn-on of the Schottky gate diode and the asymptotic behavior of Re(Z22 ) for Rch −→ 0 is calculated using Pl´a’s approximation [69]. In this way, the extrinsic resistances can be calculated while the gate-current density is kept low enough to avoid a possible device degradation. Then, the device S-parameters are measured over a grid of more than 900 different bias points. Here it is worth pointing out that in a look-up table based large-signal model the capability of predicting high order harmonics is limited by the precision of the interpolation between those points where the model is actually defined. Since the most important non-linearity is due to the drain-source current, it was decided to use variable steps for Vgs and Vds and have a tighter grid where the drain-source current changes more drastically, i.e. close to the threshold voltage for Vgs and in the linear region for Vds . In order to obtain a good accuracy up to mm-wave frequencies, it is not enough to measure the S-parameters at only one

56

InP HEMT Large Signal Model

frequency point [55]. In this case it would be also impossible to have any information about the impact ionization effects, which are strongly frequency dependent. On the other hand, in order to achieve a sufficient resolution in the low frequency range with a linear sweep, at least 201 frequency points would be necessary. This would generate a huge amount of data, not to mention the long time required by the measurements. What is needed is a good resolution at low frequencies, to extract the impact ionization related parameters, and only few points in the mm-wave frequency range, which are sufficient to extract the other parameters. Therefore, we chose to use a logarithmic frequency sweep that reaches the goal using only 59 points. This is important because it improves the measurement speed and drastically reduces the computation time required for the small-signal parameters extraction. After measurements, the bias-independent extrinsic parameters are de-embedded from S-parameters. At this point, the correct determination of the extrinsic parameters is crucial for a successful model extraction. Small errors in the extrinsic parameters extraction, in fact, can cause large inaccuracies of the de-embedded intrinsic parameters, which might result in an invalid model. The intrinsic small-signal circuit elements are calculated according to the following expressions. Gds =