Modeling and Investigation of a Geometrically Complex ... - OSU ECE

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004

1983

Modeling and Investigation of a Geometrically Complex UWB GPR Antenna Using FDTD Kwan-Ho Lee, Student Member, IEEE, Chi-Chih Chen, Member, IEEE, Fernando L. Teixeira, Member, IEEE, and Robert Lee, Member, IEEE

Abstract—A detailed analysis of ultrawide-band (UWB), dual-polarized, dielectric-loaded horn-fed bow-tie (HFB) antennas is carried out using the finite-difference time-domain (FDTD) method. The FDTD model includes realistic features of the antenna structure such as the feeding cables, wave launchers, dielectric loading, and resistive-film loading. Important antenna characteristics that are usually difficult to obtain via measurements can be obtained more directly from this FDTD model. Since the HFB antennas under consideration are intended for ground penetrating radar (GPR) applications, the effects of the half-space medium are also investigated. The simulated results serve to verify the performance of the HFB antenna design, and to optimize various antenna parameters. Index Terms—Bow-tie antenna, coaxial cable, dielectric loading, finite-difference time-domain (FDTD), ground penetrating radar (GPR), impedance, resistive, ultrawide-band (UWB).

I. INTRODUCTION

G

ROUND PENETRATING RADAR (GPR) find applications in many areas such as geophysical prospecting, archeology, civil engineering, environmental engineering, and defense technologies as a noninvasive sensing tool [1], [2]. One key component in any GPR system is the receiver/transmitter antenna(s). Desirable features for GPR antennas include broadband operation, good impedance matching, and small size. The frequency range of a GPR antenna is determined by the particular application and its relation to the nature of the target, soil constitution, desired depth of penetration, and inversion/classification method being used. For example, the frequency of operation for detection and classification of anti-tank and anti-personnel landmines is usually from 0.1 to 1 GHz [3] and from 1 to 6 GHz [4], [5], respectively. A good frequency range for detecting 6-inch drainage pipes is found to be from 100 to 400 MHz. For unexploded ordnance (UXO) detection, the 10 800 MHz frequency range is often used [6], [7]. For the detection of shallow objects where high sensitivity is not an issue, elevated antennas are often used for easier scanning and better antenna calibration. In particular, many antennas used for detection of shallow landmines [5], [8], evaluation of integrity of concrete [9] and soil hardness [10] are all elevated Manuscript received December 2, 2002; revised November 3, 2003. This work was supported in part by the Department of Defense (DoD) Strategic Environmental Research and Development Program (SERDP) Project 1122 by the National Science Foundation (NSF) under Grant ECS-0347502. The authors are with ElectroScience Laboratory, Department of Electrical Engineering, The Ohio State University, Columbus OH, 43210 USA (e-mail: [email protected]). Digital Object Identifier 10.1109/TAP.2004.832501

systems that exhibits low antenna-ground interaction. On the other hand, most GPR antennas used for the detection of deep targets are operated very close to the ground so that most of the energy is radiated into the ground to improve sensitivity. This configuration also minimizes radiation into the air to comply with the FCC regulations. The characteristics of such GPR antennas while in field operation are usually difficult to determine a priori because of the large coupling with the environment. For instance, the input impedance of the commonly used dipoles or flat bowtie dipoles are directly affected by the electrical property of the particular ground for antennas operated close to the surface. Moreover, the amount of energy coupled into the ground changes as the permittivity increases and hence the radiation patterns also depend of the soil permittivity [11]–[13]. Hence, one major disadvantage is that the antenna characteristics in the field become dependent on the electrical properties of the ground and surroundings. This also makes calibration more difficult. In order to make antenna characteristics less susceptible to ground characteristics, a new dielectric-loaded horn-fed bowtie (HFB) antenna design was introduced in [7]. The HFB antenna was designed to minimize the antenna ringing by: 1) employing a stable and well matched surge impedance and 2) using specially designed tapered resistive loadings. Unlike most conventional antennas, the surge impedance was designed to be less dependent on the ground property because the feed point is elevated off the ground. Low loss dielectric material was then used to fill the space between the feed front and the ground surface to reduce ground-surface reflections and increase the electrical height of the feed. Both single-polarized and dual-polarized HFB antenna prototypes have been built and employed in actual applications. Due to its flexibility, the finite-difference time-domain (FDTD) method has been widely used in recent years for the numerical simulation of GPR systems [14]–[17]. Some of the previous studies have modeled GPR antennas as a series of point sources or short dipoles with or without the presence of conducting shields [18], [19]. In order to better characterize HFB antennas and to provide a more convenient tool for their design and optimization, a full-scale detailed three-dimensional (3-D) FDTD model of a dual-polarized HFB prototype was developed in this work and simulated for GPR applications. To reduce the computational cost, a special partition scheme [20] is adopted for the 3-D FDTD domain. This scheme divides the whole inhomogeneous region into several small homogeneous regions. In each homogeneous region, volumetric material property matrices are replaced by constants to save the

0018-926X/04$20.00 © 2004 IEEE

1984

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004

Fig. 1.

Prototype of HFB antenna.

memory. This partition scheme for modeling the electrically large HFB antenna in the presence of ground also allows for faster simulations on a personal computer. An anisotropic perfectly matched layer (APML) especially formulated for the dielectric or lossy half spaces [16], [21], [22] is implemented. This paper is organized as follows. The HFB antenna design is discussed in Section II. Section III describes the construction of the FDTD model for the dual-polarized HFB design and the performance of the resistive-film loading which is optimized for a given length. Section IV presents various HFB antenna characteristics obtained from the FDTD simulations. II. BASIC DUAL-POLARIZED HFB ANTENNA DESIGN Fig. 1 illustrates the basic structure of the dual-polarized UWB HFB antenna design. This is somewhat similar to a planar bowtie dipole with the feed point being raised off the ground. The feed section resembles that of a small transverse electromagnetic (TEM) horn except that it is filled with low loss dielectric material. Each antenna arm is smoothly curved in the transition from the horn section to the planar bowtie dipole section. The ends of the dipoles are terminated with tapered resistive cards (R-card) to reduce antenna ringing.

HFB prototype was constructed in-house using multiple layers of commercial window films [23]. These have various sheet resistance for different percentage of light transmission. When multiple films are overlaid properly together, one can obtain a desired resistivity profile with desired taper length. Fig. 1 illustrates how the tapered R-card was constructed for the HFB prototype. The objective of the resistive card is to reduce reflections by gradually dissipating the currents propagating toward the end of each antenna arm. This requires the resistivity on the R-card to be tapered from a small value to a large value along the antenna arm. An exponential taper of the resistivity was adopted in the HFB prototype with a tapering shown as follows: (1) where is the initial sheet resistance of the R-card at the perfect electric conducting (PEC)/R-card interis the sheet resistance at the far face, and is the length of the R-card, and end of the R-card, is the distance along the R-card from the PEC arm. B. Feed Section

A. Resistive Taper Section Tapered R-cards have many useful applications for radiation and scattering control [23]–[25], but commercial tapered R-cards are often expensive and have very limited choices of tapering profile and taper length. The R-card used in the

The feed section of HFB resembles a dual polarized TEM horn except that the end of each antenna arm is curved outward gradually to be connected to the flat bowtie section, and the internal space of the horn was filled with low loss dielectric material. The geometry of the horn and the antenna arms was

LEE et al.: MODELING AND INVESTIGATION OF A GEOMETRICALLY COMPLEX UWB GPR ANTENNA USING FDTD

Fig. 2.

1985

Dimension and computation domain partitioning of the fully polarimetric dielectric filled HFB antenna.

chosen based upon the tradeoff among the dielectric constant, size, weight, and cost. The objective was to obtain a surge impedance of 100 to match to the characteristic impedance of the feeding twin-coaxial cables shown at the bottom of Fig. 1 (each cable has a characteristic impedance of 50 ohms). Although tabulated characteristic (or surge) impedances for an infinite TEM horn with arbitrary geometry are available [26], [27], the exact impedance of a dual-polarization TEM horn with dielectric filling is complicated to obtain analytically. The experimental data obtained from [28] was used during the construction of HFB prototype. Note that the center of each coaxial cable was connected to one antenna arm and coaxial cable feed one polarization. each pair of the 50 A 0–180 broadband hybrid was used as a balun for each pair of cables. Accurate FDTD models recently constructed to calculate the surge impedance for such an antenna geometry are employed here [29]. The prototype to be analyzed here has a dielectric constant of 5. The plate angle of each antenna arm is 11.5 . The horn angle itself is approximately 150 . III. FDTD MODEL DESCRIPTION A full scale model of the UWB HFB antenna prototype respace. A spatial quires a minimum of cell size of 6.3 mm was chosen to accurately model the geometrical details of the antenna and cable structure [30]. This yields approximately 96 million unknowns. The FDTD grid is shown in Fig. 2. All dimensions in the model were chosen to be as close to the actual prototype as possible. The four antenna arms were modeled as PEC plates, and the curved edges and surfaces were approximated by staircases. Each tapered R-card attached to the

end of the PEC arm is 63 cm in length and is implemented via a conductive sheet. The ground was assumed to be a lossless half space with relative permittivity of 5. A. Heterogeneous FDTD Domain Partition The antenna geometry under study is very complicated and resides in a complex environment. A traditional FDTD approach to represent the geometry would require either the storage of the material properties for each cell or else a data organization similar to what is used in the finite element method, which would also require a significant amount of memory overhead. To minimize the memory usage, we have adopted a partitioning scheme [20]. The FDTD domain is divided into blocks. The size and number of the blocks are judiciously chosen, so that the material properties within most of the blocks are homogeneous. Within the code, the FDTD algorithm is computed in different ways, and based on the properties of the block, the appropriate FDTD algorithm will be chosen. If the block is a perfect conductor, the FDTD code will recognize this, and not perform any computations for that block. Thus, there is no need to store either the fields or the material properties for that particular block. If the block is an homogeneous dielectric, the material properties are not treated as a function of the grid points within the block but instead represented just as a constant parameter. Thus only the field values need to be stored for each cell within that block. If the block is an inhomogeneous dielectric, then the FDTD algorithm used will assume a constant permeability and no conductivity. Thus, only the fields and permittivity must be stored for each cell. In our case, we divide the geometry into 196 blocks with only five of the blocks being heterogeneous as demonstrated in Fig. 2.

1986

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004

Fig. 3. Coaxial cable modeling in rectangular FDTD grid and the TEM current excitation scheme. (a) Top view, (b) side view, and (c) J (t).

B. Feed Cable Modeling In the discretized FDTD model, each coaxial cable has a square cross sectional area with a single-cell PEC wire surrounded by four PEC walls. As shown in Fig. 3(a), a relative dielectric constant of 1.5 is specified between the center wire and the PEC walls. Each cable is terminated with perfectly matched layer at one end and connected to the tip of an antenna arm at the other end. A balanced excitation is introduced to the opposite pair of cables to excite one antenna polarization as shown in Fig. 3(a) and (b). The time history of the response is also recorded at the excitation position to obtain reflection and transis obtained with the exmission data. The reflection data citation and observation points co-located in the same cable and is obtained with the observation point cross-coupling data located at the second cable. A differential Gaussian pulse is chosen as the time-domain excitation current (2) , and . where These parameters for the Gaussian pulse are determined so as to provide significant spectral energy in the frequency range of 10 to 800 MHz. Fig. 3(c) illustrates the pulse. C. Resistive Card Modeling In the FDTD model, the R-card is modeled as a single-cell corresponding to the delayer with a tapered conductivity . Conductivity along the direction sired sheet resistance where is the thickness of is calculated by

Fig. 4. Resistive card overlay configurations for the PEC launcher section = 300 = , R = 3 = ). (R

single layer. This assumption is valid when is much greater than the penetration depth but much smaller than the free space wavelength [31]. In addition to the exponential taper described in Section II-A, a linear taper with the following taper function was also investigated using the FDTD model as a comparison (3) where is the initial sheet resistance of the R-card is the end sheet at the PEC/R-card junction, resistance. The taper length is equal to that of the previous

LEE et al.: MODELING AND INVESTIGATION OF A GEOMETRICALLY COMPLEX UWB GPR ANTENNA USING FDTD

Fig. 5.

S

, S , and surge impedance of the HFB antenna. (a) Reflection coefficient S

exponential taper, i.e., 0.63 m. As it will be shown shortly, a linear taper provides a better performance, i.e., lower reflection at low frequency end, due to relatively short taper length with respect to wavelength. A more detailed analysis on this aspect can be found in [29]. Fig. 4 plots the linear resistive taper as well as its position relationship with respect to the antenna arm. The lateral edges of the R-card were kept aligned to the edges of the PEC arms to avoid undesired diffractions (see Fig. 1). IV. CHARACTERISTICS OF DUAL-POLARIZED HFB ANTENNA DESIGN A.

&

and Input Impedance

The simulated and measured reflection and transmission coand , of the HFB design are compared in efficients, Fig. 5(a). Note that the antenna is located on the surface of a half space with a dielectric of 5, corresponding to the dry sand in reis similar to since the both antenna arms have ality. The

and S

1987

and (b) antenna surge impedance.

and provide the co-polarized backscatthe same design. provides the cross-polarized kscattering data. tering data. A calibration procedure was carried out in a similar manner as done in real measurement using “short” and “matched” (PML) reference loads at the end of the feed cables

(4) is the response obtained with the coaxial In the above, cables connected to the antenna, is the response obtained is the with the coaxial cables connected to matched load, response obtained when the coaxial cables are shorted at the end is the response obtained at coaxial with a conducting wire, cable 2 with antenna connected when the excitation is applied is the incident wave. to cable 1, and It is observed that the both linear and exponential taper have similar performance at frequency above 0.3 GHz where the

1988

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004

Fig. 6. Comparison of reflected electric field difference with various ground profiles.

taper length becomes comparable or longer than one wavelength (considering dielectric constant of 5). It is also observed that the linear taper produces lower reflection level than the exponential taper at frequencies below 0.1 GHz. Overall, the reflection level is less than 10 dB above 0.05 GHz. This verifies broadband characteristic of the HFB design. The measured data is found to be on average 10 dB higher than that predicted from the simulation. This difference is most likely caused by the asymmetry of the construction of prototype antenna arms and the feed structure. Good agreement between the measurement and simulation is the result of the geometrical fidelity between of the FDTD numerical model and the prototype, including the R-card geometry, conductive plates, dielectric filling and coaxial cable feed modeling. However, the prototype measurement introduces additional environmental variables more difficult to control such as ground loss, slight asymmetry of the antenna arm design due to hand-made fabrication, and discrepancies between the equivalent conductive single layer R-card used in the FDTD model and the thin film R-card conductivity value.bac as The surge impedance can also be calculated from the shown in (5), often applying a time gate to keep only the first peak associated with the feed point near 0 ns position as shown in Fig. 6 (5) is the characteristic impedance of the twin-coaxial where cable. The resultant surge impedance is shown in Fig. 5(b). For , the surge most of the band range, as desired. impedance is found to be within B. Ground Effect In order to see how the ground properties affect the surge impedance of the HFB design, four different ground dielectric constants: 5, 7, 9, and 11 are simulated. Fig. 6 shows the reflected field from 0 to 3 ns. The height of the antenna feed above the ground is equal to 0.1 m. This causes the reflection from the ground surface to be delayed by approximately 1.5 ns since the antenna dielectric filler has a relative permittivity of 5. This

Fig. 7. Reflected (E ) field in time domain for HFB antennas with different resistive card overlay configurations and using same conductivity profile.

Fig. 8. Comparison of co-polarized (E ) reflected field in time domain from HFB antenna with the different resistive cards (R ).

agrees with the significant variations shown in the data near 1.5 ns position. Most importantly, the first reflection peak arising from the feed point remain unaffected by the ground property, as desired. C. R-Card Performance Investigation We investigate two parameters that play an important role in minimizing reflections from the truncated antenna arms. The first parameter is the overlay distance between the PEC and R-card. In the actual HFB prototype, a 5 cm overlay was used to allow the electromagnetic energy to be coupled into the R-card section because the R-card was coated with a protective insulator and could not have a direct electrical contact with the antenna arm. The second parameter is the far-end resistance value that affects the tapering rate of the R-card. If the taper is done too rapidly, undesired diffractions would be produced by the R-card. On the other hand, if the taper is too slow, the far-end reflection may still be too strong. To investigate the effect of PEC and R-card overlay distances, the following three cases were simulated as shown in Fig. 4. In case 1 through 3, the overlay distances are 11.3, 5, and 0 cm, respectively. The simulated reflection responses are plotted in

LEE et al.: MODELING AND INVESTIGATION OF A GEOMETRICALLY COMPLEX UWB GPR ANTENNA USING FDTD

1989

Fig. 9. Snap shots from FDTD simulation for E field strength in dB scale where R = 300 = . (a) t = 7:4539 ns with R-card attached-(dB) scale; (b) t = 7:4539 ns without R-card-(dB) scale; (c) t = 13:0954 ns with R-card attached-(dB) scale; (d) t = 13:0954 ns without R-card-(dB) scale.

Fig. 7. As expected, the overlay distance of the tapered R-card affects the reflection at the PEC end. Note that the R-card in the overlay section is shorted out by the PEC, this section would have an effective resistance of zero regardless of the R-card value. The larger geometric discontinuity in Case 3 provides the stronger junction reflection observed in the figure. Case 1 and 2 provide a smoother transition and result in a 35 dB reflected field at the end of the R-card. Based on the simulations, we concluded that a linear-tapered R-card with either a 11.3 or 5 cm overlay at the PEC/R-card does the best job of suppressing the reflections. , values of 100, 200, 300, and To optimize the choice of were implemented and simulated separately. From the 400 reflected field observed at the feed point, the amount of endreflection suppression was compared as shown in Fig. 8, where late time (after 20 ns) antenna reflections can be observed. These provides the maximal results indicate that suppression of the arm end reflections. D. Antenna Ringing Fig. 9 compares snapshots of the instantaneous field dis) plane with and without the tribution in the vertical (or

R-card attached to the HFB antenna arms. Without the R-card, significant diffraction and reflection at the end of the PEC arms are observed. The reflected fields later propagate back to the observation point inside the cables as shown in Fig. 9(b). On the other hand, the R-card extension significantly reduces the diffraction and reflection at the ends as depicted in Fig. 9(a) and lowers the antenna ringing by approximately 20 dB. Note that the signals that propagate back to the feed point are partially reflected due to the imperfect matching. This reflected fields generate the secondary reflection. This process repeats and becomes the well known “antenna ringing” effect, a major clutter source in GPR measurements. E. Radiated Field Distribution & Polarization The near-field radiation characteristics are investigated next. Fig. 10 depicts the simulated horizontal co-polarized and crosspolarized field distributions at a plane 40 cm below the antenna aperture, (corresponding to the ground surface plane), at the center frequency of 400 MHz. The cases with and without the R-card are also plotted for comparison. The fields are nearly linearly polarized in the principal planes. The results with the R-card clearly show a more uniform distribution, because the

1990

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004

Fig. 10. Comparison of co- and cross-polarized aperture field distributions at f = 400 MHz, depth z = 40 cm or 0.53  ,R = 300 = in R-card. (a) Co-polarized field with R-card, (b) cross-polarized field with R-card, (c) co-polarized field without R-card, and (d) cross-polarized field without R-card.

diffracted fields from the antenna arm ends modify the radiated fields that otherwise would have been close to simple spherical wavefronts. The more uniform field distributions simplify the subsequent signal processing and inverse problem and improve the overall detection/classification capabilities of a GPR system. As the observation point moves away from the principal planes, the level of depolarization increases and reaches a maximum of approximately 12 dB between the two antenna polarizations. This is, of course, due to the spherical nature of the wavefront. We note that the cross-polarized field levels with the R-card present are a little bit higher. This again may be caused by distributed diffractions along the resistive cards.

HFB antenna was calculated from the reflection coefficients and was found to be approximately 100 ohms over the entire frequency band of interest. This result confirms the broadband characteristic of the HFB design. The FDTD model also provided useful visualization of dynamic field distributions that can help identify undesired radiations and reflections sources. The near-field distributions of the co-polarized and cross-polarized fields were examined. This information is particularly useful in GPR applications where the depth of the target is unknown. Overall, the simulated results confirm that the optimized HFB antenna design is a very attractive choice for broadband, fully polarimetric GPR applications.

V. CONCLUSION

ACKNOWLEDGMENT

In this work, a detailed FDTD model was used to incorporate realistic features of UWB HFB antennas such as feeding cables, dielectric loading and tapered resistive terminations. The FDTD model is flexible enough to model different geometries, structures, and materials for both the antenna and the ground medium. Fully-polarimetric simulations were performed to obtain the radiation characteristics of HFB antennas over a broad frequency range. A parametric study on the effect of the resistive taper of the R-card termination was also performed. It was found that a linear taper performs better than the commonly used exponential taper for short taper length. It was also found that a proper overlapping between the PEC and R-card improves the transition and reduces the diffraction at the end of PEC. The R-card termination also significantly reduces the undesired antenna ringing. The surge impedance of the

The authors acknowledge the reviewers for their helpful comments. REFERENCES [1] L. Peter Jr, J. D. Young, and J. Daniels, “Ground penetration radar as a subsurface environmental sensing tool,” Proc. IEEE, vol. 82, pp. 1802–1822, Dec. 1994. [2] S. Agosti, G. G. Gentili, and S. Spagnolini, “Electromagnetic inversion of monostatic GPR: application to pavement profile,” in Proc. Int. Conf. Electromag. Adv. Applicat. (ICEAA’97), 1997, pp. 491–494. [3] L. C. Chan, D. L. Moffatt, and L. Peters Jr., “A characterization of subsurface radar targets,” Proc. IEEE, vol. 67, pp. 91–110, July 1979. [4] C.-C. Chen, S. Nag, W. Burnside, J. Halman, K. Shubert, and L. Peters Jr, “A stand-off, focused-beam land mine radar,” IEEE Trans. Geosci. Remote Sensing, vol. 38, pp. 507–514, Jan. 1998. [5] C.-C. Chen, K. R. Rao, and R. Lee, “A new ultra-wide bandwidth dielectric rod antenna for ground penetrating radar applications,” IEEE Trans. Antennas Propagat., vol. 51, pp. 371–377, Mar. 2003.

LEE et al.: MODELING AND INVESTIGATION OF A GEOMETRICALLY COMPLEX UWB GPR ANTENNA USING FDTD

[6] C.-C. Chen and L. Peters Jr, “Buried unexploded ordnance identification via complex natural resonances,” IEEE Trans. Antennas Propagat., vol. 45, pp. 1645–1654, Nov. 1997. [7] C. C. Chen, B. Higgins, K. O’Neil, and R. Detsch, “Ultrawide-bandwidth fully-polarimetric ground penetrating radar classification of subsurface unexploded ordnance,” IEEE Trans. Geosci. Remote Sensing, vol. 39, pp. 1259–1270, June 2001. [8] T. P. Montoya and G. S. Smith, “Land mine detection using a groundpenetrating radar based on resistively loaded vee dipoles,” IEEE Trans. Antennas Propagat., vol. 47, pp. 1795–1806, Dec. 1999. [9] J. Hugenshmidt, “A one-to-one comparison between radar results and reality on a concrete bridge,” in Proc. 9th Int. GPR Conf., vol. SPIE4758, May 2001, pp. 427–432. [10] M. Higgins and C.-C. Chen, “Nondestructive evaluation of soil hardness using elevated focused-beam radar,” in Proc. 9th Int. GPR Conf., vol. SPIE-4758, May 2001, pp. 54–57. [11] M. Moghaddam, W. C. Chew, B. Anderson, E. Yannakis, and Q. H. Liu, “Computation of transient electromagnetic waves in inhomogeneous media,” Radio Sci., vol. 26, pp. 265–273, Jan. 1991. [12] S. J. Radzevicius, J. J. Pariels, and C.-C. Chen, “GPR H-Plane Antenna Patterns for a horizontal dipole on a half space interface,” in Proc. 8th Int. GPR Conf., vol. SPIE-4084, Gold Coast, Australia, June 2000. [13] C. C. Chen and J. D. Young, “Unfurlable folded-dipole UWB antenna for mars explorer subsurface sensing,” in Proc. 8th Int. GPR Conf., vol. SPIE-4084, Gold Coast, Australia, Jun. 2000. [14] M. Moghaddam, E. J. Yannakakis, W. C. Chew, and C. Randoll, “Modeling of the subsurface interface radar,” J. Electromag. Waves Applicat., vol. 5, pp. 17–39, 1991. [15] J. M. Bourgeois and G. S. Smith, “A fully three-dimensional simulation of a ground-penetrating radar: FDTD theory compared with experiment,” IEEE Trans. Geosci. Remote Sensing, vol. 34, pp. 36–44, 1996. [16] K. R. Rao, K. H. Lee, C. C. Chen, and R. Lee, “Application of fullpolarmetric ground penetration radar for buried UXO Classification,” The Ohio State Univ., ElectroSci. Lab., Tech. Rep. 738 520-1, Feb. 2001. [17] K.-H. Lee, N. V. Venkatarayalu, and C.-C. Chen, “Numerical modeling development for characterizing complex gpr problems,” in Proc. Int. GPR Conf., vol. SPIE-4758, May 2002, pp. 625–652. [18] F. L. Teixeira, W. C. Chew, M. Straka, M. L. Orstaglio, and T. Wang, “Finite-difference time-domain simulation of ground penetrating radar on dispersive inhomogeneous, and conductive soils,” IEEE Trans. Geosci. Remote Sensing, vol. 36, pp. 1928–1937, Nov. 1998. [19] L. Gurel and U. Oguz, “Three-dimensional FDTD modeling of a ground penetrating radar,” IEEE Trans. Geosci. Remote Sensing, vol. 38, pp. 1513–1521, Jul. 2000. [20] J. Nehrbass, “Physics based partitioning,” in Proc. 26th General Assembly for URSI, Ontario, Canada, Aug. 2000. [21] F. L. Teixeira and W. C. Chew, “Finite-difference simulation of transient electromagnetics fields for cylindrical geometries in complex media,” IEEE Trans. Geosci. Remote Sensing, vol. 38, pp. 1530–1543, July 2000. [22] K.-H. Lee, N. Venkalayalu, C.-C. Chen, F. L. Teixeira, and R. Lee, “Application of full-polarmetric ground penetration radar for buried UXO Classification (II),” The Ohio State Univ., ElectroSci. Lab., Tech. Rep. 778 520, May 2002. [23] C. Handel, I. J. Gupta, and W. D. Burnside, “Low frequency modification of a dual chamber compact range,” The Ohio State Univ., ElectroSci. Lab., Tech. Rep. 732 264, Sep. 1997. [24] L. Chaung, T. Chang, and W. D. Burnside, “An ultrawide-bandwidth tapered resistive TEM horn antenna,” IEEE Trans. Antennas Propagat., vol. 48, pp. 1848–1857, Dec. 2000. [25] M. S. A. Mahmoud, T.-H. Lee, and W. D. Burnside, “Enhanced compactrange reflector concept using an R-card fence: two-dimensional case,” IEEE Trans. Antennas Propagat., vol. 49, pp. 419–428, Mar. 2001. [26] F. C. Yang and K. S. H. Lee, “Impedance of a Two-Conical-Plate Transmission Line,” Tech. Rep., Sensor and Simulation Company, Nov. 1976. [27] H. M. Shen, R. W. P. King, and T. T. Wu, “V-conical antenna,” IEEE Trans. Antennas Propagat., vol. 36, pp. 1519–1525, Nov. 1988. [28] C. C. Chen, “A new ground penetrating radar antenna design—the horn-fed bowtie (HFB),” in Proc. Antenna Measurement Techniques Association (AMTA) Symp., Nov. 1997, pp. 67–74. [29] N. Venkatarayalu, C.-C. Chen, F. L. Teixeira, and R. Lee, “Modeling of ultrawide-band dielectric horn antennas using FDTD,” IEEE Trans. Antennas Propagat., vol. 52, pp. 1318–1323, May 2004. [30] A. Taflove, Computational Electrodynamics. Norwood, MA: Artech House, 1995. [31] T. B. A. Senior, “Approximate boundary conditions,” IEEE Trans. Antennas Propagat., vol. 29, pp. 826–829, Sept. 1981.

1991

Kwan-Ho Lee (M’02) received the B.S. degree from the Department of Radio Science and Engineering, Kwangwoon University, Seoul, Korea, in 1997 and the M.S. degree in electrical engineering from The Ohio State University, Columbus, in 1999, where he is currently working toward the Ph.D. degree. Since 1997, he has been a Graduate Research Associate at the ElectroScience Laboratory, Department of Electrical Engineering, The Ohio State University. His research interests include computational electromagnetics, ultrawide-bandwidth antenna development, subsurface target detections and classifications, RF circuits and object oriented programming.

Chi-Chih Chen (S’92–M’97) was born in Taiwan, R.O.C., in 1966. He received the B.S.E.E. degree from the National Taiwan University, Taiwan, R.O.C., in 1988 and the M.S.E.E. and Ph.D. degrees from The Ohio State University, Columbus, in 1993 and 1997, respectively. He joined the ElectroScience Laboratory, The Ohio State University, as a Postdoctoral Researcher in 1997 and became a Senior Research Associate in 1999. His main research interests include the ground penetrating radar, UWB antenna designs, radar target detection and classification methods, automobile radar systems. In recent years, his research activities have been focused on the detection and classification of buried landmines, unexploded ordnance and underground pipes. Dr. Chen is a Member of Sigma Xi and Phi Kappa Phi.

Fernando L. Teixeira (S’89–M’93) received the B.S. and M.S. degree in electrical engineering from the Pontifical Catholic University of Rio de Janeiro (PUC-Rio), Brazil, in 1991 and 1995, respectively, and the Ph.D. degree in electrical engineering from the University of Illinois at Urbana-Champaign, in 1999. From 1999 to 2000, he was a Postdoctoral Research Associate with the Research Laboratory of Electronics, Massachusetts Institute of Technology (MIT), Cambridge. Since 2000, he has been an Assistant Professor at the ElectroScience Laboratory (ESL) and the Department of Electrical Engineering, The Ohio State University, Columbus. His current research interests include analytical and numerical techniques for wave propagation and scattering problems in communication, sensing, and devices applications. He has edited one book Geometric Methods for Computational Electromagnetics (PIER 32, EMW: Cambridge, MA, 2001), and has published over 30 journal articles and 50 conference papers in those areas. Dr. Teixeira is a Member of Phi Kappa Phi. He was awarded the Raj Mittra Outstanding Research Award from the University of Illinois, and a 1998 MTT-S Graduate Fellowship Award. He received paper awards at 1999 USNC/URSI National Symposium (Orlando, FL), and received a Young Scientist Award at the 2002 URSI General Assembly. He was the Technical Program Coordinator of the Progress in Electromagnetics Research Symposium (PIERS), Cambridge, MA, in 2000.

Robert Lee (M’92) received the B.S.E.E. degree in 1983 from Lehigh University, Bethlehem, PA, and the M.S.E.E. and Ph.D. degree in 1988 and 1990, respectively, from the University of Arizona, Tucson. From 1983 to 1984, he worked for Microwave Semiconductor Corporation, Somerset, NJ, as a Microwave Engineer. From 1984 to 1986, he was a Member of the Technical Staff, Hughes Aircraft Company, Tucson, AZ. From 1986 to 1990, he was a Research Assistant at the University of Arizona. During summer 1987 through 1989, he worked at Sandia National Laboratories, Albuquerque, NM. Since 1990, he has been at The Ohio State University, where he is currently a Professor. His major research interests are in the development and application of numerical methods for electromagnetics.