Multiband Electromagnetic-Bandgap Structures for Applications in ...

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 56, NO. 10, OCTOBER 2008

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Multiband Electromagnetic-Bandgap Structures for Applications in Small Form-Factor Multichip Module Packages Telesphor Kamgaing, Senior Member, IEEE, and Omar M. Ramahi, Senior Member, IEEE

Abstract—The design and implementation of package-level electromagnetic-bandgap (EBG) structures is presented. By using spiral-based inductance-enhanced electromagnetic-bandgap structures (IE-EBGs), the relative periodicity for achieving bandgap at extremely low frequencies is substantially reduced in comparison to traditional EBGs. Using both full-wave electromagnetic simulation and experimental characterization, it is demonstrated that this type of structure can exhibit multiple bandgaps, which are individually tunable through variation of the inductance per unit area and/or the unit cell periodicity. Sample structures with dimensions compatible with today’s microprocessor packaging technology are designed and fabricated in a multilayer organic flip-chip ball-grid array package substrate. These package-embedded IE-EBGs have unit cell dimensions less than 750 m and exhibit the first forbidden bandgaps for electromagnetic wave propagation below 10 GHz, which is a frequency band of interest for commercial wireless communication systems. Index Terms—Artificial magnetic conductor, electromagnetic-bandgap (EBG) structures, multichip packaging, multilayer organic package substrate, signal isolation.

I. INTRODUCTION HE tremendous potential of electromagnetic-bandgap (EBG) structures as circuit elements in filters, waveguides, and antennas has made them a great attraction in the microwave engineering community. While early research in this area is mainly focused on optical frequencies, the introduction of innovative structures such as the high-impedance electromagnetic surface [1], where the bandgap behavior of the structures relies primarily on their capacitive and inductive loading and less on the periodicity, has paved the way to designing EBG structures at microwave frequencies. Most recently, the use of EBG and other artificial electromagnetic conductors for mitigating switching noise in high-speed circuits has been introduced [2]–[4]. Additional research [5] has also shown that an EBG structure can be used for signal isolation in RF/analog mixed-signal systems. The solution demonstrated in [4] and similar research, however, mainly addresses an EBG

T

Manuscript received January 1, 2008, revised May 27, 2008. First published September 23, 2008; current version published October 8, 2008. T. Kamgaing is with Components Research, Intel Corporation, Chandler, AZ 85226 USA (e-mail: [email protected]). O. M. Ramahi is with the Electrical and Computer Engineering Department, University of Waterloo, Waterloo, ON, Canada N2L 3G1 (e-mail: oramahi@ece. uwaterloo.ca). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2008.2003525

that can only be implemented for printed circuit board (PCB) modules [6]–[25], where the dimensions are relatively large. In [10], the concept of spiral-based EBG structures was mainly introduced for switching noise mitigation and it was shown that enhancement of the EBG inductance can be achieved by using an additional layer of the multilayer PCB exclusively for routing the inductor. It has also been demonstrated numerically [11] that the spiral can be patterned at the same level as the patch to enhance the inductance per unit area, and hence, shift the location of the bandgap towards lower frequencies. In [17], single- and two-turn spiral EBGs were introduced as part of a systematic EBG design approach. This work, however, only focused on analyzing the first or primary bandgap and did not address the existence of higher bandgaps. From the cost standpoint, the inductance enhancement approach has the significant advantage that it does not require new technology development or inclusion of exotic materials such as high-permittivity or high-permeability materials as required by other approaches reported in the literature [23], [24]. With the continuous integration of wireless communication systems and the associated demand for form-factor reduction, a substantial downscaling of the EBG structures to package level dimensions is necessary before this technology can be used in small form-factor antennas and for signal isolation in highly integrated RF modules. According to the industry trends for multiradio integration, such a technology will have the most use in small form-factor devices such as the ultramobile personal computer (UMPC), where multimode radios including voice, data, and location have to coexist. In this paper, EBG structures are analyzed in a parallel-plate waveguide environment, which is representative for the power delivery network of a wireless communication multichip module as the one represented in Fig. 1. Besides mitigating noise propagation between the main chips of the system, the main purpose of this study is twofold, i.e., introducing a multiband EBG that can be used in place of a single wideband structure while occupying relatively smaller real estate and demonstrating the feasibility of this novel EBG at the package level without inclusion of any special materials. This paper is organized as follows. In Section II, we briefly review the concept of EBG structures and some techniques that have been used to enable operation at very low frequencies. In Section III, we introduce and discuss the concept of multiband inductance-enhanced EBG (IE-EBG). Section IV addresses the experimental verification of the IE-EBG concept through package-level implementation.

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Fig. 3. Capacitively enhanced EBG [1].

where the EBG does not support the propagation of any electromagnetic waves. The EBG is primarily characterized through the center frequency of the bandgap given by (1)

Fig. 1. Illustrative view of noise propagation in multichip-module package containing analog, digital, and mixed signal circuits from [30].

Considering a 2-D wave propagation in the structure, each main direction can be modeled as a cascade of an identical parallel resonator, whose bandwidth is given by

(2)

Fig. 2. Traditional mushroom EBG: top view, equivalent electrical model, and cross-sectional view.

II. OVERVIEW AND THEORY OF EBG STRUCTURES EBG structures are typically realized by periodically loading a grounded dielectric material with periodic metallic structures. Fig. 2 is an illustration of the high-impedance structure (HIS), also known as the mushroom EBG, originally introduced in [1]. It consists of metallic patches connected to the ground of the nonconductive dielectric substrate using vertical vias. Any two adjacent cells of this structure can be modeled as a parallel resonator, whereby , the capacitance per unit area, represents the fringing capacitance between adjacent patches and , the inductance per unit area, represents the inductive path (patch–via–ground–via–patch) from one patch to the next. A surface current induced on the patch will flow through the inductive path at low frequencies and through the capacitive path at high frequencies. This means that the mushroom EBG allows the propagation of TM waves at low frequencies and the propresagation of TE waves at high frequencies. Similar to onators, there exists a transition region, also known as the EBG,

whereby is the characteristic impedance of the highimpedance surface and is the impedance of free space. Other figures-of-merit include the fractional bandwidth, as well the upper and lower edges of the bandwidth, which can be determined using the method described in [25]. From (1), it can be seen that low-frequency EBGs require an increase in capacitance per unit area or in inductance per unit area. Inductance increase can be obtained by using longer vias or high-permeability material. Similarly, a capacitance increase can be achieved by using high-permittivity material or by substantially reducing the spacing between the adjacent patches. To circumvent the minimum trace spacing defined by given technology nodes, multilayer patches, as illustrated in Fig. 3, can be used, whereby an overlap between the patches enables capacitance density in the order of that of regular parallel-plate capacitors. Other EBG structures, such as the uniplanar compact photonic bandgap (UC-PBG) [26] and the alternating-impedance EBG (AI-EBG) [27] have been demonstrated and do not use vertical vias. This, however, results in very large periodicities for low-frequency applications [26], [27]. Furthermore, planar EBG structures can potentially lead to degradation in the signal integrity, as was demonstrated recently in [30]. III. TOPOLOGY AND CONCEPT OF SPIRAL-BASED IE-EBG A. Topology Fig. 4 shows the top and cross-sectional views of the unit cell of the spiral-based IE-EBG. It is obtained by replacing the patch of the mushroom EBG cell [1] with a spiral inductor that has a varying metallization linewidth. The spiral inductor has one end connected to a via such that the effective inductance of the via is increased. While full-wave analysis is necessary for accurate

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Fig. 4. Isometric and cross-sectional views of the IE-EBG unit cell using a spiral inductor as patch. Reduced Brillouin zone is illustrated.

determination of the inductor performance, an estimate of the inductance enhancement due to the spiral can be obtained from the thin-film inductance formulas reported in [28], which does not take into account any potential interaction of the spiral with substrate or potential substrate embedded grounds. In this approach, the total self-inductance of the spiral is first determined as the sum of the self-inductances of the individual segments, as illustrated in (3), whereby , , and represent width, thickness, and length of trace segment , respectively,

Fig. 5. Modeled dispersion diagram of an IE-EBG in parallel-plate waveguide environment. Cell dimension is 10 mm 10 mm. Normalized wavenumber is indicated for each direction of propagation.

2

(3) Second, the mutual inductance between any two parallel segand represent ments is defined by (4). In this equation, the length overlap and separation between trace segments I and I, respectively,

(4) The total mutual inductance is the obtained by summing up the individual mutual inductance pieces

2

Fig. 6. Insertion loss of a 10 cm 10 cm parallel-plate waveguide with embedded IE-EBG. Period is 10 mm and distance between the ports is 0.3 mm.

(5) The overall inductance of the spiral is then obtained as the sum of self-inductance and mutual inductance (6) For a defined technology or substrate stackup, the surface impedance can be controlled by varying the geometrical parameters of the inductor, which are the inner diameter, number of turns, linewidth, and line spacing. With reference to (2) and also resonator, increasing the inductance per in analogy to the unit area instead of capacitance per unit area leads to a larger . fractional bandwidth since the later is proportional to B. Simulation and Discussion of Results To validate the concept of the spiral-based IE-EBG, we first consider implementation with PCB dimensions, where the period or unit cell dimension is 10 mm. Limited rows of EBG cells are integrated inside a parallel-plate waveguide to evaluate its

impact on the waveguide resonant modes. Using Ansoft Corporations’ commercially available electromagnetic High Frequency Structure Simulator (HFSS) [29], we evaluate both the dispersion diagram, where we solved only the first three eigenbetween two desigmodes, and the transmission parameter nated points of the finite array of 10 10 cells. Fig. 5 shows the full – – – dispersion diagram of the structure, computed on the boundary of the reduced Brillouin zone. It represents the propagating frequencies as a function of both the wavenumber and direction of wave propagation. This analysis provides the behavior of waves propagating tangentially to the surface with the – and – segments representing waves propagating orthogonally to the surface and the – segment representing waves propagating in the diagonal direction of the surface. Two bandgaps are exhibited, one from 1.2 to 2 GHz, and the other from 3.42 to 3.75 GHz. Fig. 6 shows the insertion loss versus frequency for the same structure. The 20-dB bandgaps are exactly the same as those predicted by the dispersion diagram. In addition, there

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Fig. 7. Dispersion diagram illustrating the effect of varying the gap size on the location and width of the bandgap. Solid line is for g = 0:6 mm and dotted line is for g = 1:2 mm.

is a global minimum in the insertion loss curve at 2.6 GHz. A direct comparison with the dispersion diagram indicates that this frequency point does not represent a narrow stopband, but it is rather the frequency point where the second mode starts to transition from a predominantly TE wave to a predominantly TM wave. At this frequency point, the second mode is a pure TEM mode, as it intercepts the light line in the medium. The first mode is a TM mode, which starts as a forward propagating TEM mode at very low frequency and low wavenumber, and changes into a forward propagating TM surface wave. At very high wave numbers, a decrease in the group velocity of this mode indicates that it is propagating backward. The second mode is a hybrid mode, which starts as a TE wave at a very low wavenumber and behaves like a TM wave at a very high wavenumber. The third mode is a pure TE wave. C. Bandgap Tuning by Varying the Gapwidth The first of two methods considered in controlling the bandgap(s) of the EBG in a parallel-plate environment consists of varying the separation or gap between adjacent patches. In doing this, all other parameters are fixed. Fig. 7 shows the dispersion diagram ( – segment only) of the unit cell, where the patch separation has been varied from 0.6 to 1.2 mm. This is done by increasing the width of the outer loop of the spiral inductor. The upper band or second bandgap moves towards lower frequencies with continuous decrease in the bandwidth. When the gap reaches a certain value, as indicated by the solid line, the second bandgap disappears. On the other hand, it can be seen that the lower or first bandgap remains unaffected. When the gap between patches is very large, a second bandgap is generated. As the size of the gap decreases, the third mode moves faster towards lower frequencies and at about the same rate across the entire spectrum (wavenumbers). The second mode also moves towards lower frequency. This shift is more pronounced at higher frequencies. Since the second mode shifts much more slowly, the second bandgap will eventually merge into the first band due to the simultaneous presence of both capacitive and magnetic coupling between the adjacent cells.

Fig. 8. Effect of number of turns on the fundamental stopband of the IE-EBG in parallel-plate waveguide environment.

D. Bandgap Tuning by Varying the Number of Turns of the Spiral Inductor The second method considered for tuning the bandgaps consists of varying the number of turns of the spiral inductor used as the IE-EBG patch. In this case, we consider a unit cell of a period of 5 mm. The substrate is lossless with a relative dielectric constant of 4.4. The EBG height and the separation between the two plates remain 1.54 mm. The gap between the patches is 0.6 mm and the via diameter is 0.4 mm. The main electrical performance attributes are all derived from the – section of the dispersion diagram. The number of turns , ( ) of the spiral inductor was varied form 1 to 2.5. The properties of the obtained bandgaps are summarized in Fig. 8 along with those of the . The stanstandard mushroom structure represented by dard EBG exhibits only one bandgap, which is located at much higher frequencies than the fundamental bandgap of structures with an IE-EBG. As the number of turns of the spiral inductor increases, the surface inductance increases very fast, leading to the lowering of the center frequency. At the same time there is a decrease in the absolute bandwidth of the forbidden bandgap. A potential reason for the absolute bandwidth is the redistribution of the electrical charges along the edges of the spiral inductors, which is the same electrical phenomenon that explains self-resonance of the inductors. It is also important to note that as the inductance goes beyond a certain critical value, the second bandgap will also disappear. This specific case is illustrated in Fig. 9. IV. PACKAGE LEVEL IMPLEMENTATION OF EBGs AND ELECTRICAL CHARACTERIZATION RESULTS A. Design and Fabrication In order to evaluate the electrical performance of the IE-EBGs for low-frequency applications, several structures with periodicities ranging from 400 to 750 m have been designed and fabricated on the multilayer organic flip-chip ball-grid array (FCBGA) substrate described in [32]. The performance of these structures is evaluated by inserting them

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Fig. 9. Dispersion diagram illustrating the disappearance of the second bandgap due to appropriate increase in inductance.

Fig. 10. Representative cross section of multilayer organic package substrate: (a) with and (b) without embedded EBG unit cell and (c) illustrative side view with testing pads.

in a parallel-plate waveguide environment, as illustrated by the cross-sectional representation of Fig. 10. Fig. 11 shows the top view of the fabricated EBGs, whereby m in both all cells have a unit cells of periodicity - and - direction. EBG_1 and EBG_2 are traditional mushroom EBGs and have unit cell size of 750 m and patch to patch gap of 100 and 300 m, respectively. EBG_3 and EBG_4 use spiral patches with the same periodicity and gap between the patches as their mushroom counterpart EBG_1. For simplicity, the inductor trace has a uniform width for all turns. The only key difference between EBG_3 and EBG_4 resides in the inductance density, which, in this case, is controlled by varying the number of turns and trace width of the spiral patches. EBG_3 uses a two-turn spiral patch with a starting inner diameter of 200 m, a uniform trace width of 100 m, and a spacing of 26 m between adjacent turns. EBG_4 uses a three-turn spiral patch with 200- m starting inner diameter and trace-to-trace spacing of 26 m. The associated trace width is 50 m. B. Measurement and Discussion of Results The electrical performances of the fabricated structures were measured in the form of two-port scattering ( ) parameters

Fig. 11. Snapshots of the top view of fabricated EBG structures.

using a 50-GHz performance network analyzer (PNA) and on-wafer ground–signal–ground (GSG) probes, as illustrated in Fig. 10(c). The two test ports are separated by 15 EBG cells. In addition, the separation between the EBG columns, where the test port is located, is m, as illustrated in Fig. 10(c). A standard short-open-load-through (SOLT) calibration was used. Fig. 12 shows the transmission coefficient of the reference parallel-plate waveguide in the absence of the IE-EBG. It can be seen that several resonant modes are excited. The -parameters of the traditional EBG_1 is plotted in the same plot. The traditional EBG does not exhibit any stopband

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Fig. 12. Measured electrical performance of parallel-plate waveguide with and without traditional EBG (period = 750 m, spacing = 100 m).

Fig. 14. Modeled and measured electrical performance of IE-EBG using twoturn spiral as patch (period = 750 m, spacing = 100 m).

Fig. 15. Measured electrical performance of IE-EBG using three-turn spiral as patch (period = 750 m, spacing = 100 m). Fig. 13. Modeled electrical performance of traditional EBG in parallel-plate waveguide environment.

V. CONCLUSION behavior below 50 GHz, which is the maximum frequency that can be measured with the PNA. In Fig. 13, we show electrical modeling results of the traditional EBG. Two different via lengths are investigated. In the case of short vias, the separation between the underlying ground plane and the EBG patches is 30 m and the separation between the EBG patches and the signal plane is 75 m. In the case of the long via, the via length is increased to 75 m. It can be noted that the EBG with the short via exhibits a bandgap above 60 GHz, whereas the structure with long vias exhibits a bandgap starting above 40 GHz. Fig. 14 shows the modeled and measured electrical performance of the two-turn spiral IE-EBG. This structure exhibits a total of four bandgaps below 100 GHz, including two below 50 GHz. The lower first fundamental bandgap is around 13 GHz. Fig. 15 shows the measured electrical performance of the three-turn spiral IE-EBG. This EBG exhibits six bandgaps below 100 GHz including three below 50 GHz. The first two bandgaps are located at 9 and 27 GHz, respectively.

Spiral-based EBG structures have been investigated using electromagnetic modeling and experimental characterization. It was shown that this type of EBG exhibits multiple forbidden bandgaps, whereby the first bandgap is determined by the dimension of the unit cell. The second bandgap was shown to be primarily dependent on the capacitive and inductive loading of the unit cell and can be tuned independently of the primary bandgap. Package-level structures, exhibiting bandgaps in the 9- and 27-GHz frequency ranges, have been designed with less than 750- m periodicity and implemented in a standard microprocessor multilayer organic packaging substrate. The realized feature indicates that this technology is an enabler for applying EBGs into portable wireless communication systems including wireless local area networks (WLANs) and ultra-wideband (UWB). While the smallest frequency demonstrated in this study is 9 GHz, it is expected that a mere optimization of the spiral geometry could lead to operation at lower frequencies without any change in the package material or processing. Such EBG structures will then be suitable for compact antennas and noise isolation for future technology platforms requiring multimode radios.

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ACKNOWLEDGMENT The authors would like to thank all members of the Radio Frequency Packaging Integration Working Group, Intel Corporation, Chandler, AZ, for various technical discussions. The authors would also like to acknowledge L. Wojewoda, Assembly and Test Technology Development, Intel Corporation, for electrical measurement support. REFERENCES [1] D. Sievenpiper, L. Zhang, R. F. J. Broas, N. G. Alexopolous, and E. Yablonovitch, “High-impedance electromagnetic surfaces with a forbidden frequency band,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 11, pp. 2059–2073, Nov. 1999. [2] T. Kamgaing and O. M. Ramahi, “Design and modeling of high-impedance electromagnetic surfaces for switching noise suppression in power planes,” IEEE Trans. Electromagn. Compat., vol. 47, no. 3, pp. 479–489, Aug. 2005. [3] T. Kamgaing and O. Ramahi, “A novel power plane with integrated simultaneous switching noise mitigation capability using high impedance surface,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 1, pp. 21–23, Jan. 2003. [4] R. Abhari and G. V. Eleftheriades, “Metallo-dielectric electromagnetic bandgap structures for suppression and isolation of the parallel-plate noise in high-speed circuits,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 6, pp. 1629–1639, Jun. 2003. [5] J. Choi, V. Govind, R. Mandrekar, S. Janagama, and M. Swaminathan, “Noise reduction and design methodology in mixed-signal systems with alternating impedance electromagnetic bandgap (AI-EBG) structure,” in IEEE MTT-S Int. Microw. Symp. Dig., Long Beach, CA, Jun. 12–17, 2005, pp. 849–852. [6] J. Park, A. C. W. Lu, K. M. Chua, L. L. Wai, J. Lee, and J. Kim, “Double-stacked EBG structure for wideband suppression of simultaneous switching noise in LTCC-based SiP applications,” IEEE Microw. Wireless Compon. Lett., vol. 16, no. 9, pp. 481–483, Sep. 2006. [7] T. L. Wu, C. H. Wang, Y. H. Lin, T. K. Wan, and G. Chang, “A novel power plane with super-wideband elimination of ground bound noise on high speed circuits,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 3, pp. 174–176, Mar. 2005. [8] J. Qin and O. M. Ramahi, “Ultra-wideband mitigation of simultaneous switching noise using novel planar electromagnetic bandgap structures,” IEEE Microw. Wireless Compon. Lett., vol. 16, no. 9, pp. 487–489, Sep. 2006. [9] J. Lee, H. Kim, and J. Kim, “High dielectric constant thin film EBG power/ground network for broad-band suppression of SSN and radiated emissions,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 8, pp. 505–507, Aug. 2005. [10] T. Kamgaing and O. M. Ramahi, “Inductance-enhanced highimpedance electromagnetic surfaces for broadband simultaneous switching noise mitigation in power planes,” in IEEE MTT-S Int. Microw. Symp. Dig., Philadelphia, PA, Jun. 8–13, 2003, pp. 2165–2168. [11] T. Kamgaing and O. M. Ramahi, “Electromagnetic magnetic band-gap structures for multiband mitigation of resonant modes in parallel-plate waveguides,” in Proc. IEEE Int. AP-S Symp., Monterey, CA, Jun. 20–26, 2004, vol. 4, pp. 3577–3580. [12] S. H. Joo, D. Y. Kim, and H. Y. Lee, “An S-bridged inductive electromagnetic bandgap power plane for suppression of ground bounce noise,” IEEE Microw. Wireless Compon. Lett., vol. 17, no. 10, pp. 709–711, Oct. 2007. [13] K. H. Kim and J. E. Schutt-Aine, “Design of EBG power distribution networks with VHF-band cutoff frequency and small unit cell size for mixed-signal systems,” IEEE Microw. Wireless Compon. Lett., vol. 17, no. 7, pp. 489–491, Jul. 2007. [14] S. Shahparnia and O. M. Ramahi, “Design, implementation, and testing of miniaturized electromagnetic bandgap structures for broadband switching noise mitigation in high-speed PCBs,” IEEE Trans. Adv. Packag., vol. 30, no. 2, pp. 171–179, May 2007. [15] A. Tavallaee and R. Abhari, “2-D characterisation of electromagnetic bandgap structures employed in power distribution networks,” IET Microw. Antennas Propag., vol. 1, no. 1, pp. 204–211, Feb. 2007. [16] M. S. Zhang, Y. S. Li, C. Jia, and L. P. Li, “Signal integrity analysis of the traces in electromagnetic-bandgap structure in high-speed PCBs and packages,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 5, pp. 1054–1062, May 2007.

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[17] C. L. Wang, G. H. Shiue, W. D. Guo, and R. B. Wu, “A systematic design to suppress wideband ground bounce noise in high-speed circuits by electromagnetic-bandgap-enhanced split powers,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 12, pp. 4209–4217, Dec. 2006. [18] T. L. Wu and S. T. Chen, “A photonic crystal power/ground layer for eliminating simultaneously switching noise in high-speed circuit,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 8, pp. 3398–3406, Aug. 2006. [19] R. Pena-Rivero, H. J. Aguilar, and R. L. Y. Miranda, “Optimum use of high-impedance surface in PCB to mitigate the simultaneous switching noise and radiated emission,” Microw. Opt. Technol. Lett., vol. 48, no. 7, pp. 1446–1449, Jul. 2006. [20] G. Chen and K. L. Melde, “Cavity resonance suppression in power delivery systems using electromagnetic band gap structures,” IEEE Trans. Adv. Packag., vol. 29, no. 1, pp. 21–30, Feb. 2006. [21] T. L. Wu, Y. H. Lin, T. K. Wang, C. C. Wang, and S. T. Chen, “Electromagnetic bandgap power/ground planes for wideband suppression of ground bounce noise, and radiated emission in high-speed circuits,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 9, pp. 2935–2942, Sep. 2005. [22] S. D. Rogers, “Electromagnetic-bandgap layers for broad-band suppression of TEM modes in power planes,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 8, pp. 2495–2505, Aug. 2005. [23] D. J. Kern, D. H. Werner, and M. J. Wilhelm, “Active negative impedance loaded EBG structures for the realization of ultra-wideband artificial magnetic conductors,” in Proc. IEEE Int. AP-S Symp./USNC/CNC/URSI North American Radio Sci. Meeting, Columbus, OH, Jun. 22–27, 2003, vol. 2, pp. 427–430. [24] R. H. Trimm, E. J. Tuck, G. Tuck, M. C. Buncick, M. Kranz, P. Reiner, M. G. Temmen, and P. R. Ashley, “Dynamic MEMS based photonic band-gap filters,” in Proc. IEEE Sens., Jun. 2002, vol. 1, pp. 43–48. [25] T. Kamgaing, “High-impedance electromagnetic surfaces for mitigation of switching noise in high-speed circuits,” Ph.D. dissertation, Elect. Comput. Eng. Dept., Univ. Maryland at College Park, College Park, MD, 2003. [26] F. Yang, K. Ma, Y. Qian, and T. Itoh, “A uniplanar compact photonic-bandgap (UC-PBG) structure and its applications for microwave circuits,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 8, pp. 1509–1514, Aug. 1999. [27] T. H. Kim, D. Chung, E. Engin, W. Yun, Y. Toyota, and M. Swaminathan, “A novel synthesis method for designing electromagnetic band gap (EBG) structures in packaged mixed signal systems,” in Proc. Electron. Compon. Technol. Conf., San Diego, CA, May 2, 2006, pp. 1645–1651. [28] H. M. Greenhouse, “Design of planar rectangular microelectronics inductors,” IEEE Trans. Parts, Hybrids, Packag., vol. PHP-10, no. 2, pp. 101–109, Jun. 1974. [29] HFSS. ver. 10, Ansoft Corporation, Pittsburgh, PA, 2006. [30] M. Swaminathan, J. H. Kim, and I. Novak, “Power distribution networks for system-on-package: Status and challenges,” IEEE Trans. Adv. Packag., vol. 27, no. 2, pp. 286–300, May 2004. [31] J. Qin, O. M. Ramahi, and V. Granatstein, “Novel planar electromagnetic bandgap structures for mitigation of switching noise and EMI reduction in high-speed circuits,” IEEE Trans. Electromagn. Compat., vol. 49, no. 3, pp. 661–669, Aug. 2007. [32] S. A. Chickamenahalli, H. Braunisch, S. Srinivasan, J. He, U. Shrivastava, and B. Sankman, “RF packaging and passives: Design, fabrication, measurement, and validation of package embedded inductors,” IEEE Trans. Adv. Packag., vol. 28, no. 4, pp. 665–673, Nov. 2005.

Telesphor Kamgaing (S’00–M’04–SM’05) received the Diplom.-Ingenieur degree in electrical engineering from the Darmstadt University of Technology, Darmstadt, Germany, in 1997, and both the M.S. and Ph.D. degrees in electrical engineering from the University of Maryland at College Park, in 2003. In 1999, he was a Guest Researcher with the National Institute for Standards and Technology (NIST), Gaithersburg, MD, where he was involved in the modeling and applications of silicon–carbide devices. From 2000 to 2004, he was with the Digital DNA Laboratories, Motorola Inc., Tempe, AZ, where he was involved in the research and development of silicon integrated passives and RF modules for wireless communication. Since 2004, he has been with the Intel Corporation, Chandler, AZ, where he has held several technical leadership positions including Research and Development Technology and Development Manger responsible for the electrical analysis of RF and low density interconnect packaging. Most recently, he has been a Staff Research Scientist with a main focus on radio coexistence on ultra-small form-factor platforms and packaging

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development for millimeter-wave applications. He has authored or coauthored over 50 technical papers in refereed journals and conference proceedings. He holds one U.S. patent with over 15 patents pending. His research interests include CPU power delivery and EBG structures for various applications. Dr. Kamgaing is a senior member of the IEEE Microwave Theory and Technique Society and the IEEE Components and Packaging Manufacturing Technology Society. He was the recipient of numerous awards including a government scholarship, several Best Student Awards and conference travel grants.

Omar M. Ramahi (S’86–M’90–SM’00) received the B.S. degrees in mathematics and electrical and computer engineering (summa cum laude) from Oregon State University, Corvallis, in 1984, and the M.S. and Ph.D. degrees in electrical and computer engineering from the University of Illinois at Urbana-Champaign, in 1986 and 1990, respectively. From 1990 to 1993, he was a Visiting Fellow with the University of Illinois at Urbana-Champaign. From 1993 to 2000, he was with the Digital Equipment Corporation (now Hewlett-Packard), where he

was a member of the Alpha Server Product Development Group. In 2000, he joined the faculty of the James Clark School of Engineering, University of Maryland at College Park, as an Assistant Professor, and later as a Tenured Associate Professor. At the University of Maryland at College Park, he was also a faculty member of the CALCE Electronic Products and Systems Center. He is currently a Professor and the Natural Sciences and Engineering Research Council of Canada (NSERC)/Research in Motion (RIM) Industrial Research Associate Chair with the Electrical and Computer Engineering Department, University of Waterloo, Waterloo, ON, Canada. He holds cross-appointments with the Physics Department and Astronomy and Mechanical Engineering Department, University of Waterloo. He was instrumental in the development of computational techniques to solve a wide range of electromagnetic radiation problems in the fields of antennas, high-speed devices and circuits, and electromagnetic interference (EMI)/electromagnetic compatibility (EMC). He is a consultant to several companies. He was a cofounder of EMS-PLUS LLC and Applied Electromagnetic Technology LLC. His research interests include experimental and computational EMI/EMC studies, high-speed devices and interconnects, biomedical applications of electromagnetics, novel optimization techniques, and interdisciplinary studies linking electromagnetic application to novel materials. He has authored or coauthored over 170 journal and conference papers. He coauthored EMI/EMC Computational Modeling Handbook (Springer-Verlag, 2001, 2nd ed.).

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