On Dynamic Effects Influencing IGBT Losses in Soft ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 1, JANUARY 2011

On Dynamic Effects Influencing IGBT Losses in Soft-Switching Converters Per Ranstad, Member, IEEE, and Hans-Peter Nee, Senior Member, IEEE

Abstract—Two different dynamic effects influencing the insulated gate bipolar transistor (IGBT) losses in soft-switching converters are demonstrated. The first one, the Dynamic tail-charge effect shows that the tail charge is dependent not only on the absolute value of the current at turn-off, but also on the dynamics of the current. This effect may have a significant impact on the optimization of zero-current-switching converters. The Dynamic conduction losses originate from the conductivity modulation lag of the IGBT. It is shown by experiments that the on-state losses depend on the operating frequency. Different methods to accurately determine the on-state losses are evaluated. It was found that the best method is an indirect measurement, where the stray inductance is identified by the use of an oscillating circuit. The experiments are performed under a sinusoidal current excitation at a fixed amplitude (150 A) for different frequencies (up to 104 kHz). The switching devices used are IGBT modules rated 300–400 A/1200 V in a bridge-leg configuration. From the experiments performed, it is found that IGBTs of a modern punch-though (PT) designs have the lowest losses in the series-loaded resonant converters studied in this paper. Index Terms—Conductivity modulation, insulated gate bipolar transistor (IGBT), resonant converters, soft switching, tail current.

I. INTRODUCTION N MANY industrial applications with power electronic converters, it makes sense to operate the converter at highswitching frequencies. There are two main reasons to this fact. The first is that the necessary amount of material in most passive main-circuit components is inversely proportional to the switching frequency, which implies low initial costs at high-switching frequencies. The second reason is that the dynamic bandwidth is usually tightly correlated to the switching frequency, enabling a very good dynamic system performance at high-switching frequencies. However, unless considerable design efforts are made, by different means, to shape the switching waveforms of the semiconductor switches, the switching losses of the converter will be excessive at high switching-frequencies. An attractive solution to this problem is soft-switching [1]–[9]. One class of converters called series-loaded resonant (SLR) convert-

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Manuscript received December 29 2009; revised May 3, 2010; accepted June 14, 2010. Date of current version December 27, 2010. Recommended for publication by Associate Editor E. Santi. P. Ranstad is with the Alstom Power, Växjö S-35112, Sweden (e-mail: [email protected]). H.-P. Nee is with the Kungliga Tekniska Högskolan Royal Institute of Technology School of Engineering, Stockholm S-10044, Sweden (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2010.2055581

Fig. 1.

Circuit diagram SLR.

ers [10]–[12] is frequently suggested for industrial applications with high switching-frequencies (see Fig. 1). When this type of converter is operated above the resonance frequency, the turn-on transitions exhibit both zero-voltage switching (ZVS) and zero-current switching (ZCS) properties. The turn-off transitions, however, are “hard,” and therefore, capacitive snubbers are often employed to reduce the turn-off losses. The capacitive snubber can reduce the turn-off losses considerably, but not eliminate them. The reason to this is that most insulated gate bipolar transistors (IGBTs) have a certain magnitude and duration of the tail current, which creates turn-off losses even though a capacitive snubber is employed. The established method of control of SLR converters is frequency control [10], [11], and [13]. An alternative control method is the phase-shift (duty-cycle) control [14]. However, if these two methods are used in combination, one of the phase legs can be made to turn off at the zero crossing of the current, thus obtaining ZCS [15]–[20]. This control strategy has been named Dual-Control [17]. Fig. 2 shows the typical waveforms when using Dual Control. If the ZCS turn off is assumed to be lossless, as a consequence of a zero turn-off current, the switching losses are reduced by more than 50% compared to frequency control. This is a result of lower switching frequency and that every second turn off is lossless. In [17], this idea has been evaluated experimentally on a 60 kW/25 kHz converter using 1200 V/400 A high-speed non-punch-through (NPT) [21]–[25] IGBTs. It was shown that the reduction in the switching losses was far less than 50% compared to the frequency-control case. The hypothesis of the authors is that the reason to this is the tail current [24]–[26], and its dynamic dependence of the collector current. In [17], also a comparison of the measured losses to the calculated conduction losses is done. The actual current waveforms and the on-state voltage drop, published by the manufacturer,

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RANSTAD AND NEE: ON DYNAMIC EFFECTS INFLUENCING IGBT LOSSES IN SOFT-SWITCHING CONVERTERS

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TABLE I LISTS OF TEST DEVICES

Fig. 2.

Dual control typical waveforms.

are used in the calculations. It was found that the measured losses were significantly higher than the calculated ones. A possible explanation to the additional losses is that the conduction losses exhibit substantial dynamics, i.e., a voltage drop, which depends not only on the instantaneous value of the current, but also on the history of the current [25], [27]–[29]. This is referred to as conductivity modulation lag. If this is the case, different IGBT designs may exhibit more or less of this phenomenon. An interesting consequence of the conductivity modulation lag, as described in [29], is that the resulting effect on the conduction losses would be zero if the forward current is a sinusoidal half wave. According to the experimental results in [17], however, this does not seem to be true. Moreover, [21], [22], and [30] state a coupling between the conductivity modulation lag and the tail current in the case, where the current has a constant magnitude, but variable pulse length. These investigations, however, only consider NPT designs. Since different IGBT designs have different dynamic and static properties, it makes sense to survey how IGBTs presently available on the market perform with respect to conductivity modulation lag and the tail-current dynamic dependence of the collector current [31], [32]. Additionally, it is valuable to indicate, which IGBTs are optimized for the actual application, as, for instance, in [33] and [34]. Studies on switching losses under soft-switching conditions have been performed on IGBTs with a high-blocking voltage capability [35]. Also 1200 V IGBTs have been studied in the past [36]–[38]. In [39], the ZVS case is investigated extensively. Still, however, state-of-the-art components in the 1200 V range suitable for 20–200 kW resonant converters are not covered. New, or improved, IGBTs are presented on a regular basis [40]–[43]. In this paper, therefore, it is the intention of the authors to identify the reason to why ZCS does not imply zeroswitching losses and why the conduction losses may be higher at high-switching frequencies in the case, where the current varies during the conduction cycle, and especially when the waveform is a sinusoidal half wave. Additionally, it is the intention of the authors to identify, which kind of IGBT design that is most suitable for high-frequency soft-switching converters in the 20– 200 kW range. This has already been done for ZVS converters

[44] with an IGBT design of the 1990s. This paper, however focuses on the ZCS case. This paper is organized as follows. In Section II, the test devices and their respective designs are discussed. In Section III, the Dynamic tail-charge effect is investigated. The aspects on the Dynamic conduction losses are presented in Section IV. In Section V, are the combined result of the effects discussed based on experiments on a 60 kW/25 kHz series-resonant converter. Section VI provides a conclusion of the work performed. II. TEST DEVICES Anumber of different IGBT modules have been selected for the measurements. These have been selected on their potential merits as switching devices in resonant converters, as stated in Section I. In order to be able to compare the results among the devices, the ratings are kept as similar as possible. The selected devices are all half-bridge modules with a rated blocking voltage of 1200 V. The current ratings range from 300 to 400 A. Table I lists the tested devices. The selected test devices represent different IGBT technologies [24], [25], and [27]. The type description also includes information on the gate structure, where (P ) indicates a planar structure and (T ) indicates a trench structure. Device A is an NPT-type with lifetime control of the charge carriers. Device B is of a similar design as A. Device C and D include a buffer layer and are thus of the PT design [21]–[25]. They are described by the manufacturer as a trench-field stop design [24], [25], [41], where Device C is said to be “high-speed” and Device D “soft turn-off’.” Device E is a PT-type of an older design. Device F is a “high-speed” PT-type [42], and thus of a similar structure as C and D. Devices G and H are of a similar design as Device A, i.e. NPT. Device I is a similar design as F, but optimized for lower on-state losses. All test devices are using the same drive circuit in the different experiments, i.e., UGE = ±16 V and RG = 3 Ω. III. TAIL CURRENT The aim of this section is to investigate the dynamic tailcharge effect, i.e., how the tail charge QT of the IGBT depends not only on the turn-off current IOFF , but also on the dynamics of the conducted current.

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Fig. 4. Fig. 3.

Test circuit, tail charge.

Collector current. TABLE II LIST OF TURN-OFF CURRENTS AND CONDUCTION ANGLES.

QT is measured for different conduction angles α on a sinusoidal collector current, as shown in Fig. 3 and indicated in Table II. The hypothesis is: if the dynamic tail-charge effect exists, QT has a lower value on the rising slope of the current compared to the falling slope, for the same value of IOFF . This hypothesis will be experimentally validated in this paper. A. Methodology An IGBT bridge leg is used to excite an LC circuit, as shown in Fig. 4. The gate signals are timed in such a way that T2 , which serves as the device under test (DUT), is turned off at different occasions on the first half period of the sinusoidal current. QT is measured at five different conduction angles α, as listed in Table II. Initially, T1 is in the on-state, keeping C fully charged. The oscillation is initiated by the switching from T1 to T2 . The length of the gate pulse G2 determines the instant of the turn off of T2 . The frequency f0 and the amplitude of the current ˆı, are given by (1) and (2) 1 √ . 2π LC E ˆı = L/C

f0 =

(1) (2)

where L and C are the inductance and capacitance values of the resonant tank and E is the dc-link voltage of the bridge. The tests are performed with ˆı = 150 A at two different values of f0 30 and 104 kHz. Fig. 5 shows typical measured waveforms.

Fig. 5. Tail-charge measurement. Tr1: G2 (10 V/div), Tr2: I (100 A/div), Tr3: IE (10 A/div), and time: 0.5 μs/div.

The test is repeated at two different junction temperatures Tj (25 ◦ C and 125 ◦ C) in order to investigate the impact of the temperature. The high-temperature turn-off performance of early generation IGBT designs has also been studied in [46]. It was found that the tail current increased significantly when increasing the temperature. Three test Devices (A, C, and D) are compared (see Table I). B. Results The results of the tail-charge measurements are shown in Figs. 6–9. The current waveform (dotted line) is shown in the figures as a reference for the reader. Fig. 6 shows that QT does not follow IOFF for Device A. At 25 ◦ C QT (5) is 50% higher compared to QT (1). In addition, QT (3) < QT (5). At 125 ◦ C, the same behavior, even more pronounced, is observed. It is found that QT (5) is 100% higher than QT (1). From these observations, it is evident that QT has a dynamic dependence of the collector current, when operated at 150 A and 30 kHz. The results shown in Figs. 7 and 8 indicates that no dynamic dependence of QT can be observed for Devices C and D when operated at 150 A and 30 kHz. The tail-charge curve shows a very good symmetry (QT (5) = QT (1) and QT (4) = QT (2)) with respect to the peak of the current at 90◦ . The tail-charge curve conserves the sinusoidal shape of the current. Hence, the tail charge depends only of the absolute value of IOFF (within the accuracy of the measurement). In order to investigate the


Fig. 6.

Fig. 7.

Device A, Q T versus α at 30 kHz.

Device C, Q T versus α at 30 kHz.

behavior at higher frequencies, Device C was also measured at 150 A and 104 kHz. The result is shown in Fig. 9. It appears that at this frequency, the tail-charge curves are nonsymmetrical, i.e., QT (5) > QT (1) and QT (4) > QT (2) for both 25 ◦ C and 125 ◦ C operation. Thus, it is indicated that QT has a dynamic dependence of the collector current when measured at 150 A and 104 kHz. C. Discussion From the measurements, it is found that all the investigated Devices (A, C, and D) show evidence of the dynamic tail-charge effect. However, the onset of this effect occurs at different frequencies. For Device A, which is an NPT device, the effect is evident already at 30 kHz. For Device C, which is a PT (trench gate, field stop) design the effect is found at the frequency 104 kHz. When evaluating the losses due to the tail current, the absolute values of QT have to be considered. A comparison of the absolute values of QT for Devices A and C shows that Device A generally has 50% lower values of QT than Device C. However, if it is possible to switch Device C at the optimum switching instant, it is possible to obtain a very low QT at frequencies up to 100 kHz. This means that even though Device C generally has a higher QT compared to Device A, at high-switching frequencies and ZCS, it shows a lower absolute value of QT . It is the opinion of the authors that the onset frequency is determined by the time constant of the conductivity modulation. It is assumed that at frequencies well above the onset frequency

Fig. 8.

Device D, Q T versus α at 30 kHz.

Fig. 9.

Device C, Q T versus α at 104 kHz.

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QT will show lower values. However, at these frequencies, the on-state losses will increase drastically. The conductivity modulation will have a minor impact on the forward voltage drop, only the MOSFET channel of the IGBT is active. In order to determine the optimal turn-off instant of the IGBTs in the ZCS leg in the converter of Figs. 1 and 2, a set of experiments are performed. In different publications [34], [45], it has been shown that the minimum loss is achieved when switching off slightly before the zero crossing of the current. Fig. 10 shows the total losses in the ZCS bridge leg as a function of IOFF . The converter, rated 60 kW/540 Vdc , is operated at a load point of U0 = 54 V and I0 = 120 A. The switching frequency is 22 kHz and Tj is 60 ◦ C. IOFF is defined as the value of the collector current at the negative edge of the gate signal. The losses of Devices A and C are compared. As shown in Fig. 10, Device C can be operated at significantly lower losses compared to Device A. In order to achieve the minimum losses of Device C, IOFF has to be carefully tuned, while Device A is less sensitive to variations in IOFF . Obviously, Device A has a more significant tail-charge effect than Device C at this operating frequency and temperature. This is, in accordance, with the measurements of QT , as shown in Figs. 6 and 7. IV. CONDUCTION LOSSES This section of the paper deals with the dynamic part of the on-state losses. The aim is to investigate the dynamic behavior of the conductivity modulation. The IGBT is conducting a half

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Fig. 12. Excitation of resonant tank. Tr1: IE (50 A/div), Tr2: U C E (2 V/div), and time: 0.2 ms/div.

Fig. 10.

Fig. 11.

IGBT losses versus IO F F .

Fig. 13. Measurement of u C E (t) and iE (t). Tr1: iE (t) (100 A/div), Tr2: u C E (t) (2 V/div), and time: 5 μs/div.

Test circuit.

period of a sinusoidal current. It is expected to observe a frequency dependence of the conduction losses. This is referred to as dynamic conduction losses. A. Methodology The dynamic properties of the on-state voltage are studied under a sinusoidal current excitation. The current and voltage of the DUT are recorded. The instantaneous power loss is calculated as p(t) = i(t)u(t).

(3)

The IGBT conduction losses are calculated as the average of p(t) as 2 T /2 P = p(t)dt (4) T 0 where T is the period time of the sinusoidal current. The conduction losses P are calculated at varying temperatures and frequencies and are compared among the different test objects. The test circuit shown in Fig. 11 is used. It consists of two halfbridges connected to a series-resonant tank. The switches T1 /D1 and T2 /D2 are used to initiate an oscillation in the resonant tank. The switch T4 /D4 serves as the DUT and is continuously in the on-state. The on-state voltage UCE and the conducted current IE of T4 /D4 are recorded. The switch T3 /D3 is continuously in the off-state. As defined by (1) and (2), E is used to control the amplitude of the current and the resonant tank (L and C) determines the frequency.

Figs. 12 and 13 show oscillograms of the test circuit. Fig. 12 shows the oscillation of the tank current. The duration of the current is in the range of 1 ms. In order not to, significantly, heat the chip, the oscillations are repeated at a low frequency 0.5 Hz. The junction temperature Tj is kept constant during the tests. Tj is controlled by controlling the temperature of the heat sink. Fig. 13 shows iE (t) and uCE (t) during one of the periods of the oscillation. The voltage uCE (t) is measured at the power terminals of the module. E is adjusted such that the peak value ˆıE of IE is 150 A. This value is kept constant for all measurements. In order to minimize the errors in the power loss calculation, it is vital to keep the difference in delays between the recorded values of iE (t) and uCE (t) low. An important aspect of the measurements is an accurate measurement of uCE (t). At the actual testing conditions, the inductive voltage drop of the stray inductance LS of the module is significant. Modeling of the stray inductances is discussed in [47] and [48]. Fig. 14 shows an equivalent circuit of a power module. Since the test is performed on the lower switch, the current flows from the C2/E1 terminal to the E2 terminal. As a consequence, the equivalent circuit can be simplified, as shown in Fig. 15, where all relevant inductances have been lumped into one LS . Two different methods to measure parasitic elements of MOSFET modules are discussed in [48]. The first one uses an impedance analyzer. This is not applicable in the actual case as the circuit includes the nonlinear impedance of an IGBT. In the second method, stray capacitances are identified by the use of an external inductance and a measurement of the resulting resonance frequency. In this paper, this method has been adapted


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Fig. 16. Test setup. (a) Uncovered IGBT module. (b) Detail of probe connection.

Fig. 14. Equivalent circuit of a half-bridge IGBT module including stray inductances.

Fig. 17. Direct measurement of U C E . Tr1: iE (t) (50 A/V), Tr2: u C E 1 (t), Tr3: u C E 2 (t), and Tr4: u L s (t).

Fig. 15.

Simplified circuit.

to the measurement of stray inductance by the connection of an external capacitance (see Method 4 in the following). Four different methods to measure uCE (t) have been evaluated. 1) Direct measurement, on the IGBT chip. 2) Direct measurement, through the gate connections. 3) Indirect measurement, identification of LS by a current ramp. 4) Indirect measurement, identification of LS by an oscillating circuit. The Methods 1 and 2 aim for a measurement of uCE (t), not including the voltage drop of LS , while Methods 3 and 4 compensate for LS in the postprocessing of the recorded data. 1) Direct Measurement, on the IGBT-Chip: The power module is opened, as shown in Fig. 16(a), and a dedicated voltage probe is used to measure the voltage directly on the chip, as shown in Fig. 16(b). The bonding pads at the surface of the chip and the copper foil to which the chip is soldered are used as measurement points. The silicone gel, which protects the different chips inside the module, is penetrated in order to make contact. This method gives excellent measurement results, but it also drastically reduces the blocking capability and the relia-

bility of the device. The result of the measurement is shown in Fig. 17. The voltage uCE1 (t) is measured at the power terminals of the module, while uCE2 (t) is measured directly on the chip. The voltage uL s (t) is calculated as the difference of uCE1 (t) and uCE2 (t). As shown in Fig. 17, uL s (t) is leading by 90◦ . Hence, the parasitic impedance is inductive and the inductance LS is determined by LS =

u ˆL s 2πfˆıE

(5)

where f is the frequency, ˆıE is the peak value of iE (t), and u ˆL s is the peak value of uL s (t). 2) Direct Measurement, Through the Gate Connections: An analysis of the layout of the opened device, used in Method 1, yields that the inductances LSE1 and LSC2 in Fig. 14 are small. Additionally, the DUT is continuously on which implies that the gate current is constant and very low ( 0), the time needed to fully develop the conductivity modulation causes an additional voltage drop. In a similar way, for a decreasing current (diE (t)/dt < 0), the voltage drop is reduced due to the excess of charges. At the lower frequency 7.5 kHz, these two effects are balancing and do not add significantly to the losses. However, at the highest frequency 104 kHz, the increase of the on-state voltage during the first part of the sinusoidal current is heavily dominating over the decrease during the latter part. As a consequence, the power loss is significantly higher at the highest frequency. Hence, the measurement proves the frequency dependency of the on-state losses, referred to as dynamic conduction losses. It is also evident, from a comparison of Devices A and C, that the frequency response is different among the devices. The operational temperature of the chip also affects the frequency response. D. Results The IGBT conduction losses P have been determined using (2). Fig. 28 shows the resulting losses for the different test


Fig. 27. Device C, U C E , Tr1: dc eqv, Tr2: 7.5 kHz, Tr3: 30 kHz, and Tr4: 104 kHz. (a) T j 25 ◦ C. (b) T j 125 ◦ C.

Fig. 28. IGBT conduction losses; 1: 7.5 kHz 25 ◦ C, 2: 30 kHz 25 ◦ C, 3: 7.5 kHz 125 ◦ C, and 4: 30 kHz 125 ◦ C.

devices in Table I. The losses are presented, for each device, at two different frequencies 7.5 and 30 kHz, and at two different temperatures 25 ◦ C and 125 ◦ C. When analyzing Fig. 28, it is evident that all devices have a higher conduction loss at the higher frequency compared to the lower one. This is valid for low temperature as well as for high temperature. It is also observed that, for all devices, the loss increase is higher at the higher temperature. It is also


Fig. 29. IGBT conduction losses; 1: dc equivalent, 2: 7.5 kHz, 3: 30 kHz, and 4: 104 kHz.

observed that for some Devices E and F, the conduction losses have negative temperature dependence. The conduction losses decrease at higher temperatures. This is typical for PT devices [24]. In order to investigate the dynamic conduction losses in a wider frequency range, two devices were selected. Devices A and C were studied at 104 kHz as well as at a very low frequency dc equivalent. The result is shown in Fig. 29. The analysis of Fig. 29 indicates a better high frequency performance of Device C compared to Device A. This is valid both regarding the absolute and the relative loss increase when comparing 30 and 104 kHz operation. This result is the same at 25 ◦ C as well as at 125 ◦ C operational temperatures.

E. Discussion The IGBT conduction losses have been measured and compared among different designs, each design being represented by one single specimen. In order to estimate the variation of the losses among different specimens of the same design, a statistical approach has been used. In a production process (60 kW resonant converters), the IGBT losses are measured, as a part of the quality control procedure. From a population of 96 units, representing 192 IGBT modules, the standard deviation of the losses was 2.74%, while the standard deviation of the load current was 1.0%. Hence, it is the opinion of the authors that the results of the measurements, on single specimens, presented in this paper can be generalized to the different IGBT designs. As discussed in Section IV-A, any difference in the delays of the recorded signals iE (t) and uCE (t), may contribute to the total error of the power loss calculation. In the actual test setup, the maximum value of the difference is estimated to 100 ns, which at 104 kHz corresponds to a phase angle α = 3.7◦ . Consequently, the contribution to the total error may be estimated from P = S·cosα = S·0.998, which indicates a contribution