Power Adaptor Device for Domestic DC Microgrids ... - IEEE Xplore

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 60, NO. 3, MARCH 2013

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Power Adaptor Device for Domestic DC Microgrids Based on Commercial MPPT Inverters Hugo Valderrama-Blavi, Member, IEEE, Josep M. Bosque, Student Member, IEEE, Francisco Guinjoan, Member, IEEE, Luis Marroyo, Member, IEEE, and Luis Martínez-Salamero, Member, IEEE

Abstract—This paper presents a power adaptor device, referred to as smart panel device, allowing the connection of additional energy sources and storage elements to a domestic photovoltaic (PV) grid-connected system. The adaptor output port is designed to behave as a power source/sink, thus enabling its hot-swap parallel connection to renewable power sources without modifying their maximum power point (MPP). Moreover, the adaptor device features a power characteristic with a single controllable MPP and allows the control of the injected power within the operating range of the dc–ac grid-connected inverter. The work presents the design principles of such device by describing the operation of a slidingmode controlled quadratic-boost converter. The proper operation of the device is experimentally verified for several scenarios in a small PV-based microgrid system including a fuel-cell stack, a 1-kW three-phase wind turbine, a battery charger–discharger, and commercial grid-connected PV inverters. Index Terms—Active grid cooperation policies, distributed power generation, energy storage, power converters, PV systems, sliding-mode control.

I. I NTRODUCTION

R

EDUCING economic dependence from third countries, fossil fuel consumption, and greenhouse-effect gas emission have caused the expansion of renewable energies. Some years ago, however, that expansion was seriously limited by the high cost of such systems and the lack of mature technologies. Progressively reducing costs, increasing efficiency, and particularly institutional subsidies have been promoting such energies, namely, wind and photovoltaic. For instance, the European Union has scheduled a very ambitious goal for 2020. By this date, renewable energy production must

Manuscript received September 14, 2011; revised January 5, 2012 and March 15, 2012; accepted April 18, 2012. Date of publication May 7, 2012; date of current version October 16, 2012. This work was supported in part by the Spanish Ministry of Science and Innovation, and by the European Union Program for Regional Development (FEDER), under the grants DPI200914713-C03-01, DPI2009-14713-C03-02, DPI2009-14713-C03-03, and RUE CSD2009-00046 from the 2010 Consolider-Ingenio Program. H. Valderrama-Blavi, J. M. Bosque, and L. Martínez-Salamero are with the Department of Electrical Engineering, Universitat Rovira i Virgili, 43007 Tarragona, Spain (e-mail: [email protected]; [email protected]; [email protected]). F. Guinjoan is with the Department of Electronic Engineering, Universitat Politècnica de Catalunya, 08034 Barcelona, Spain (e-mail: francesc.guinjoan@ upc.edu). L. Marroyo is with the Department of Electrical and Electronic Engineering, Universidad Pública de Navarra, 31006 Pamplona, Spain (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2012.2198038

achieve a share of 20%. In addition, greenhouse emissions should be reduced by the same amount, taking as reference the 1990 levels [1]. Currently, the expansion limiting factor is quite different. The stochastic production and the passive grid behavior of such systems can cause severe control problems to the grid if the ratio between renewable and conventional plants increases beyond a certain threshold [2]. To favor grid reliability and, consequently, renewable energy integration and expansion [3], it is mandatory to participate in active grid policies. This means that renewable plants also contribute to grid management and control [4]–[9]. Distributed generation systems [10] reduce greatly the transport losses and production randomness. New regulations on wind systems require that such systems have sufficient reactive reserves to compensate voltage sags [11]. Although these policies can help, other active policies must be taken for increasing grid stability and renewable promotion. Existing domestic grid-tie PV systems would also contribute to these policies provided that additional sources and storage elements could be connected to the PV system without perturbing its maximum power point (MPP) operation fixed by the PV inverter. In this sense, the design of a power stage properly interfacing these additional elements to the grid connected system has no antecedents in the technical literature to the authors’ knowledge. This paper presents a power adaptor device, referred to as smart panel device (SPD), allowing the hot-swap connection of additional sources, loads, and storage elements in existing domestic grid-tie PV systems preserving the MPP operation of the solar field. The SPD linking these additional elements to the PV system is based on switching converters controlled as power sources/sinks featuring a smart voltage-to-power curve which approximates the characteristics of a solar array. The remainder of this paper is organized as follows. In Section II, the three basic operation modes of the SPD are discussed. Some SPD applications and benefits are explained in Section III. In Section IV, the SPD is proposed as a building block to develop three dc-microgrid architectures. The implementation of the smart PV panel curve is described in Section V. An example of SPD design, for a low-voltage fuel cell, is presented in Section VI. Section VII is devoted to experimentally verify the proper operation of the SPD under different scenarios in a domestic PV-based microgrid system including a fuel-cell stack, a 1-kW three-phase wind turbine, a battery charger–discharger, and a 2-kW commercial grid-connected PV

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inverter. Finally, the last section draws the conclusions of this paper. II. SPD O PERATION M ODES Commercial maximum power point tracking (MPPT) inverters are the key element in a grid-connected PV system. These inverters track the solar field MPP only when the voltage is inside the MPPT range. In fact, this means that any source, not only solar panels, with the adequate voltage and a power–voltage (P −V ) curve similar to a solar array will be compatible with the MPPT inverter. Additional information about MPPT systems and applications, including solar inverters, can be found in [12]–[19]. If the source has not the adequate voltage level and P −V curve, an adaptor would be required. If the adaptor output port behaves as a controllable power source with a similar P −V curve of a solar array, any source/sink device can be easily connected to the grid-connected PV system preserving the standard operation of the MPPT inverter. Assuming this behavior, the grid-connected PV system can operate in three operation modes as described in the following. A. First Operation Mode: Additional Source With No PV Array Power Generation Enabling any kind of source to be compatible with an MPPT inverter is one of the key issues of this work. This is illustrated in the circuit of Fig. 1(a), which corresponds to the first operation mode. Indeed, the original motivation for developing the SPD concept was to inject energy during the night, when there is no solar generation. Among the additional features of the SPD, the adaptor input conductance can be controlled. As a result, the power extracted from the source and injected into the grid becomes fully controllable. Moreover, the SPD can also adapt the voltage of the source to fall within the MPPT inverter voltage input range. As the adaptor output has a power source characteristic [20], the parallel connection of the output ports of several adaptors is realizable, and different sources can be processed by a single MPPT inverter. According to Singer et al. [20], [21], the equivalent circuit of a parallel or series connection of various power sources and/or sinks is a single power source totalizing the contribution of the partial powers delivered by the diverse sources. Consider now that diverse adaptor sources have an MPP. As the adaptor output port is a power source, all the MPPT algorithms associated to the sources are totally decoupled, thus avoiding interferences among the diverse sources. Finally, the power source characteristic of the adaptor output simplifies the inclusion and extraction of the sources, loads, and storage elements. Thus, the architecture changes can be done without shutting down the MPPT inverter or the remaining adaptors. This hot-swapping feature reduces the energy loss during the installation maintenance. As shown in Fig. 1, the adaptor output port behaves as a power source, where the power delivered to the inverter PO (VO , Gg ) can be controlled varying the adaptor input conductance Gg since P0 = Gg .Vg2 .

Fig. 1. SPD operational modes. (a) First mode. (b) Second mode. (c) Third mode. (d) Power source and power sink I-V characteristics.

Fig. 2.

SPD operation in the first canonical mode.

Nevertheless, the adaptor output port is controlled so that the voltage-to-power curve at the inverter input is a piecewise approximation of a solar array P −V curve shown in Fig. 2, whose expression is given in  O PO (VO ) = Pref · VVAB 0 < VO ≤ VAB    PO (VO ) = Pref
  VAB < VO ≤ VBC  O −VBC VBC < VO ≤ VCD . PO (VO ) = Pref · 1 − VVCD −VBC    PO (VO ) = 0 VO > VCD   2 PO (Gg , VO ) = Gg · Vg 0 < VO ≤ VCD (1) In Fig. 2, PO stands for the SPD output power, whereas Pref is the reference power level to be delivered by the new source

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of the third mode where, now, PINV = Pp + P0 with P0 < 0. Note that power sinking is only possible if there is sufficient solar power. If the power to be sunk Pref has an absolute value higher than the panel power Pp , all the power is sunk, and the inverter has no power to inject in the grid; see case Pp3 + PO in Fig. 3(c). On the other hand, the solar panel operating points (a, b, c) remain unchanged since a flat region B (see Figs. 2 and 3) has been designed to include all solar panel MPPs by choosing the values of VAB and VBC to be as much separated as possible. Note that adding sources using the second mode is always possible, but for the third canonical mode, power sinking will be only possible if there is sufficient solar power generation. The remaining SPD design criteria are as follows: 1) VIU < VAB to avoid repeated inverter shutdowns and time-consuming restart procedures, and finally, 2) VCD must be near VOC , the panel open circuit voltage. III. SPD A PPLICATIONS AND B ENEFITS In this section, we investigate different applications using the SPD to show its wide possibilities. A. Any Source for Any MPPT Inverter

Fig. 3.

SPD operation in the second and third canonical modes.

Vg . The MPPT inverter range is defined by the under and over voltage limits VIU and VIO , respectively, and VAB , VBC , VCD are the voltage levels defining the piecewise approximation and the operating regions A, B, C, and D. The selection criteria of these voltages are addressed in the next section.

This application lay directly on applying the first canonical mode. Frequently, the source and the available inverter have incompatible voltage ranges, and as the SPD includes a power converter, this inconvenience is cleared. Aside from matching voltage ranges, the adaptor output exhibits a convex piecewise curve with its own MPP; thus, any source having or not having an MPP becomes compatible with a commercial MPPT inverter. In addition, if the source has its own MPP, any kind of MPPT algorithm can be used to modulate the input conductance Gg so that the SPD input will track this MPP at any time.

B. Second and Third Operation Modes: Additional Sources/Loads With PV Array Power Generation The second and third canonical modes are shown in Fig. 1(b) and (c), respectively, and deal with the connection of an additional source/load when the PV array is generating power. The second mode shown in Fig. 1(b) corresponds to the connection of an additional power source, whereas the third one enables the connection of a power sink. Note that the power source symbol used at the inverter input port in Fig. 1(b) is changed to a power sink symbol in Fig. 1(c). The power currentversus-voltage curves of a power source/sink [20] are shown in Fig. 1(d). Using the adequate mode, any kind of source/load or storage element (in charge/discharge mode) can deliver/sink a certain power PO from the system, without modifying the PV array MPP. This feature is shown in Fig. 3, where Fig. 3(a) shows the P −V curves of a solar array Pp1 , Pp2 , and Pp3 under different irradiances. Fig. 3(b) shows the P −V curves corresponding to the second mode, when a solar panel is parallel connected with an SPD source, thus leading to an overall input power PINV = Pp + P0 with P0 > 0. Fig. 3(c) shows the operation

B. Creating an Energy Reservoir The improvement of a previous grid-connected PV plant, allowing the system to implement several active cooperation policies, requires the addition of an energy reservoir. This can be made using adequately the three SPD canonical modes, using as many adaptors as required sources or sinks, and connecting their outputs in parallel to add all contributions. According to Singer et al. [20], [21], the equivalent circuit of a parallel or series connection of several power sources and/or sinks is a single power source whose power corresponds to the overall power of the connected sources. For instance, during night, when there is no PV production and the energy reservoir is being discharged, the SPD operates according to the first canonical mode [see Fig. 1(a)]. During the day, if the solar panels are producing more energy than required for a certain scheduled grid-injection profile, the remaining power can be used to recharge the energy reservoir, using the third canonical mode [see Fig. 1(c)]. Finally, if during the day, the PV production is under the scheduled injection, both the

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Fig. 4. Irradiation and consumption profiles around 40◦ N [22], [23].

solar panels and the reservoir can deliver the required energy, as proposed in the second canonical mode.

Fig. 5.

Irradiation at diverse U.S. sites [27]. TABLE I Y IELD G AINS AT VARIOUS C ITIES [22], [26]

C. Enhancing Inverter Use During a day, solar production does not match necessarily with electrical consumption. Fig. 4 shows some consumption and generation profiles [22], [23] during a day. The consumption profiles belong to Spain (36◦ N–43◦ N), whereas the irradiation ones come from a 40◦ N latitude site with the panels south oriented with a tilt equal to the latitude. In Fig. 4, the curves with legends PW and PS are respectively the irradiation profiles at December and June, and CPm and CPa are constant grid-injection profiles adapted to the maximum and average June irradiations. DW and DS are the December and June domestic consumption profiles, and finally, GW and GS are the December and June global consumption profiles, scaled a factor of 1/3, to fit the graphic. As expected, global profiles are flatter because of the industrial customers. By simple inspection of the CPm, PW, and PS profiles, we can deduce than the inverter capabilities are seriously wasted. As the inverters are sized according to the solar field, in a gridconnected PV plant, most of the day, the inverters are working at low power where they are less efficient, and they are stopped during the night. Additionally, in winter where there are less irradiation hours, the situation is even worse. Using the SPD, new energy sources and storage elements can be incorporated to the system creating an energy reservoir, and then, the power injected to the grid becomes controllable, decoupling the grid injection from the weather and day cycles. Using an energy reservoir, the following diverse injection profiles now become possible: 1) constant injection at diverse levels, for instance, CPm and CPa; 2) injection tracking a given consumption profile; and 3) collaboration in primary, secondary, or tertiary regulation tasks [24], [25]. D. SPDs and Inverter Sizing Ratio As previously addressed, a grid-tie connected inverter is working most of the time at low or even zero power, causing

an important efficiency loss due to both day cycling and year cycling. To enhance the inverter use reducing the influence of day cycling, we have proposed to include an energy reservoir in the system using SPD devices, but to compensate long-term or year cycling, the inverter rating must be adapted to the site latitude [26]. Solar resources are strongly dependent on the latitude, as shown in Fig. 5. As the latitude is lower, the available energy per day is more uniform during the year. The data shown in Fig. 5 belong to diverse U.S. sites [27], from Puerto Rico to Alaska, with the exception of latitude 50◦ N, obtained averaging data from 45◦ and 55◦ sites, as latitude 50◦ N is outside U.S. borders. At all the sites, the solar arrays were south oriented with a tilt equal to the latitude site. According to Velasco et al. [26], the sizing factor is defined as the ratio between the inverter rated power and the installed solar field. Moreover, to optimize the PV plant, the sizing factor must be higher at low latitudes and, conversely, lower at high latitudes. For instance, at 40◦ N, it is recommended that the inverter power rating be equal to the solar field  M YD = 24 · SF (2) = M YD , Y = M YD . Y G min

Imax

G max

Imin

In Table I, the sizing factor SF and irradiation densities of certain sites are given. Using these data, the maximum daily yield M YD is defined in (2) as the maximum energy per day and surface unit that can be injected into the grid using an inverter

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with the appropriate sizing ratio. Moreover, the yield gain YG also defined in (2) is the ratio between that maximum and the energy deliverable from such irradiation levels. As can be seen, in Table I, all the minimum yield gains YGmin are quite similar, but the maximum gains YGmax are quite different, as expected from the low irradiation levels at high latitude winters. This can be improved using the SPD to incorporate alternative energy sources as wind generators. Finally, it is worth noting that, at high latitudes, creating an energy reservoir using only storage elements may become expensive in cost and size.

IV. DC M ICROGRIDS : SPD AS B UILDING B LOCK In the previous section, the main applications and advantages using the SPD are shown. Maybe the most important is the possibility of creating energy reservoirs adding different energy sources and storage elements [28]. As a result, the PV-based power plant becomes a dc microgrid. In this section, three microgrid architectures using SPDs are explained.

A. Multiple-SPD Grid-Tie Derived DC Microgrid This architecture is derived directly from adding elements to a simple grid-tie PV plant, using an SPD for each source, storage element, or load. The only irreplaceable element is the MPPT inverter. The remaining elements can be connected and disconnected, and the system remains working. Consequently, this is the most modular architecture. Furthermore, the main solar field does not need a converter [29], this being an additional feature of this architecture.

B. Stand-Alone Derived DC Microgrid With One SPD The second architecture, shown in Fig. 6(b), addresses a previous off-grid microgrid based on a central battery dc bus. By adding a conventional MPPT inverter using a single SPD, the stand-alone system becomes grid connected.

C. Single-SPD Grid-Tie Derived DC Microgrid This microgrid architecture is a hybrid version of the previous ones. It is also based in a simple grid-tie PV plant, but as in the second architecture, it is based in a central battery. Comparing the previous architectures, we can observe some differences. The first architecture [see Fig. 6(a)] is the most modular since it allows an easy expansion of the microgrid by connecting additional elements with their own SPD. The other structures [see Fig. 6(b) and (c)] use the SPD only as an interface to connect the main battery to the MPPT inverter. In addition, the use of a single SPD facilitates the grid active policy implementation due to the reduction in the communications usually required in this kind of systems [30], [31]. Finally, it can be pointed out that the storage management [4]–[7], [32], [33] will be a critical issue in all of these architectures in order to assure the feeding of critical loads and energy reservoir.

Fig. 6. (a) Multiple-SPD grid-tie derived microgrid. (b) Stand-alone derived dc microgrid using one SPD. (c) Grid-tie derived dc microgrid using one SPD.

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V. SPD G ENERAL D ESIGN C ONSIDERATIONS The adaptor (SPD) device consists of two parts: a power converter and a control system. The first matches the voltage between the source and the inverter, and the second forces the adaptor output to behave according to the shape shown in Fig. 2. Since this shape approximates a solar panel curve, the device and the control loop will be referred to as “SPD” and “panel shaper,” respectively. A. Converter Selection The power converter must be selected for each application and mainly depends on the voltage difference between the source and the MPPT inverter, which can lead either to buckor boost-derived type topologies. Galvanic insulation, source immunity to pulsating currents, and, finally, power flow bidirectional requirements are other important factors. For instance, to manage a battery, the converter must be bidirectional and will be able to operate according to any of the three canonical modes. The converter control is based on the loss-free resistor (LFR) concept. As stated in [21], the output ports of LFR devices behave as power sources and can be connected in series or in parallel [20]. This feature is very useful when adding new elements to the system since it assures that different sources and loads are decoupled [34]. B. Panel Shaper The realization of the shape shown in Fig. 2 approximating a solar array P −V curve will be undertaken by means of a sliding-mode control of the SPD [35], [36]. Sliding control has been selected for its implementation simplicity by means of hysteretic comparators and for its inherent robustness against parameter variations [37]–[39]. A simple and stable control design to regulate the energy flow in switching converters can be achieved by controlling the current of the inductor acting as an energy buffer. With this goal, the panel shaper implementation must take into account the converter selection since, for buck-based converters, the inductor current control corresponds to the control of the SPD output current. Conversely, for boost-based converters, the inductor current control corresponds to the control of the SPD input current (the case of a buck–boost converter is addressed in Section V-B3). In addition, the SPD efficiency results in a slight modification of the piecewise characteristic, as addressed in the following. 1) Buck-Derived Converters: As previously stated, buckbased converters lead to a direct SPD output power control. In this case, the P −V curve seen by the MPPT inverter is

SA (X) : iL = Iref (Pref , Vg , Vo ) = SB (X) : iL = Iref (Pref , Vg , Vo ) = SC (X) : iL = Iref (Pref , Vg , Vo ) =

Pref VAB Pref VO Pref VO

SD (X) : iL = Iref (Pref , Vg , Vo ) = 0

Fig. 7. SPD (gray) input and (black) output power–voltage curves. (a) Buckderived type SPD converter. (b) Boost-derived type SPD converter.

the trapezoidal shape given in Fig. 2 and reproduced here in Fig. 7(a) in black color. Notice that the SPD input power PIN , in gray color, is slightly higher than the output power PO due to the converter’s efficiency. Considering the first operation mode [see Fig. 1(a)], the inverter MPPT will find the MPP around VAB or VBC points. As regards the sliding-mode control design, the control surfaces for the four regions A, B, C, and D given in (3) shown at the bottom of the page can be derived from (1) considering that PO = VO · IO , where IO is the output current. As can be deduced from (3), the output power control is achieved forcing the converter output current (i.e., iL ) to track a given reference corresponding to the panel shape. 2) Boost-Derived Converters: In this case, the control acts on the SPD input power (PIN ) tracking the trapezoidal waveform depicted in gray in Fig. 7(b). Due to the SPD efficiency η, the P −V curve seen by the MPPT inverter will verify P0 = η.PIN and is shown in black in Fig. 7(b). The MPP of this curve has a controllable power PAM given in (4), with VAM being the MPP voltage, which is the point inside the flat region B where the adaptor exhibits its maximum efficiency. At this point, the adaptor voltage gain is VAM /Vg , and the MPP coordinates are (PAM , VAM ). As the inverter MPPT will find this point, the SPD will operate at its maximum efficiency PAM = PO (VAM ) = η · Pref (VAM ) = η · Vg2 Gg .

(4)

The control surfaces for boost-derived topologies are given in (5), shown at the bottom of the next page, and can also be deduced from (1) considering now that PO = ηVg Ig , where Ig is the averaged value of IL . The input power control is achieved


· 1−

VO −VBC VCD −VBC



 0 < VO ≤ VAB    VAB < VO ≤ VBC  VBC < VO ≤ VCD     VO > VCD

(3)

VALDERRAMA-BLAVI et al.: POWER ADAPTOR DEVICE FOR DOMESTIC DC MICROGRIDS

forcing the converter input current to track a given reference corresponding to the panel shape. The design of the sliding-mode control laws corresponding to the surfaces given in (3) and (5) can be undertaken according to the principles given in [37] and [38]. In this regard, a detailed design procedure is given in Section VI for a quadratic-boost converter. In addition, the conditions ensuring a sliding motion over the previous surfaces for buck and boost converters are given in Table IV. The implementation of the four surfaces requires sensing the adaptor output voltage V o and then selecting the appropriate surface by sending to the hysteretic comparator the respective current reference IRef for the state variable IL . 3) Buck–Boost Derived Converters: Some applications would require the SPD to step up and step down the voltage of the additional source, as in the case of a battery which voltage range during the charging–discharging process can be higher or lower than the inverter input voltage operating range. Although the simplest choice is to design an SPD by means of a buck–boost converter, this solution entails the following drawbacks. 1) The input and output current discontinuities increase both resistive losses and electromagnetic interferences. 2) Neither the input nor the output power can be directly controlled from the inductor current because the inductor is switching between the input and output ports. However, the sliding-mode control is still possible through the following switching surface: S(X) = IL = Iref (Pref , Vg , V0 )

Fig. 8. Quadratic-boost SPD.

amplifiers as described in [36] or digitally using a low performance controller [35], [40]. VI. E XAMPLE OF SPD D ESIGN The proposed design example deals with a low-voltage fuel cell (4 V, 25 A) that must be adapted to a domestic gridconnected PV system [35]. The solar field is formed by four series-connected PV panels with a rated power of 680 Wp and an open circuit voltage VOC around 170 V. The PV panels are directly connected to a 2-kW PV inverter. A. Converter Selection: Quadratic-Boost Converter In this case, the voltage difference between the panels and the source is important, and the converter must have a high voltage gain. A suitable topology supporting this high voltage gain is the cascade connection of two elementary boost cells as shown in Fig. 8, where the main switches M 1 and M 2 operate in synchronous mode. As the steady-state voltage gain (9) MQB has a quadratic dependence with the complementary duty ratio, this converter can be named quadratic boost

(6)

where, now, the current reference Iref for the buck–boost inductor can be obtained introducing an outer loop with a proportional integral controller defined as t Iref (Pref , Vg , V0 ) = KP · eP (t) + Ki eP (τ )dτ (7) eP (t) = Pref (V0 ) − V0 I0

MQB =

(8)

where eP (t) stands for the power error between the reference power Pref (VO ) of the panel shape and the real output power (8). It is worth noting that this solution entails a reduction of the panel shaper bandwidth. Alternatively, the step-up/step-down conversion can be carried out by selecting conversion topologies including an in´ ductor at the input port, such as Cuk or cascaded boost–buck converters. Aside from being controlled as boost-based ones, these converters exhibit nonpulsating currents at both input and output ports. Finally, for all the previous cases, the surface selection in terms of V o can be implemented analogically using operational

SB (X) : iL = Iref (Pref , Vg , Vo ) = SC (X) : iL = Iref (Pref , Vg , Vo ) =

Pref Vg Pref Vg Pref Vg

SD (X) : iL = Iref (Pref , Vg , Vo ) = 0

1 1 1 · = . 1 − DM 1 1 − DM 2 (1 − D)2

(9)

Nevertheless, according to the literature, a single-switch quadratic-boost converter, exhibiting the same dynamics as the previous one, can be found in [41] and [42]. However, the two-switch topology has been finally adopted in the design since the switch M 1 must be a low-voltage highcurrent device, whereas M 2 must be rated for higher voltages and lower currents. In contrast, the device in the single-switch quadratic boost must carry M 1 + M 2 currents but must be rated with the same breakdown voltage as M 2. This switch will therefore operate under high stress conditions, and a commercial device properly rated will have a larger on-resistance. On the other hand, quadratic-boost element sizing results from a tradeoff between diverse constraints: fast dynamics plant, converter variable ripple, and loss considerations. The switching frequency, although variable, is in the range of 80–120 kHz. The following considerations have been adopted

0

SA (X) : iL = Iref (Pref , Vg , Vo ) =

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·




· 1−

    < VO ≤ VBC 

0 < VO ≤ VAB

VO VAB VO −VBC VCD −VBC



VAB

VBC < VO ≤ VCD     VO > VCD

(5)

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TABLE II Q UADRATIC -B OOST E LEMENT VALUES

TABLE III Q UADRATIC -B OOST S LIDING -M ODE E XISTENCE C ONDITIONS

in the design: As the fuel-cell voltage is very small, the value of L1 has to be quite low to process a high current level with an acceptable core size. However, since low values of L1 entail high input current ripples, a capacitor C0 is parallel connected to reduce the current ripple to an adequate value for the fuel cell. Furthermore, to avoid decreasing the inverter MPPT tracker speed, the output capacitor C2 , which is parallel connected to the MPPT inverter input capacitor, is set to a low value. In addition, to avoid a high-ripple SPD output current that can also perturb the MPPT inverter, L2 is set to a high value. As a result of the preceding tradeoffs, the values of the quadraticboost elements are designed and given in Table II. B. Panel Shaper Analysis In the circuit of Fig. 8, the model of the MPPT inverter has been simplified and includes only an input conductance Gop in parallel with the capacitance C. This conductance should follow the internal MPPT system, maximizing the power absorbed by the inverter. No 100-Hz ripple effects have been included in the model, as the inverter control loop eliminates them. A good example of these techniques can be found in [17]. The circuit shown in Fig. 8 is a variable-structure system [37]–[39] with two electrical topologies, referred to as ON and OFF. The system dynamics can be described in general by the following linear state equations: ∂X(t) = AON X(t) + BON ∂t ∂X(t) = AOFF X(t) + BOFF . ∂t These equations can be compacted as shown in ∂X = AOFF X + BOFF ∂t + [(AON − AOFF )X + (BON − BOFF )] · u(t)

(10) (11)

where n is the diode ideality factor, VT is the thermal potential (26 mV at 300 K), and N S = 288 is the number of solar cells (four panels with 72 cells for each one). An equivalent capacitor CEQ = C2 //C replaces C2 and C. As explained in Section V, the panel shaper consists of four sliding-mode control laws (5), one for each shape region, which can be analyzed using a generic surface given in SK = iL − Iref

where u(t) stands for the control signal u(t) which takes the values u(t) = 1 and u(t) = 0 during the ON and OFF states, respectively. Particularizing (12) to the circuit of Fig. 8 and neglecting resistive losses lead to a dynamical model (13) of the system  Vg ∂iL1 vC1  ∂t = L1 − L1 [1 − u(t)]   ∂iL2 vC1 VO   = − [1 − u(t)]  ∂t L2 L2  ∂vC1 −iL2 iL1 = + [1 − u(t)] ∂t C1 C1 

 (13)
 ∂VO VO 1 − Gop VO   ∂t = CEQ ISC − Id exp nNS VT    iL2 + CEQ [1 − u(t)]

= 0,

K = A, B, C, D. (14)

Following the sliding-mode control design guidelines given in [37]–[39], the first step is devoted to find the existence conditions of a sliding motion over the previous surfaces. This sliding motion is ensured if SK ·

(12)

K (Pref , Vg , V0 )

dSK < 0 K = A, B, C, D (15) dt  dIref o dIref o dIref o dSK ˙ = iL − + + P V V dt dPref ref dVg g dV0 0 K = A, B, C, D

(16)

where the upper dot stands for the time derivative. From (13)–(16), the existence conditions can be found and are shown in Table III for the nontrivial surfaces K = A, B, C, since for K = D, the surface is SD (X) = IL = 0. It is worth noting that these conditions give the maximum derivatives of Pref , Vg , and V0 to assure a sliding motion. Moreover, it can be noted that the panel characteristic is not directly involved in these conditions, i.e., there is no influence of the time derivative dISC /dt in the existence of the sliding-mode regime. However, referring to Fig. 8, any transient

VALDERRAMA-BLAVI et al.: POWER ADAPTOR DEVICE FOR DOMESTIC DC MICROGRIDS

unbalance between the panel current and the current absorbed by the inverter GOP · VO will affect the voltage derivative of the capacitor CEQ = C//C2 . Assuming, for instance, a step irradiance change of ∆ISC , then ∆ISC = CEQ

dV0 . dt

(17)

Therefore, the maximum irradiance step ensuring the sliding motion will be given by

 dV0 ∆ISCM AX = CEQ (18) dt M AX where (dV0 /dt)MAX is given by the boundaries of Table III. For instance, in the case of the surface SA of Table III, these irradiance step boundaries are given by   Vg · V0 − Vg VAB · Vg CEQ · < ∆ISCM AX − L · Pref CEQ · VAB · Vg2 < . (19) L · Pref The next steps are devoted to find the equivalent control ueq [37]–[39], the ideal sliding dynamics, and the equilibrium point and, finally, to investigate the system stability. After the generic analysis, each result must be particularized for each shape region. To reduce the extension of this work, only some results are given here. For instance, the equivalent control for all regions is given by (20) and (21), and the expressions of the equilibrium point at region B are shown in (22) and (23). This equilibrium point is interesting because region B is the region where the adaptor will work, in all canonical modes, at steady-state conditions   V   O Pref VO Gop VO −ISC +Id exp nNS VT Vg  +  L1 VAB Vg CEQ ueq_A = 1 − Pref ·iL2 vC1 (20) L1 + VAB Vg CEQ   Vg ueq_B = 1 − vC1 , ueq_D = 0   V Pref VO Gop VO −ISC +Id exp nN OV Vg S T L − (VBC −VCD )Vg CEQ ueq_C = 1 − 1 . (21) Pref ·iL2 vC1 − L1 (VBC −VCD )Vg CEQ The expression of V o∗ in (23) has been deduced using a linear approximation for the solar panel exponential curve, given in (24). Consequently, the expression of V o∗ is exact for the first canonical mode VOF M , where there is no contribution from the solar panel, and approximated for the second operational mode VOSM ∗ ∗ XB = [i∗L1 , i∗L2 , vC1 , VO∗ ]T  ∗ vC1 = Vg VO∗ i∗L2 = √Pref ∗ V V g O

i∗L1

=

Pref Vg

VO∗ SM

Ip = ISC −Id exp

≈

ISC +

VO∗ FM 

 =

Pref Gop

I

2 +4P d ISC ref Gop + nN V

  I 2· Gop + nN dV S T  VO∗ Id VO∗ . ≈ ISC − nNS VT nNS VT

S T



   

(22)

  

(23)

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equilibrium point. Moreover, two dynamic components can be distinguished, a fast oscillation (ωA ) around the MPP due to the tracking algorithm and the slow changes of the MPP itself (ωMP ) due to weather changes. As a result, the inverter input conductance can be expressed as Gop (t) = GM + ∆gA · sin(ωA t) + ∆gM P · sin(ωM P t). (25) On the other hand, to prevent instabilities, the converter time constants must be chosen to avoid resonances at these frequencies. The system poles for region B are given in (26), and the low frequency or dc gain of the V o(s)/∆g(s) transfer function is shown in (27)  D(s) = C1 s3 +αs2 +βGM s+ L22γ  C EQ 
 ∗  VO Id C1  α = Vg GM + CEQ GM + nNS VT 

 ∗ (26) V V Id g 1 O 1  + CEQ β = GM2L2 2+ CCEQ ∗ VO Vg GM + nNS VT  
    1 γ = GM + CCEQ GM + nNISdVT  VO (s)  −VO∗ = .  ∆g(s) s=0 2GM + nNIdV S

(27)

T

Note that the dc gain in (27) is negative and independent from the adaptor energy storage elements L1 , L2 , C1 , and C2 . Thus, an increment of the inverter conductance reduces the inverter input voltage. It can be finally pointed out that expressions (26) and (27) are related to the second canonical mode where the same panel linearization given in (24) has been used. The expressions for the first operation mode can be obtained eliminating the term Id /nNS VT in (25)–(27). VII. E XPERIMENTAL R ESULTS We have developed a renewable energy system as presented here. The solar field and the test workbench are shown in the photographs of Fig. 9(a) and (d). The workbench includes PV solar panels, various MPPT inverters, fuel cells, a battery bank, and two wind turbines. A preliminary experiment consisted in testing the design of an SPD and a panel shaper with shape voltages of VAB = 100 V, VBC = 150 V, and VCD = 170 V, which was carried out according to the design example of Section VI. The test results shown in Fig. 9(b), where Ch1 (horizontal axis) is V0 and Ch2 (vertical axis) is the current reference and confirms the proper operation of the panel shaper. In the remaining experiments, diverse sources were added to the domestic grid PV system defined in Section VI, namely: a solar field of four series-connected panels NT-170U1 from SHARP totalizing 680 Wp and an open circuit voltage of V oc = 170 V. The MPPT inverter is a Steca model of 2 kW.

(24)

To investigate the system stability, we assumed that the MPPT algorithm adjusts continuously the inverter input conductance Gop (t) and the SPD adaptor never reaches a true

A. Adding a Low-Voltage Fuel-Cell Stack The first element to be adapted was a seven-cell proton exchange membrane (PEM) fuel-cell stack from H2-Economy.

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Fig. 9. (a) Solar field facilities. (b) Panel shaper operation. (c) Adding a low-voltage fuel cell. (d) Laboratory test bed facilities. (e) 1.2-kW fuel-cell extraction transient. (f) 1-kW three-phase wind turbine. (g) Battery adaptor (charger–discharger). (h) Battery charge/discharge transient. (i) Day–night–day transient.

It is rated at 100 W, 4 V–25 A. This is the case studied in the previous section. In Fig. 9(c), the 4.2-V source was suddenly connected to the system. As expected, after a short transient, the PV array voltage V o was restored, the MPP was recovered, and the inverter input power was increased by 100 W. B. Extracting a 1.2-kW Nexa Fuel-Cell Stack This experiment was devoted to test a 1.2-kW Ballard Nexa PEM fuel-cell stack. In this case, a boost converter is enough to match the stack voltage (27 to 43 V) and the inverter voltage V o. The fuel-cell stack and the solar panels were delivering energy to the inverter, and suddenly, the fuel-cell stack was shut off. During the transient, the inverter had to accommodate a sudden input power reduction from 1.6 to 0.6 kW. Fig. 9(e) shows the fuel-cell voltage V g (Ch1), the fuel-cell current Ig (Ch2), the inverter input voltage V 0 (Ch3), and the inverter input current (Ch4). As seen, before the transient, the power delivered by the source was 1020 W. Four seconds after the disconnection, the voltage of the PV array was near the same

as before the shutoff, and the solar field MPP was recovered and held. Nevertheless, as the inverter control bandwidth is generally designed to track slow and small solar irradiation changes, it cannot respond to this sudden changes. Focusing on the transient of Fig. 9(e), when the input power is drastically reduced due to the fuel-cell shutoff, the inverter could not reduce sufficiently fast the energy injected into the grid, and the inverter input capacitor was instantaneously depleted, dropping its voltage under the inverter operation threshold at 80 V. As a result, the inverter stopped. Subsequently, the inverter input voltage V o rose again until the open circuit voltage of the solar field was reached (170 V). At the end of the transient, the MPPT began to increase the inverter input current searching a new MPP, which was finally restored to its initial value. This temporary loss of the inverter MPP will happen for any source extraction if the power of this source is high enough compared with the power delivered by other sources connected to the system. This transient where the MPP is lost and even the inverter stops can be eliminated using a soft-stop circuit.

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TABLE IV S LIDING -M ODE E XISTENCE C ONDITIONS OF B UCK AND B OOST C ONVERTERS

C. 1-kW Three-Phase Wind Turbine A 1-kW alternator from a Whisper200 wind generator was mechanically coupled with a 4-kW induction machine supplied with a programmable inverter. That generator, a three-phase ten-pole permanent magnet alternator, was connected to a threephased boost adaptor with power factor correction. Fig. 9(f) shows the rectified generator phase voltage (Ch1), the sum of the rectified input currents of the single phase adaptors (Ch2), the microgrid bus voltage V o (Ch3), and the current delivered to the inverter (Ch4). As it can be seen, the generator was rotating at 55.5 Hz and feeding the system with 390 W. D. Battery Charger/Discharger A battery charger/discharger prototype shown in Fig. 9(g) was implemented with a bidirectional boost converter and tested with the PV grid-connected system. Fig. 9(h) shows several transients of the battery charger. At the beginning, the battery charger current (Ibat , Ch4) was sinking 1 A from the solar field, reducing the inverter input current (Ch2) by the same amount. This experiment proves the feasibility of sinking power, as considered by the third canonical mode. Next, the battery adaptor was not working, and finally, the battery converter was sourcing 1 A to the system, increasing the inverter input current. In this discharge mode, the SPD was working according to the second canonical mode. E. Day–Night–Day Transient The previous experiments have verified on the one hand the MPP recovery after the connection and disconnection transients. On the other hand, these experiments have also confirmed the proper operation of the SPD in the second and third canonical modes. The last experiment was devoted to verify the SPD operation in the first canonical mode and emulated the operation of the PV system with an additional

source during a day–night transient. The results are shown in Fig. 9(i) where (Ch1) is the inverter input voltage V o, (Ch2) is the inverter input current Iinv, (Ch3) is the additional source voltage V g, and, finally, (Ch4) is the current delivered by this source In. During the day, the inverter input received the energy coming from the PV panels and from a 360-W source connected through a boost SPD operating in the second canonical mode. During the night, the solar power was no longer present, and the SPD was operating according to the first canonical mode, feeding the inverter with the expected 360 W. It can be pointed out that, during the night, the inverter MPP was located around 120 V, but during the day, the MPP was around 130 V. In addition, as the power drop from day to night was smaller and slower than the transient shown in Fig. 9(e), the inverter did not stop, thus avoiding the restart procedure. VIII. C ONCLUSION The notion of a versatile bidirectional converter whose output port behaves as a power source or as a power sink depending respectively on the mode of operation, i.e., delivering energy to the grid or absorbing this one from the ac mains, has been presented under the name SPD. The novelty of the proposed device with respect to other bidirectional converters employed in the interface of different energy sources with a dc bus constituting the input of a centralized inverter in many grid-connected electric architectures of renewable energies is the fact that the sliding-mode control employed in the feedback loop of the converter forces the power stage to behave as a programmable power source which emulates the voltage-to-power (P/V0 ) characteristic of a PV array. This fact is the most relevant feature of the power adaptor in a clear cut contrast with the behavior of existing devices whose output corresponds either to a voltage source or to a current source. While the connection of current or voltage sources to a dc bus has to employ in each case a topologically compatible electrical architecture, the

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connection of a power source has not this type of constraints, and therefore, it offers much more flexibility, the resulting calculations being directly in terms of power and efficiency. As it has been shown in this paper, the SPD allows the hot-swap connection of additional sources, loads, and storage elements in domestic grid-tie PV systems including commercial MPPT inverters without perturbing the PV inverter operation and preserving the MPP of the solar array. Moreover, the power of each additional element can be independently controlled. As it has been suggested in Section III, the SPD can be used as a building block to implement different microgrid architectures, such as a multiple-SPD grid-tie derived dc microgrid, stand-alone derived dc microgrid with one SPD, and singleSPD grid-tie derived dc microgrid. Moreover, since the SPD allows the inclusion of an energy reservoir, it can potentially contribute in existing grid-tie PV systems to several active grid cooperation policies such as smoothing renewable random production, injecting power into the grid according to some arbitrary profiles, and even cooperating with grid regulation tasks. Finally, using an energy reservoir also allows an enhanced use of the inverter by increasing the plant yield. After detailing the sliding-mode control design steps for a quadratic-boost SPD converter, several SPD have been designed to experimentally verify on a 2-kW domestic grid-tie PV system the feasibility of the proposed approach. With this aim, this paper has included a set of experimental results testing the panel shaper, the system operation under day–night transient, and the connection of additional elements such as lowvoltage 1.2-kW fuel cells, a 1-kW wind turbine, and a battery charger–discharger. All these tests have shown that the SPD properly operates as a power source/sink without perturbing the MPP of the solar field.

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Hugo Valderrama-Blavi (S’98–M’01) received the Ingeniero and Ph.D. degrees from the Universitat Politècnica de Catalunya, Barcelona, Spain, in 1995 and 2001, respectively. He is currently an Associate Professor in the Departament d’Enginyeria Electrònica, Elèctrica i Automàtica, Universitat Rovira i Virgili, Tarragona, Spain. During the academic year 2001–2002, he was a Visiting Scholar at Laboratoire d’Automatique et Analyse des Systémes-Centre National de la Recherche Scientifique, Toulouse, France. His current research interests are power electronics, renewable energies, silicon carbide devices, and nonlinear control.

Josep M. Bosque (S’10) received the Enginyer Tècnic Industrial en Electrònica Industrial degree and the Enginyer en Electrònica master degree from the Universitat Rovira i Virgili (URV), Tarragona, Spain, in 2005 and 2009, respectively, where he is currently working toward the Ph.D. degree. Since 2004, he has been a Research Technician with the Automatics and Industrial Electronics Group, URV. His research interests are power electronics and renewable energy.

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Francisco Guinjoan (M’92) received the Ingeniero de Telecomunicación and Doctor Ingeniero de Telecomunicación degrees from the Universitat Politecnica de Catalunya (UPC), Barcelona, Spain, in 1984 and 1990, respectively, and the Docteur es Sciences degree from the Université Paul Sabatier, Toulouse, France, in 1992. He is currently an Associate Professor in the Departamento de Ingenieria Electrónica, Escuela Técnica Superior de Ingenieros de Telecomunicación Barcelona, UPC, where he teaches courses on power electronics. He has coauthored more than 80 papers in international journals and conference proceedings, and in 2011, he was a Guest Coeditor of the Special Issue on “Smart Devices for Renewable Energy Systems” of the IEEE T RANSACTIONS ON I NDUSTRIAL E LECTRONICS. His research interests include power electronics modeling and control for renewable energy systems.

Luis Marroyo (M’04) received the M.Sc. degree in electrical engineering from the University of Toulouse, Toulouse, France, in 1993, and the Ph.D. degree in electrical engineering from the Universidad Pública de Navarra (UPNA), Pamplona, Spain, in 1997, and from the Laboratoire d’Electronique et Electrotechnique IndustrielleEcole Nationale Supérieure d’Electrotechnique, d’Electronique, d’Informatique, d’Hydraulique, et des Télécommunications Institut National Polytechnique Toulouse, Toulouse, France, in 1999. From 1993 to 1998, he was an Assistant Professor in the Department of Electrical and Electronic Engineering, UPNA, where he has been an Associate Professor since 1998. He is the Head of the Ingeniería Eléctrica, Electrónica de Potencia y Energías Renovables research group. He has been involved in more than 60 research projects mainly in cooperation with industry, is the coinventor of 11 international patents, and has coauthored more than 70 papers in international journals and conference proceedings. His research interests include power electronics, grid quality, and renewable energy.

Luis Martínez-Salamero (M’85) received the Ingeniero de Telecomunicación and Ph.D. degrees from the Universidad Politécnica de Cataluña, Barcelona, Spain, in 1978 and 1984, respectively. From 1978 to 1992, he taught courses on circuit theory, analog electronics, and power processing at the Escuela Técnica Superior de Ingenieros de Telecomunicación de Barcelona, Barcelona. During the academic year 1992–1993, he was a Visiting Professor at the Center for Solid State Power Conditioning and Control, Department of Electrical Engineering, Duke University, Durham, NC. He is currently a Full Professor in the Departamento de Ingeniería Electrónica, Eléctrica y Automática, Escuela Técnica Superior de Ingeniería, Universidad Rovira i Virgili, Tarragona, Spain. During the academic years 2003–2004 and 2010–2011, he was a Visiting Scholar at LAAS-CNRS, Toulouse, France. Dr. Martínez-Salamero was recognized as a Distinguished Lecturer of the IEEE Circuits and Systems Society for the period 2001–2002 and was the President of the IEEE Spanish Joint Chapter of the IEEE Power Electronics and IEEE Industrial Electronics Societies for the period 2005–2008.