Predicting Field Coupling to an IC Using Measured ... - IEEE Xplore

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 56, NO. 6, DECEMBER 2014

1287

Predicting Field Coupling to an IC Using Measured Coupling Factors Ji Zhang, Xiang Li, Richard Moseley, Member, IEEE, David Pommerenke, Senior Member, IEEE, and Daryl G. Beetner, Senior Member, IEEE

Abstract—High-strength electric and magnetic fields can capacitively or inductively couple energy to integrated circuits (ICs) and cause them to fail. While measurements can show when an IC will fail, they do not provide insight into the mechanisms for failure. Modeling the response of the IC to these fields is challenging, in part because of the small features of the IC and the large amount of circuitry information that must be included from the IC and printed circuit board. The goal of the following work is to develop a methodology for predicting the voltage or current on the pins of the IC from incident electric or magnetic fields. The method is based on measuring “coupling factors,” which show the relationship between a specific field component and the IC response. These coupling factors can be determined by placing the IC in a known electric or magnetic field within a transverse electromagnetic cell and measuring the response. The developed technique was validated by predicting the response of a commercially available 8-bit microcontroller to the electromagnetic fields generated by the nearby discharge of an electrostatic discharge gun. The proposed approach allows the prediction of the waveforms and a better understanding of failure mechanisms without the need to know or model IC geometry and circuitry. Index Terms—Electromagnetic coupling, electrostatic discharge (ESD), IC immunity, modeling, prediction.

Fig. 1.

Electromagnetic fields coupled to an IC during an ESD event.

Fig. 2.

Field coupling test using an ESD generator.

I. INTRODUCTION NTEGRATED circuits (ICs) are susceptible to errors caused by electrostatic discharge (ESD) events, electrical fast transients, or pulsed or continuous RF signals [1]. While most electromagnetic energy is conductively coupled to the IC through traces on the printed circuit board (PCB), it may also be capacitively or inductively coupled directly to the IC package or die. Measurements are often used to determine the level of electric or magnetic fields required to cause the IC to fail, but these measurements are often very time consuming, and more importantly, may not provide a deeper understanding of the mechanisms that caused the error. The goal of the following work is to provide a

I

Manuscript received August 23, 2013; revised March 18, 2014; accepted August 25, 2014. Date of publication September 19, 2014; date of current version December 11, 2014. This work was supported in part by the National Science Foundation under Award 0855878. J. Zhang is with the Electrical and Computer Engineering, Missouri University of Science and Technology, Rolla, MO 65401 USA, and also with Cisco Systems Inc., San Jose, CA 95134 USA (e-mail: [email protected]). X. Li and R. Moseley are with the Freescale Semiconductor, Inc., Austin, TX 78735 USA (e-mail: [email protected]; [email protected]). D. Pommerenke is with the Electromagnetic Compatibility Laboratory, Missouri University of Science and Technology, Rolla, MO 65401 USA (e-mail: [email protected]). D. Beetner is with the Electrical and Computer Engineering, Missouri University of Science and Technology, Rolla, MO 65401 USA (e-mail: daryl@ mst.edu). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TEMC.2014.2355719

method of predicting the voltage or current waveforms on pins of an IC in the presence of incident electric or magnetic fields. Predicting the waveforms from the fields allows one to better predict when the IC might fail and to obtain a better understanding of the failure mechanisms. A typical scenario, where energy is coupled to an IC through electromagnetic fields is illustrated in Fig. 1. An ESD event occurs a short distance from an IC mounted on a PCB. Electric and magnetic fields generated by the ESD event couple to the IC, possibly causing a disruption of the IC. In laboratory tests, the ESD events are usually created by an ESD generator (gun) discharging to the return plane as illustrated in Fig. 2 [2]. Unlike tests for conducted immunity, where signals are applied directly to traces connected to the IC, the signals coupled to individual pins are difficult to predict and depend on parameters like the package geometry, the IC circuitry, the relative location and orientation of the source and the IC, the source waveform, and many others. Typical measurement protocols look only for a failure of the IC, and not the waveform on individual pins. While these waveforms could be measured, doing so on many pins is difficult and time consuming. The difficultly is compounded by the fact that thorough testing requires the test be repeated for many positions and orientations of the IC and source. One approach for predicting the currents and voltages in the IC is to use a full-wave model of the ESD gun and IC package

0018-9375 © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

1288


[3], [4]. Full-wave modeling of the IC package requires the use of a fine mesh, while the relatively large size of the ESD gun requires a large computation volume. Consequently, the simultaneous presence of the large ESD gun and small-scale IC package results in a large number of mesh elements and very long simulation times. In addition, accurate full-wave simulations require accurate models of the circuit elements inside the die [5], [6]. These circuit elements may not be known, and if known, may not be easily incorporated into the full-wave model. Another method of predicting the current and voltage waveforms is to use a SPICE circuit model of the IC, where field coupling is represented within the SPICE model [7]. The coupled electrical and magnetic energy is represented with equivalent current and voltage sources. A detailed SPICE model of the IC package and die is required [8]. This method is much more computationally efficient than pure full-wave simulations and allows rapid simulation of many package and source configurations. Detailed information about the die and package, however, may not always be easy to obtain. The following paper presents a method for predicting the voltage or currents on the pins of an IC from incident electromagnetic field using measured coupling factors. The coupling factors represent the frequency-dependent transfer function from the incident field to the voltage or current at the pin. Coupling factors are found using a transverse electromagnetic (TEM) cell and a vector network analyzer. Coupling factors are found for the tangential magnetic fields, Hx and Hy , and for the vertical electrical field, Ez . The voltage or current at a pin can be found from the superposition of these three components. The advantages of this method are that no information is required about the internal characteristics of the IC or connected support circuitry and that simulations may be performed very quickly for a wide variety of incident fields. Information about the incident fields can be obtained through full-wave simulations or measurements. The proposed method is described in detail in the following sections. The theoretical basis behind the coupling factors and methods for determining their values are described first, followed by an experimental validation of the approach using a commercial microcontroller. The predicted voltage waveform on the power pin of the microcontroller during an ESD test is shown to closely match the measured response. II. COUPLING FACTORS A. Definition To simplify prediction, the mechanisms resulting in EM field coupling are treated as a “black box” whose transfer characteristics can be determined through experiments. The transfer function can be determined by applying known EM fields to the IC as input variables and measuring the IC’s response. The transfer functions between the fields and measured response are the “coupling factors.” The voltage and current coupled to an IC is a superposition of contributions from multiple field components. In most scenarios, the IC is mounted very close to a ground-plane as shown in Fig. 2. In this case, the tangential electric field and vertical magnetic field incident on the IC are small and typically

Fig. 3. Coupling factors are defined for the vertical electric field EZ and the tangential magnetic field components H x and H y .

can be ignored. The primary coupling comes from the vertical electric field component Ez , and the tangential magnetic field components Hx and Hy . The coupling factors are consequently defined according to these three field components, as illustrated in Fig. 3. The coupling factor for each field component is defined as the ratio of the voltage (or current) at the measurement point (Vm ea ) resulting from the incident field to the strength of the incident field component at the IC location Vm ea E z (f ) Ez (f )

(1)

CFH x =

Vm ea H x (f ) Hx (f )

(2)

Δ

Vm ea H y (f )

Δ

CFE z = Δ

CFH y =

Hy (f )

(3)

where Vm ea E z is the measured voltage resulting from the incident vertical electric field, and Vm ea H x and Vm ea H y are the measured voltages resulting from incident tangential magnetic fields in the x- and y-directions, respectively. The total coupled voltage is a superposition of the contributions from each field component Vinduced = CFE z · Ez + CFH x · Hx + CFH y · Hy .

(4)

B. Measurement To find the coupling factors, the IC must be excited with wellknown electric and magnetic fields and the response to these fields must be measured. Here, a TEM cell, a hybrid coupler, and a vector network analyzer (VNA) are used to perform the measurement, as illustrated in Figs. 4 and 5. The IC under test is mounted on a test board and placed in the TEM cell above the septum. Within the TEM cell, the IC is exposed to an incident TEM wave generated by the VNA. The propagating wave ideally contains three field-components: Ez , Hx , and Hy . The values of these components can be easily calculated from the VNA stimulation voltage [9], [10]. The resulting voltage at the IC pin, Vinduced , is measured outside the TEM cell using the VNA. To separate the response of the IC to the electric field from the response to the magnetic fields, the TEM cell is fed two signals that are either in-phase or out-of-phase with one another. The signal from the VNA is split into two channels by a 180° hybrid coupler, as shown in Figs. 4 and 5. The

ZHANG et al.: PREDICTING FIELD COUPLING TO AN IC USING MEASURED COUPLING FACTORS

Fig. 4.

1289

Experimental setup for extracting the coupling factors (top view).

Fig. 6. Transfer function from input to outputs of the hybrid coupler for the case where outputs are 180° out-of-phase.

Fig. 5:

Experimental setup for extracting the coupling factors (cutaway view).

hybrid generates two signals that are either in-phase or 180° out-of-phase with one another. Changing the phase allows either the electric field or the magnetic field to dominate at the IC location, as will be explained later. The two cables from the hybrid to the TEM cell should be the same length, so the signals in each cable arrive at the TEM cell at the same time. The receiving port of the network analyzer is connected to the IC pin under test to measure the response voltage. Two 225Ω 0402 SMT resistors are inserted in series between the IC pin and the coaxial probe so that a 10:1 probing system is constructed to avoid changing the voltage/current at the pins. The characteristics of the hybrid coupler must be measured to determine the coupling factors. The hybrid coupler has four ports: two input ports and two output ports. Depending on which input is selected, the phase difference between the outputs will either be 0° or 180°. The S-parameter transfer functions between the inputs of the hybrid coupler and the outputs are illustrated in Fig. 6 for the case where the phase difference between the outputs is 180°. The transfer functions for this case will be denoted as H180 a and H180 b . For the other case where the two outputs are inphase (separated by 0°), the transfer functions from input to output port “a” and “b” will be denoted as H0 a and H0 b . These measured transfer functions for the hybrid coupler are used in future equations so that no additional deembedding is required.

Fig. 7.

EM field distribution within the TEM cell.

Driving the TEM cell inputs with signals that are in-phase or 180° out-of-phase can enhance or cancel electric or magnetic fields at the center of the cell, as illustrated in Fig. 7 [9]. When the input on the left-hand side of the TEM cell is of the same phase as the input on the right-hand side of the TEM cell, the electric fields generated by the two signals are in-phase, but the magnetic field is of opposite phase (since the currents are traveling in the opposite direction). Ideally, the electric field at the center of the TEM cell is twice the electric field produced by a single input and the magnetic field at the center of the TEM cell is zero. More precisely, the vertical electric field, Ez (f ), and tangential magnetic field, Ht (f ), at the center of the TEM cell (at the IC location) are given by Ez (f ) =

V0,a (f ) V0,b (f ) Vin (f ) · H0 a (f ) + = dTEM dTEM dTEM +

Vin · H0 b (f ) dTEM

(5)

1290


V0,b (f ) Vin (f ) · H0 a (f ) V0,a (f ) − = dTEM · η dTEM · η dTEM · η

Ht (f ) =

−

Vin · H0 b (f ) dTEM · η

(6)

where V0,a and V0,b are the in-phase input voltages at inputs a and b of the TEM cell (corresponding to the hybrid output ports a and b), Vin is the signal at the input of the hybrid coupler, which is equivalent to the signal at the output of the VNA, dTEM is the distance from the septum to the TEM cell wall, and η is the wave impedance (120 π in air). When the TEM cell inputs are 180° out-of-phase, the magnetic fields generated by the two signals are in-phase, but the electric fields are of opposite phase. The fields at the center of the TEM cell (at the IC location) are given by

1) In-phase inputs (primarily Ez ): H0 a (f ) H0 b (f ) + + CFH y (f ) S21E z (f ) = CFE z (f ) dTEM dTEM H0 a (f ) H0 b (f ) − × + CFH x (f ) · 0. dTEM · η dTEM · η (9) 2) 180° out-of-phase outputs and 0° rotation of the IC (primarily Hy ): H180 a (f ) H180 b (f ) S21H y (f ) = CFE z (f ) + dTEM dTEM H180 a (f ) H180 b (f ) − + CFH y (f ) dTEM · η dTEM · η + CFH x (f ) · 0.

Ez (f ) =

V180,a (f ) V180,b (f ) Vin (f ) · H180 a (f ) + = dTEM dTEM dTEM +

Ht (f ) =

Vin (f ) · H180 b (f ) dTEM

(7)

3) 180° out-of-phase outputs and 90° rotation of the IC (primarily HX ): H180 a (f ) H180 b (f ) S21H x (f ) = CFE z (f ) + dTEM dTEM

Vin (f ) · H180 a (f ) V180,a (f ) V180,b (f ) − = dTEM · η dTEM · η dTEM · η −

Vin (f ) · H180 b (f ) dTEM · η

+ CFH y (f ) · 0 + CFH x (f ) H180 a (f ) H180 b (f ) − × . dTEM · η dTEM · η

(8)

where V180,a and V180,b are the 180° out-of-phase inputs to the TEM cell. Applying inputs to the TEM cell that are in-phase exposes the IC to a strong vertical electric field and allows one to determine the coupling factor for the Ez field. Applying inputs that are out-of-phase generates a strong tangential magnetic field and, by rotating the IC, allows one to determine the coupling factors for the Hx and Hy fields. So long as the dimension of the IC (e.g., 1.5 cm × 1.5 cm) are small compared to the wavelength of energy from a typical ESD event (e.g., no more than a few gigahertz), it is reasonable to assume that the electromagnetic fields are uniformly distributed over the IC, both during TEM and ESD gun tests. Furthermore, since the IC is located at the center of the TEM cell (on one wall), the relative phase of the wavefronts propagating from the two TEM ports to the IC are only determined by the hybrid coupler’s response (as shown in Fig. 6). C. Calculation of Coupling Factors Values of S21 from the hybrid coupler input to the IC pin voltage (or current) can be expressed in terms of coupling factors. The relationship is found from the expression for the electric and magnetic fields in the TEM cell [(5)–(8)], from the expression for inducted voltage in (4), and using the fact that S21 = Vinduced /Vin . The relationship varies depending on the configuration of the hybrid coupler and the orientation of the IC. These relationships are given as follows.

(10)

(11)

In matrix form, these relationships are given as ⎤ ⎡ S21E z ⎥ ⎢ ⎢ S21H y ⎥ = ⎦ ⎣ S21H x ⎡

H0 a + H0 b ⎢ dTEM ⎢ ⎢ H180 + H180 a b ⎢ ⎢ dTEM ⎢ ⎢ ⎣ H180 a + H180 b dTEM ⎡ ⎤ CFE z ⎢ ⎥ CFH y ⎥ ×⎢ ⎣ ⎦ CFH x

H0 a − H 0 b dTEM · η H180 a − H180 b dTEM · η 0

⎤ 0 0 H180 a − H180 b dTEM · η

⎥ ⎥ ⎥ ⎥ ⎥ ⎥ ⎥ ⎦

(12)

where values of S21, of the coupling factors, and of hybrid coupler transfer functions are dependent on frequency. The coupling factors can be found from measurements of S21 by solving this set of linear equations. Once the coupling factors have been identified, the IC’s response to any incident field can be found using (4) when the incident fields are known. III. VALIDATION Application of the coupling factors was demonstrated by using them to predict the voltages at pins of a microcontroller during an ESD test, where an ESD gun was discharged a short distance away from the IC.


Fig. 8.

1291

ESD test setup. Fig. 10. Experimentally determined coupling factors when all pins were connected.

through a 10:1 voltage divider (consisting of a 450-Ω SMT resistor and the 50-Ω input impedance of the instrument). All circuitry besides the IC under test (including the power supply and GPIO control) were located on the bottom of the PCB as shown in Fig. 9 so that the measurement circuitry could be isolated from the ESD event. Three different configurations were tested: all power/return pins connected to the power/return rails on the PCB; only the VDD pin connected to the power rail while the other power/return pins were unconnected (floating), and only the VDD and VSS1 pin connected to power/return net while the other pins were unconnected. The I/O pins were left floating in each case. Values for electric and magnetic fields were found using a full-wave simulation model of the ESD gun [3]. B. Predicted Results

Fig. 9. IC on test board: (a) top and bottom view of PCB (b) application/measurement circuit.

A. Test Setup The ESD gun was discharged to the return plane at a point 4 cm from the IC, as shown in Fig. 8. Tests were performed with the ESD gun set to a 1-kV charge. Only the IC was exposed to the EM field; all measurement equipment and cables were shielded from the ESD event. The voltage on the VDD pin (see Fig. 9) was measured during the ESD event. An oscilloscope was connected to the VDD pin through a 2.2-nF coupling capacitor. The voltage was scaled

Fig. 10 shows the coupling factors for each field component when all five power and return pins were connected. The Ez , Hy , and Hx coupling factor are marked in red, blue, and green, respectively. Not surprisingly, the coupling factors all rise at the rate of 20 dB/decade, corresponding to the increasing capacitive or inductive coupling. This rising trend stops around a few gigahertz, where the energy begins to decrease. The rising trend is broken in part by the 1-GHz bandwidth limitation of the TEM cell. The trend is also impacted by the self-inductance and capacitance of the IC package, which will attenuate the portion of the coupled energy that reaches the measurement port at high frequencies. The induced voltage on the VDD pin can be predicted using (4) from the calculated coupling factors and the simulated EM fields from the ESD gun. The predicted and measured results are shown in Fig. 11 in the frequency domain and in Fig. 12 in the time domain. The major features of the measured voltage are generally predicted well in both the time and frequency domains, though the agreement between simulation and measurement is poor below 100 MHz. There are two possible reasons for this poor agreement. First, the inductive and capacitive coupling is small at low frequency so that the S21 data used to find the coupling factors are less accurate at these frequencies due to noise. Second, the

1292


Fig. 13.

Measured coupling factors when only VDD/VSS1 were connected.

Fig. 11. Spectral density of measured and predicted voltage on the Vdd pin (all pins connected).

Fig. 14. Spectral density of measured and predicted voltage on the Vdd pin (only Vdd/Vss1 pins connected).

Fig. 12. Time-domain waveform of measured and predicted voltage on the Vdd pin (all pins connected).

predicted voltage is only as good as the fields predicted by the ESD gun model. The accuracy of the ESD gun model is also not good below 100 MHz [3]. Both measured and simulated voltages drop rapidly beyond 1 GHz due to the rapidly decreasing EM field from the ESD gun. Coupling factors and measured and predicted voltage on the Vdd pin were similarly found for the case where only the VDD and VSS1 pins were connected. Fig. 13 shows the coupling factors and Figs. 14 and 15 show the corresponding voltage on the VDD pin. A strong resonance is shown around 600 MHz in this case associated with the die and package. This resonance is predicted well in both the time and frequency domains. Coupling factors and measured and predicted voltages on Vdd when only the VDD pin was connected are shown in Figs. 16–18.

Fig. 15. Time-domain waveform of measured and predicted voltage on the Vdd pin (only Vdd/Vss1 pins connected).


Fig. 16.

1293

Measured coupling factors when only the VDD pin was connected.

Fig. 18. Time-domain waveform of measured and predicted voltage on the Vdd pin (only Vdd pin connected).

Fig. 17. Spectral density of measured and predicted voltage on the Vdd pin (only Vdd pin connected).

As before, a strong resonance appears around 600 MHz that is predicted well in both the time and frequency domains. The predicted time-domain waveform in Fig. 18 demonstrates some noncausality, in the sense that the response to the pulse begins before the pulse actually occurs. This noncausality is likely due to the translation of the frequency-domain model into the time domain. Despite this difference, the model still captures the major features of the waveform even in this challenging case. Each of the ESD tests shown here were performed with the IC powered OFF. Powering the IC may change the coupling factors and the measured voltages at the IC, due to changes in the internal IC circuit (e.g., due to the voltage-dependent on-die decoupling capacitance). The basic approach used to measure the coupling factors here, however, should work well for the powered case. IV. CONCLUSION Predicting the voltage and current on the pins of an IC due to an incident electromagnetic field is challenging. This paper introduced a methodology for predicting this voltage based on the measurement of coupling factors to the IC. These

coupling factors could be determined by applying known electric or magnetic fields to the IC in a TEM cell and measuring the IC response. The coupling factors predict the response due to the vertical electric field and the tangential magnetic fields, the fields most responsible for coupling to the IC. Comparison of simulations and measured voltages on the VDD pin of an 8-bit microcontroller show the method is capable of generating good results. While the methodology does not predict if the IC will fail, it does allow the user to predict the voltage and current on specific pins, and thus, gives them insight as to why the IC may fail under certain conditions. While a full-wave model could yield similar information, the use of measured coupling factors has some advantages over the full-wave technique. The coupling factors can be found without significant knowledge of the IC or PCB geometry or circuitry. Once the coupling factors are known, the voltages or currents can be calculated very quickly for a wide variety of incident fields. The incident fields can be found through full-wave simulations (which may be done quickly since they do not include the small features of the IC) or through measurements. This allows the user to relatively easily predict the response of the IC to different sources or changes in the relative magnitude or relative position of a particular source. REFERENCES [1] Integrated circuits - Measurement of electromagnetic immunity, IEC International Standard, IEC 62132, 2012. [2] EMC-Part 4-2: Testing and measurement techniques-Electrostatic discharge immunity test, IEC International Standard 61000-4-2, 2007. [3] C. Qing, J. Koo, A. Nandy, D. Pommerenke, J. S. Lee, and B. S. Seol, “Advanced full wave ESD generator model for system level coupling simulation,” in Proc. IEEE Int. Symp. Electromagn. Compat., Aug. 2008, pp. 1–6. [4] K. Wang, D. Pommerenke, R. Chundru, T. Van Doren, J. Drewniak, and A. Shashindranath, “Numerical modeling of electrostatic discharge generators,” IEEE Trans. Electromagn. Compat., vol. 45, no. 2, pp. 258–271, May 2003.

1294


[5] J. Koo, L. Han; S. Herrin, R. Moseley, R. Carlton, D. G. Beetner, and D. Pommerenke, “A nonlinear microcontroller power distribution network model for the characterization of immunity to electrical fast transients,” IEEE Trans. Electromagn. Compat., vol. 51, no. 3, pp. 611–619, Aug. 2009. [6] M. Stockinger, J. W. Miller, M. G. Khazhinsky, C. A. Torres, J. C. Weldon, B. D. Preble, M. J. Bayer, M. Akers, and V. G. Kamat, “Boosted and distributed rail clamp networks for ESD protection in advanced CMOS technologies,” in Proc. Electr. Overstress/Electrostatic Discharge Symp., Sep. 2003, pp. 21–25. [7] J. Zhang, D. G. Beetner, R. Moseley, S. Herrin, and D. Pommerenke, “Modeling electromagnetic field coupling from an ESD gun to an IC,” in Proc. IEEE Int. Symp. Electromagn. Compat., Aug. 14–19 2011, pp. 553–558. [8] J. Zhang, J. Koo, D. G. Beetner, R. Moseley, S. Herrin, and D. Pommerenke, “Modeling of the immunity of ICs to EFTs,” in Proc. IEEE Int. Symp. Electromagn. Compat., Jul. 2010, pp. 484–489. [9] V. Kasturi, S. Deng, T. Hubing, and D. Beetner, “Quantifying electric and magnetic field coupling from integrated circuits with TEM cell measurements,” in Proc. 2006 IEEE Int. Symp. Electromagn. Compat., Aug 14–16, 2006, vol. 2, pp. 422–425. [10] L. Ren and Z. Chen, “Improvement of expression for excitation by an electric dipole in GTEM cell,” J. Electron., vol. 19, no. 1, pp. 94–98, 2002. Ji Zhang received the B.S. and M.S. degree in electronic engineering from Tsinghua University, Beijing, China, in 2005 and 2008, respectively, and the Ph.D. degree from the Electromagnetic Compatibility Laboratory, Missouri University of Science and Technology, Rolla, MO, USA, in 2013. He is currently working for Cisco Systems Inc., San Jose, CA, USA. His research interests include the immunity of integrated circuits (IC) to electrical fast transient events and electrostatic discharge events, IC power delivery network (PDN) modeling, reconstruction of emission sources, modeling/optimization for high-speed series-link channel, and power integrity of board and ASIC package.

Xiang Li received the B.S. degree in electronic information engineering from Jilin University, Changechun, China, in 2007 and the M.S. degree from Electromagnetic Compatibility Laboratory, Missouri University of Science and Technology, Rolla, MO, USA, in 2010. She is currently an EMC Engineer working for Freescale Semiconductor, Inc., Austin, TX, USA. Her research interests include the radiated emission and immunity of integrated circuits (IC), conducted emission and immunity of the IC, electrostatic discharge event modeling, and crosstalk within harness bundles.

Richard Moseley (M’09) received the B.S.E.E. degree from Texas A&M University, College Station, TX, USA, in 1981. In 1981, he joined Motorola, and in March 2005, he joined the EMC engineering team as an Applications and Systems Engineer. He is currently a Senior EMC Engineer with the Microcontroller Solutions Group, Freescale Semiconductor, Inc., Austin, TX, USA, where he is involved in EMC test and validation engineer.

David Pommerenke (SM’03) received the Ph.D. degree from the Technical University Berlin, Berlin, Germany, in 1996. He was with Hewlett Packard for five years. In 2001, he joined the Electromagnetic Compatibility Laboratory, Missouri University of S&T, Rolla, MO, USA, where he is currently a Professor. He has authored or coauthored more than 200 papers and is inventor on 13 patents. His current research interests include system-level electrostatic discharge (ESD), electronics, numerical simulations, EMC measurement methods, and instrumentations and measurement/instrumentation ESD and EMC. Dr. Pommerenke is an Associate Editor for the IEEE TRANSACTIONS ON EMC.

Daryl G. Beetner (S’89–M’98–SM’03) received the B.S. degree in electrical engineering from Southern Illinois University at Edwardsville, Edwardsville, IL, USA, in 1990, and the M.S. and D.Sc. degrees in electrical engineering from Washington University in St Louis, St Louis, MO, USA, in 1994 and 1997, respectively. He is currently a Professor and the Chair of Electrical and Computer Engineering at the Missouri University of Science and Technology (formerly called the University of Missouri - Rolla), Rolla, MO, USA. He conducts research with the Electromagnetic Compatibility Laboratory at Missouri S&T on a wide variety of topics including EMC at the chip and system level and detection and neutralization of explosive devices. Prof. Beetner is an Associate Editor for the IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT.