Predistortion Linearization of an X-Band TWTA for ... - IEEE Xplore

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Predistortion Linearization of an X-Band TWTA for Communications Applications Xin Hu, Gang Wang, Zi-Cheng Wang, and Ji-Run Luo, Senior Member, IEEE

Abstract—Using a predistortion linearizer is an important way to improve traveling-wave tube amplifier (TWTA) linearity. However, predistortion linearizers cannot usually control amplitude modulation (AM)/AM and AM/phase modulation (PM) conversion separately, so it is difficult to satisfy linearization requirements for AM/AM and AM/PM conversion simultaneously. This paper proposes a predistortion linearizer with a relatively simple circuit structure, which provides good control of gain expansion over the input power dynamic range while keeping the amount of phase expansion small. Good linearity improvement has been obtained for a 45-W X-band TWTA with this predistortion linearizer. Index Terms—Expansion of gain and phase, nonlinear distortion, predistortion linearizer, traveling-wave tube amplifier (TWTA).

I. I NTRODUCTION

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HE CONSTANT demand for higher data rates in communication systems has resulted in more complex digital modulation techniques that require efficient and linear power amplifiers. Traveling-wave tube amplifiers (TWTAs) are still a very effective means for microwave and millimeter-wave power amplification [1]. Telecommunication payloads frequently utilize TWTAs. At saturation, TWTAs for communication applications can efficiently provide tens or hundreds of watts of output power, but their linearity is often inadequate for modern communications specifications. When multiple signals are sent through a communications system, TWTAs typically must be operated at a reduced power level with reduced efficiency in order to keep distortion at an acceptable level. Distortion is often measured as the carrier-to-intermodulation (C/IM) ratio (expressed in decibels) when multiple closely spaced frequencies are amplified. High levels of IM products can affect neighboring frequency bands and are not desirable. Placing a low-power predistortion linearizer in front of a TWTA is an effective way of reducing the IMs at a given output power level [2]–[4]. For a communications system, if a carrier-to-noise ratio of 15 dB [10 dB frequency modulation threshold and 5 dB for channel fading] is required and the IMs are to have a negligible effect, then a C/IM ≥ 25 dB is needed. To satisfy this Manuscript received November 12, 2010; revised January 19, 2011, February 2, 2011, February 12, 2011, and February 27, 2011; accepted February 28, 2011. The review of this paper was arranged by Editor W. L. Menninger. The authors are with the R&D Center for High Power Microwave Device, Institute of Electronics, Chinese Academy of Sciences, Beijing 100190, China (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TED.2011.2128321

Fig. 1. Schematic of a typical predistortion linearizer.

requirement, a TWTA (without a linearizer) would typically have to be driven such that its output power backoff (OPBO) from saturation is 5–7 dB [5]. With a predistortion linearizer, the same TWTA OPBO may be reduced to 3 dB, that is to say, the TWTA output power can be increased by 2–4 dB, which improves the margin of the communications link [5], [6]. Several predistortion methods using dual-gate GaAs FETs [4] and Schottky diodes [8]–[11] have been reported, both of which are capable of improving the amplitude and phase linearity of the TWTA. However, these predistortion linearizers cannot usually control amplitude modulation (AM)/AM and AM/phase modulation (PM) conversion separately, so it is difficult to satisfy the linearization requirement for AM/AM and AM/PM conversion simultaneously. Reference [12] reports a type of predistortion linearizer using two linearizers in parallel to independently control AM/AM and AM/PM conversion. However, this linearizer is aimed at solid-state power amplifiers applied in CDMA2000 multicarrier applications and is relatively complicated. In this paper, we propose a predistortion linearizer with a relatively simple circuit structure by combining a FET with a Schottky diode, which may provide good control of gain and phase expansion over the expected input power dynamic range. II. D ESIGN OF THE P REDISTORTION L INEARIZER Most modern predistortion linearizers are based on Schottky diodes, such as the small topology designed in [8] or the complex circuit structure designed in [9]–[11]. Near saturation, the nonlinearity of the Schottky diode is used to compensate the TWTA’s amplitude and phase response via these circuit topologies. The structure of a typical predistortion linearizer [9], [10] is shown in Fig. 1. jωt is fed into a 3-dB In Fig. 1, the RF input signal v in e power splitter and then divided into two equal components that pass through the delay-line path and the nonlinear path, respectively. The delay-line path contains a time-delay element to compensate for the delay through the various elements in

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the nonlinear path and ensures that signals recombine with the correct time relationship in the output terminal. The main element is a Schottky diode found in the nonlinear path [9], [10]. The nonlinear path gain is designed to decrease as the input power nears saturation. Assume that the delay-line path gain and the nonlinear path gain are G1 and G2 , respectively, and ◦ assume that the phase of the RF input signal ∠(v in ) is 0 and that the delay-line and nonlinear path phase shifts are ϕ1 and ϕ2 , respectively. When the phase difference (ϕ1 − ϕ2 ) between the two path signals is adjusted to an angle by the phase shifter such that cos(ϕ1 − ϕ2 ) < 0, the predistortion linearizer gain G and the phase variation Δϕ can be written as 1 1 G = |v (G1 − G2 )2 + (1 + α)G1 · G2 (1) out /v in | = 4 2 v1 sin(ϕ1 − ϕ2 ) Δϕ = ∠(v + ϕ2 out /v in ) = arctan v1 cos(ϕ1 − ϕ2 ) + v2 G1 · sin(ϕ1 − ϕ2 ) = arctan (2) + ϕ2 G1 · α + G2 where α = cos(ϕ1 − ϕ2 ) (α < 0), v1 = (1/2)v in · G1 , v2 = (1/2)v in · G2 . As the drive power level increases, the gain of the Schottky diode in the nonlinear path decreases, and the phase difference (ϕ1 − ϕ2 ) between the two path signals remains at a large obtuse angle (cos(ϕ1 − ϕ2 ) < 0). In (1), if G1 stays constant, the positive value (1/4)(G1 − G2 )2 will increase faster than the positive value (1/2)(1 + α)G1 · G2 as G2 decreases, so the overall gain G increases. Because the value of the phase variation Δϕ is between ϕ2 and ϕ1 , one can infer that 0 < (Δϕ − ϕ2 ) < (ϕ1 − ϕ2 ) < π and that the arctangent term in (2) is in the range 0–π. G1 · α + G2 in (2) will change from a positive value to a negative value with a decrease in G2 . Because arctan(x) ∈ (0, π) is a monotonically increasing function of x, the overall phase variation Δϕ increases as G2 decreases. Therefore, once the attenuator and phase shifter are adjusted properly, the amount of linearizer gain and phase shift expansion is determined; however, it is difficult to satisfy a linearization requirement for AM/AM and AM/PM conversion simultaneously. If there is an electronic device that can produce the gain expansion in the Schottky diode linear region and a bigger phase shift than the Schottky diode simultaneously, one can replace the microstrip line and phase shifter in Fig. 1 with this electronic device. It can be seen in (1) and (2) that the overall gain G and phase variation Δϕ increases with an increase in G1 if G2 keeps constant and cos(ϕ1 − ϕ2 ) < 0. Simulations and analyses with the Advanced Design System (ADS) code [13] show that some FETs have a gain response that gradually increases versus drive, while that the gain response for a Schottky diode stays nearly constant over a −40 to 0 dBm input power dynamic range, as shown in Fig. 2. The FET is operating close to cutoff. When the gate voltage Vgs of the FET is close to cutoff, its transconductance is approximately equal to zero, but a small Vgs voltage variation can cause a large change of the FET’s transconductance, and so the FET behaves nonlinearly [14]. It can be seen in Fig. 2 that the FET has gain expansion over a −20 to 0 dBm input power dynamic range. The Schottky

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Fig. 2.

Gain response of a FET and a Schottky diode versus input power.

diode gain curve, at Vb1 bias voltage, shows that the Schottky diode gain remains constant over a −40 to 0 dBm input power dynamic range. Different devices will cause different phase shift variations, thus, the angle between the FET and the Schottky diode can be an obtuse angle (cosine < 0) over a −40 to 0 dBm input power dynamic range. Fig. 3 shows the phase shift variation of the FET operating close to cutoff [Fig. 3(a)] and the Schottky diode at Vb1 bias voltage [Fig. 3(b)]. As the input power becomes greater than −20 dBm, the FET phase shift gradually increases, with the total amount of phase shift variation being about 13◦ over a −40 to 0 dBm input power dynamic range. On the other hand, the total amount of phase shift variation for the Schottky diode is about 1◦ over the −40 to 0 dBm input power dynamic range. When the same input power is fed into the FET and the Schottky diode, respectively, the angle between them is about 120◦ and the total amount of phase shift variation is only about 13◦ over the input power dynamic range. Based on the above arguments, gain expansion and an obtuse phase angle difference between two path signals can thus be realized with a FET operating close to cutoff and a Schottky diode operating in the linear region. This is different from the typical predistortion linearizer in which the Schottky diode operates in its saturation region [8]–[11]. Thus, in this paper, we propose to replace the attenuator and phase shifter in Fig. 1 with a FET, as shown in Fig. 4. The input signal is again fed into a 3-dB power splitter, and it is divided into two equal components, which respectively pass through the FET path and the Schottky diode path, with corresponding gains of G1 and G2 , respectively. In the FET path, the gain G1 will increase as the FET nears saturation, whereas in the Schottky diode path, the gain G2 will remain the same at the Vb1 bias voltage. Thus, the overall gain G and phase variation Δϕ will increase. If one increases the bias voltage of the Schottky diode from Vb1 to Vb2 (Vb2 > Vb1 ), the gain and phase response of the Schottky diode with the different bias can be obtained using ADS computer simulations, as shown in Fig. 5. At a small input power level, the Schottky diode path gain remains constant for either bias. At a large input power level, the Schottky diode path gain with the Vb1 bias voltage still remains G2 , and the path gain with the Vb2 bias voltage changes from G2 to G2 + ΔG2 (ΔG2 < 0) as the result of the nonlinearity introduced by

This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. HU et al.: PREDISTORTION LINEARIZATION OF AN X-BAND TWTA FOR COMMUNICATIONS APPLICATIONS

Fig. 3.

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Phase variation of a FET and a Schottky diode over the same input power dynamic range. (a) FET. (b) Schottky diode.

III. S IMULATION AND E XPERIMENTAL R ESULTS

Fig. 4.

Schematic of proposed predistortion linearizer.

the increased bias voltage. The change of the bias voltage has little effect on the phase shift variation of the Schottky diode path, however, i.e., only 4◦ in the input power range of −40 to 0 dBm. When the Schottky diode bias voltage is set to Vb2 , the overall gain response can be written as in (3), shown at the bottom of the page. Compared to (1), which is the overall gain response of the original linearizer design at the Schottky diode Vb1 bias voltage, the increased bias voltage with the new linearizer design generates an additional (1/4)(ΔG2 )2 + (1/2)ΔG2 · (G1 · α + G2 ) term inside the square root in (3). When the input power is greater than −10 dBm, the FET gain G1 is much greater than G2 , which makes |G1 α| greater than G2 (α < 0), i.e., (1/2)ΔG2 · (G1 · α + G2 ) > 0. In the proposed predistortion linearizer, the amount of overall gain expansion at the Vb2 bias voltage is therefore greater than that at the Vb1 bias voltage over the rated input power dynamic range. However, the change of ΔG2 has little effect on the amount of phase expansion. Thus, by controlling the bias voltage of the Schottky diode, the proposed predistortion linearizer can effectively adjust and control the amount of gain expansion with only a small phase expansion variation.

G= = =

An X-band predistortion linearizer circuit, monolithic microwave integrated circuit (MMIC) amplifiers, and a variable attenuator were designed and fabricated on a 10-mil-thick twometallization-layer substrate with εr = 2.55. The predistortion linearizer was developed for the frequency range 8.38– 8.58 GHz. A photograph of the X-band linearized TWTA (LTWTA) is shown in Fig. 6. The predistortion linearizer concept presented here is not limited to X-band—it can also be applied at higher frequencies, such as Ku- or Ka-band. A model of the predistortion linearizer discussed here was simulated with ADS to explore its gain and phase expansion. The results show that the predistortion linearizer proposed here can achieve the appropriate gain and phase expansion over a −40 to −6 dBm input power dynamic range in order to effectively linearize a TWTA. Then, in order to verify the simulation results, the predistortion linearizer circuit was tested. Fig. 7 shows the simulated and measured gain and phase response of the predistortion linearizer. With regard to the simulated results in Fig. 7, the rising slope of the gain response for the Vb2 bias is larger than for the Vb1 bias [Fig. 6(a)], i.e., the gain difference between the two biases increases from about 0 dB at the −40 dBm input power level to 1.4 dB at the −6 dBm input power level. The phase response, on the other hand, varies only slightly for the different bias voltages, i.e., the phase difference between the two biases varies from about 2◦ at the −40 dBm input power level to 0.5◦ at the −6 dBm input power level, as shown in Fig. 7(b). The measured results are in good agreement with the simulated results shown in Fig. 7. Thus, the predistortion linearizer proposed here is not only capable of providing good control of the gain and phase expansion, it can also reach greater gain expansion with little effect on the

1 2 1 1 G1 + (G2 + ΔG2 )2 + G1 (G2 + ΔG2 ) cos(ϕ1 − ϕ2 ) 4 4 2 1 1 1 1 1 (G1 − G2 )2 + (1 + α)G1 · G2 + (ΔG2 )2 + ΔG2 · G2 + αΔG2 · G1 4 2 4 2 2 1 1 1 1 (G1 − G2 )2 + (1 + α)G1 · G2 + (ΔG2 )2 + ΔG2 · (G1 · α + G2 ) 4 2 4 2

(3)



Fig. 5. Gain and phase responses of a Schottky diode with different bias voltages. (a) Gain. (b) Phase.

Fig. 6. Photograph of the X-band LTWTA, showing a predistortion linearizer, MMIC amplifiers, a variable attenuator, and a high-power TWTA.

phase expansion by adjusting the bias voltage of the Schottky diode. For a typical and practical TWTA, the dynamic range of its input power is about 22 dB from small signal to saturation, which means that the predistorted signal at the input terminal of the TWTA must compensate its nonlinearity over a 22-dB dynamic range [10], [11]. Fig. 8 shows the measured gain and phase responses of the predistortion linearizer over the 8.38–8.58 GHz range for different two different input power levels. The linearizer delivers nearly 6 dB gain expansion and 45◦ phase expansion from small-signal input drive (−22.96 dBm) to saturated input drive (−6.92 dBm), and the gain of the linearizer varies by less than 1 dB over the frequency band. The AM/AM and AM/PM conversion characteristics of the TWTA and that of the LTWTA were tested at the center frequency of 8.48 GHz, as shown in Figs. 9 and 10. In Fig. 9, the gain and phase change of the TWTA is 6 dB and 45.8◦ over a −30 to −13.96 dBm input power dynamic range [the input power −13.96 dBm (point M1 on the curves) corresponds to the TWTA saturated output power]. The measured saturated output power of the TWTA is 45.32 dBm. In Fig. 10, the gain and phase change of the LTWTA from small-signal input drive (−30 dBm) to saturated input drive (−6.92 dBm and marked by M2) are improved to 1 dB and 3◦ , respectively.

A two-tone signal of 8.48 and 8.49 GHz was injected into the TWTA and also into the LTWTA at +25 ◦ C and at varying drive levels. The two-tone C/IM performance versus OPBO is shown in Fig. 11. As the OPBO is increased, the LTWTA C/IM is improved much more than the TWTA C/IM. At 1 dB OPBO, the improvement in C/IM from TWTA to LTWTA is 0.5 dB. At 3 dB OPBO, the improvement is 11.8 dB. In other words, a 25.5-dB C/IM with only 3 dB OPBO can be achieved with the LTWTA, whereas without the linearizer, an OPBO of nearly 7 dB would be needed to achieve the same 25.5-dB C/IM. Our experiments show that the electrical performances of the TWTA discussed here change very little in an environmental temperature range of −25 to +75 ◦ C. Therefore, the linearizer discussed here was tested over a temperature range from −25 ◦ C to +75 ◦ C, and the TWTA was kept at +25 ◦ C all the time. In an environmental temperature range of +10 to +50 ◦ C, the performance of the linearizer changes very little. When the environmental temperature is out of this range, the performance change of this linearizer reduces the LTWTA C/IM. However, this problem can be improved by adjusting the bias voltage of the linearizer. The experiments show, in order to obtain the same C/IM as that at +25 ◦ C, that the gate bias voltage must be decreased and the drain bias voltage must be increased when the environmental temperature is higher than +50 ◦ C, and that the gate bias voltage must be increased and the drain bias voltage must be decreased when the environmental temperature is lower than +10 ◦ C. The measured LTWTA twotone C/IM versus OPBO for various temperatures −10 ◦ C, +10 ◦ C, +40 ◦ C, and +65 ◦ C are shown in Fig. 12 before and after the adjustment for the bias voltages at the temperatures of −10 ◦ C and +65 ◦ C. If without gate bias voltage adjustments, as the OPBO is increased, the LTWTA C/IM at −10 ◦ C and +65 ◦ C are deteriorated much more than the LTWTA C/IM at 25 ◦ C. However, with proper gate bias voltage adjustments, these two-tone C/IM values are in good agreement with the C/IM at +25 ◦ C (shown in Fig. 11). IV. C ONCLUSION The simulation, design, and experimental demonstration of a predistortion linearizer for a TWTA have been presented. The predistortion linearizer combines a FET and a Schottky diode to compensate for the TWTA’s nonlinear behavior. With


Fig. 7.

Gain and phase responses of the predistortion linearizer. (open circle: measured results; filled triangle: simulated results). (a) Gain. (b) Phase.

Fig. 8.

Measured gain and phase responses of the predistortion linearizer in the frequency range of 8.38–8.58 GHz. (a) Gain. (b) Phase.

Fig. 9.

Measured gain and phase characteristics of the TWTA at 8.48 GHz. The TWTA output power saturates at −13.96 dBm (M1). (a) Gain. (b) Phase.

a simple circuit structure, the amount of gain expansion can effectively be controlled over an input power dynamic range with small phase change. The experimental linearizer, operating

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over the frequency range 8.38–8.58 GHz, generated 6 dB gain expansion and 45◦ phase shift, accurately inverting the gain and phase characteristics of a 45-W saturated output power TWTA



Fig. 10. Measured gain and phase characteristics of the LTWTA at 8.48 GHz. The LTWTA output power saturates at −6.92 dBm (M2). (a) Gain. (b) Phase.

over an input power dynamic range of −25 to −6.92 dBm. The resulting LTWTA required only 3 dB OPBO to reach a two-tone C/IM ratio of 25.5 dB, compared to 7 dB OPBO for the unlinearized TWTA. Variation in the LTWTA performance resulting from changes in the environmental temperature was able to be compensated by properly adjusting the bias voltages of the Schottky diode in the linearizer. R EFERENCES

Fig. 11. Two-tone C/IM comparison between the X-band TWTA and LTWTA.

Fig. 12. C/IM comparison between the X-band TWTA and the LTWTA for different environmental temperatures: −10 ◦ C, +10 ◦ C, +40 ◦ C, and +65 ◦ C without and with proper adjustments of the linearizer’s FET bias voltage at each temperature.

[1] J. Weekley and B. Mangus, “TWTA versus SSPA: A comparison of onorbit data,” IEEE Trans. Electron Devices, vol. 52, no. 5, pp. 650–652, May 2005. [2] R. Gray, A. Katz, and R. Dorval, “Advances in millimeter-wave linearization,” in Proc. 13th Ka Broadband Commun. Conf., Torino, Italy, Sep. 24–26, 2007, pp. 76–79. [3] J. Yi, Y. Yang, M. Park, W. Kang, and B. Kim, “Analog predistortion linearizer for high-power RF amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 12, pp. 2709–2713, Dec. 2000. [4] M. Kumar, J. C. Whartenby, and H. J. Wolkstein, “Predistortion linearizer using dual-gate MESFET for TWTA and SSPA used in satellite,” IEEE Trans. Microw. Theory Tech., vol. MTT-33, no. 12, pp. 1479–1488, Dec. 1985. [5] A. Katz, “TWTA linearization,” Microw. J., vol. 39, no. 4, pp. 78–90, Apr. 1996. [6] P. B. Kenington, High-Linearity RF Amplifier Design. Boston, MA: Artech House, 2000. [7] R. Inada, H. Ogawa, S. Kitazume, and P. Desantis, “A compact 4-GHz linearizer for space use,” IEEE Trans. Microw. Theory Tech., vol. MTT-34, no. 12, pp. 1327–1332, Dec. 1986. [8] K. Yamauchi, K. Mori, M. Nakayama, Y. Mitsui, and T. Takagi, “A microwave miniaturized linearizer using a parallel diode with a bias feed resistance,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 12, pp. 2431–2435, Dec. 1997. [9] J. F. Villemazet, P. Moroni, B. Cogo, T. Peyretaillade, J. Maynard, J. L. Cazaux, C. Laporte, and L. Lapierre, “Wide band linearizer for improved Ka-band telecommunication satellite,” in Proc. AIAA 19th ICSS Conf., 2001, pp. 1–5. [10] W.-M. Zhang and C. Yuen, “A broadband linearizer for Ka-band satellite communication,” in Proc. IEEE MTT-S Dig., 1998, pp. 1203–1206. [11] H.-Y. Jeong, S.-K. Park, N.-S. Ryu, Y.-C. Jeong, I.-B. Yom, and Y. Kim, “A design of K-band predistortion linearizer using reflective Schottky diode for satellite TWTAs,” in Proc. 35th Eur. Microw. Conf., Paris, France, 2005, pp. 341–344. [12] C. H. Park, F. Beauregard, G. Carangelo, and F. M. Ghannouchi, “An independently controllable AM/AM and AM/PM predistortion linearizer for CDMA2000 multi-carrier applications,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Phoenix, AZ, May 2001, pp. 53–56. [13] Advanced Design System, Version 2002. [14] D. M. Pozar, Microwave Engineering, 3rd ed. New York: Wiley, 2006.


Xin Hu was born in 1985. He received the B.S. degree in electrical engineering from Huazhong University of Science and Technology, Wuhan, China, in 2007. He is currently working toward the Ph.D. degree in physical electronics from the R&D Center for Microwave Device and Technology, Institute of Electronics, Chinese Academy of Sciences, Beijing, China. His doctoral research concerns a transmitter design including MPMs, RF power amplifier linearization techniques, and high efficient power amplifier design with applications for radar and wireless

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communications.

Zi-Cheng Wang received the B.S. degree in physics and the M.S. degree in solid-state electronics from Yunnan University, Kunming, China, in 1986 and 1989, respectively, and the Ph.D. degree in microwave electronics from the Beijing Vacuum Electronics Research Institute, Beijing, China, in 1998. He joined the Institute of Electronics, Chinese academy of Science, Beijing, in 1999 and became a Senior Engineer in 2002 and a Senior Researcher in 2004. His recent work focus on the modeling, designing, and manufacturing of TWTs, BWOs, and other vacuum devices especially in terahertz domain.

Gang Wang received the B.E. degree in electronic engineering from Harbin Shipbuilding Engineering Institute, Harbin, China, in 1992. In 1992, he joined Yangzhou Shipborne Electronic Instruments Institute and was engaged in development research on power transmitter based on microwave vacuum tube. From 2003 to 2007, he was engaged in development research on radar system. From 2007, he was engaged in development research on spaceborne traveling wave tube amplifier (TWTA) at the Institute of Electronics, Chinese Academy of Science, Beijing, China. His main direction is the development of electronic power conditioners (EPCs).

Ji-Run Luo (SM’10) was born in 1957. He received the B.Sc. degree in radio physics from Jiangxi University (now Nanchang University), Nanchang, China, in 1982 and the M.Eng. and Ph.D. degrees in electron physics from the Institute of Electronics, Chinese Academy of Sciences (IECAS), Beijing, China, in 1985 and 2000, respectively. He joined the High-Power Microwave Devices Laboratory at IECAS in 1985. From 1985 to 1993, he was involved in theoretical and experimental research on gyrotron oscillators, solid-state microwave power amplifiers, and microwave energy applications in ceramic material sintering and crude oil dehydration. He became a Senior Engineer in 1993. From 1994 to 2002, his research activities were mainly devoted to high power klystron development and millimeter-wave materials processing. He became a Research Professor in IECAS, China, in 1998 and a Professor in the Graduate University, Chinese Academy of Sciences, in 2002. From December 2002 to November 2003, he was with the Institute for Pulsed Power and Microwave Technology, Research Center Karlsruhe (FZK), Karlsruhe, Germany, where he worked on the theoretical model and experimental verification of the microwave heating process of powdered metals. In December 2003, he returned to IECAS. His current research projects are the development of high-power gyro-amplifiers and klystrons as well as microwave energy applications for materials processing.