Quad-Band High-Temperature Superconducting ... - IEEE Xplore

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 62, NO. 12, DECEMBER 2014

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Quad-Band High-Temperature Superconducting Bandpass Filter Using Quadruple-Mode Square Ring Loaded Resonator Haiwen Liu, Senior Member, IEEE, Baoping Ren, Xuehui Guan, Member, IEEE, Pin Wen, and Yan Wang

Abstract—In this paper, a compact quad-band high-temperature superconducting (HTS) bandpass filter using a quadruple-mode square ring loaded resonator (SRLR) is introduced. The even- and odd-mode method is applied to investigate the equivalent circuits of the proposed quadruple-mode SRLR. The design graphs for the relationship of the parameters of electrical length and resonance performances are then set up. Based on the analysis above, four allocated resonant modes can be simultaneously excited and easily tuned. Meanwhile, multi-transmission zeros are created due to the different propagation paths of the square ring structure. Signal-interference theory is also adopted to explain the generating mechanism of transmission zeros. Moreover, a meander coupled-line technique is realized to adjust one uncertain resonant frequency to meet the target specifications of the designed quad-band filter. To verify this methodology, a second-order microstrip HTS filter operating at 2.45/3.5/5.2/5.8 GHz for wireless local area networks and worldwide interoperability for microwave access potential applications is designed using two quadruple-mode SRLRs. The quadruple-mode SRLRs are coupled with a pseudo-interdigital coupling structure for achieving the desired coupling degree conveniently, which also miniaturize the circuit size. This filter was fabricated on a 2-in-diameter 0.5-mm-thick MgO wafer with double-sided YBa Cu Oy thin films. The filter component was measured at the temperature of 77 K. Measured results agree with the theoretical results and show that the insertion losses in passbands are less than 0.3 dB, which exhibit superiority in midband insertion loss. Index Terms—Bandpass filter (BPF), high-temperature superconducting (HTS), quad band, quadruple mode, square ring loaded resonator (SRLR).

I. INTRODUCTION

W

ITH THE increasing development of multi-service wireless communication networks, microwave components and systems that support various modern communication standards have become a widespread tendency. Therefore, as a key passive component in the RF front-end, multi-band

Manuscript received May 16, 2014; revised August 03, 2014 and October 16, 2014; accepted October 23, 2014. Date of publication November 11, 2014; date of current version December 02, 2014. This work was supported by the National Science Foundation of China under Grant 61061001 and Grant 61161005 and by the International Cooperation Funds and Science and Technology Innovation Team of Jiangxi Province of China under Grant 20121BDH80015 and Grant 20122BCB24025. H. Liu, B. Ren, X. Guan, and P. Wen are with the School of Information Engineering, East China Jiaotong University, Nanchang 330013, China (e-mail: [email protected]). Y. Wang was with the School of Information Engineering, East China Jiaotong University, Nanchang 330013, China. She is now with Railway Communications, Railway Design Institute, Shanghai 200070, China Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2014.2366147

bandpass filter (BPF) design with compact size, high performance, and low loss is in great demand for enhancing system functionality. Being an essential part of the multi-band operation system, the quad-band BPF has achieved great attention over recent years [1]–[9]. Basically, these reported quad-band design methods can be classified into three typical categories. The first approach is utilizing the multi-layered structure to achieve the quad-band response [1]–[3]. In [1], a dual-plane structure composed of a microstrip and a defected ground structure slot is employed to obtain quad-band responses. In [2], a quad-band BPF was proposed using coupled microstrip uniform impedance resonators (UIRs) and defected ground structures resonators. In [3], the authors proposed an improved coplanar waveguide (CPW)-fed quad-band BPF based on dual-mode double-square-ring resonators. However, the aforementioned quad-band BPFs require multiple types of resonators or multi-layered fabrication technology, which increase fabrication difficulty and cost. The second approach is to cascade two types of the dual-band BPF in parallel, such as symmetric stepped-impedance resonators (SIRs) [4], [5], and fork-type resonators [6]. Each set of resonators produce two passbands, and four passbands are therefore achieved by two sets of these resonators, but the circuits always occupy a relatively large size. The third approach is adopting coupled quadruple-mode resonators for quad-band BPFs [7]–[9]. Four resonant modes are excited in desired frequencies using one quadruple-mode resonator, thus, a compact filter configuration can be obtained. Nevertheless, their higher order passbands are with a noticeable insertion loss of more than 1 dB. Recently, high-temperature superconducting (HTS) materials become more and more attractive in designing the RF/microwave filters because of their lower loss and excellent performance. Generally, the current study on HTS filter mainly focuses on the single passband filter [10], [11], and only a few attempts have thus far been made on multi-band HTS BPFs [12]–[16]. For example, dual-/triple-band HTS filters were designed by using multi-spiral resonators in [12] and [13]. In addition, dual-band HTS filters have been realized by using quarter-wavelength SIRs [14], embedded split-ring resonators [15], or multi-stub loaded resonators [16]. It is noted that some complex synthesis algorithms are required and a coupling matrix needs to be extracted for achieving good performance. Thus, they take more effort to fulfill design parameters. Additionally, very little research has been reported thus far for quad-band HTS BPF design in comparison with those of dual-/tri-band filters.

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Today, a square ring loaded resonator (SRLR) was firstly proposed in [17] and its multi-mode feature has been proven in [18]. Furthermore, a modified quadruple-mode SRLR unit cell was proposed and applied to design a compact and high-selectivity dual-band BPF [19]. Comparing with the transversal signal-interference filtering cell [20], two extra open stubs are distributed in both sides of the SRLR. These two stubs act as the input and output terminals of the resonator, which differs from the works in [21]. Thus, more interesting resonant characteristics will be obtained. In addition, the stubs can be in an arbitrary location and such that some higher order resonant modes can be more flexibly produced and controlled than the ones discussed in [22] and [23]. In this work, the conventional multi-mode ring loaded resonator (RLR) is rearranged and its transmission line model (TLM) is established. The even- and odd-mode method is applied to investigate its equivalent circuits and resonant characteristics. Moreover, the design procedure of a quad-band BPF using a quadruple-mode SRLR is further studied. A second-order quad-band HTS filter based on two coupled quadruple-mode resonators is then designed and implemented for demonstration. HTS thin-film YBCO deposited on a 0.5-mm-thick MgO substrate is utilized to fabricate the circuit. This paper is organized as follows. In Section II, the characteristics as well as design principles of the multi-mode SRLR are investigated. Section III presents the design of a second-order quad-band HTS BPF using the proposed coupled quadruplemode SRLRs. Its implementation and verification are illustrated in Section IV. Finally, Section V draws some brief conclusions for this filter design.

Fig. 1. SRLR. (a) Basic structure. (b) TLM with two lumped capacitors (C).

II. CHARACTERISTICS OF QUADRUPLE-MODE SRLR A. Structure Analysis Fig. 1(a) depicts the basic structure of the proposed quadruple-mode RLR. It is composed by a one-wavelength square ring resonator and two open microstrip lines that are attached to both sides of the ring. Its TLM is given within the dashed box in Fig. 1(b). to and to are denoting the physical lengths and widths of the corresponding microstrip line segments, respectively. This TLM consists of six transmission-line sections. Its corresponding electrical lengths and characteristic admittances are referred to and , respectively. Here, , , , and is the propagation constant. For analysis and design simplicity, it is noted that is assumed in this work. For the proposed RLR in Fig. 1(a), its physical length , , and can be chosen to an arbitrary value so that more design freedoms and some interesting features will be obtained. As presented in Fig. 2, four special cases for this new resonator with different electrical length combinations are listed and discussed. B. Resonance Characteristics Fig. 1(b) shows an equivalent circuit model of the proposed RLR with two identical lumped capacitors that are located at the input and output ports. The lumped capacitor is applied to provide a weak excitation for this structure.

Fig. 2. Special cases for SRLR with different electrical length combinations.

To explore the resonant characteristics of the multi-mode RLR from Fig. 2, as an example, , , and are set at 70 , 20 , and 64 , respectively. All the electrical lengths are in respect to a fundamental frequency of 3 GHz and the characteristic impedance of transmission lines is set to 50 . The corresponding frequency response of the proposed RLR is presented in Fig. 3. It is observed that four resonant modes, called , , , and , are excited and located at 3, 3.48, 5.62, and 6.13 GHz, respectively. This circuit model is simulated by Agilent Technologies’ simulation tool Advance Design System (ADS) 2010. In Fig. 3, two frequency differences between the resonant modes are defined as (1) To further study the mode splitting of the RLR, Fig. 4 illustrates the variations of and with different electrical length ratio, , of the square ring. Here, to be stated, that remains

LIU et al.: QUAD-BAND HTS BPF USING QUADRUPLE-MODE SRLR

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Fig. 5. (a) Even- and (b) odd-mode excitation equivalent circuits of the proposed SRLR in Fig. 1.

Fig. 3. Frequency response of pF with ,

magnitude under weak coupling , and .

one-port network with open- and short-circuited ends in the M.W. and E.W. locations, respectively, as shown in Fig. 5. As illustrated in Fig. 5(a), and represent the corresponding input admittances from the left and right sides of the one-port bisection network under the condition of even-mode excitation. Its resonant condition can be derived as (2) where (3) (4) represents the input Similarly, as indicated in Fig. 5(b), admittance from the right side of the one-port bisection network under the condition of odd-mode excitation. Thus, its resonant condition occurs at

Fig. 4. Variation of under the condition of

and

with different electrical length ratio and .

(5) where

almost unchanged while the other three frequencies, i.e., , , and , are varying with . As depicted in Fig. 4, it can be seen that decreases and increases as is enlarged. Besides, there are three highlight points, , , and , indicated in Fig. 4. At point , its corresponding and are equal to 1 and 0, respectively. At point , and are equal to 2.4 and 0, respectively. It suggests that two resonant frequencies of and or and will merge when is chosen to 2.4 or 1, respectively. Accordingly, only three transmission poles may appear in this case. At point , is equal to when is chosen to about 1.6. In this case, it implies that quadruple-mode characteristics of the new resonator could be produced with identical frequency bandwidths by the two pairs of resonant modes. Thus, by adjusting , the proposed RLR can provide more freedom to exhibit different characteristics for dual-/tri-/quad-band or wideband designs. To give a deep insight into the above-mentioned phenomenon, an even- and odd-mode method is utilized. Under the even- or odd-mode excitations, the plane of symmetry in Fig. 1(b) behaves as a perfect magnetic wall (M.W.) or an electric wall (E.W.), respectively, and its bisection becomes a

(6) Substituting (3) and (4) into (2), the resonance condition at the even modes can then be expressed as (7) Similarly, by substituting (3) and (6) into (5), the resonance condition at the odd modes yields (8) Therefore, the resonant frequencies at the even- and oddorder modes can be ascertained from the roots of (7) and (8). Here, is assumed for simplification. Thus, (7) and (8) can be rewritten as

(9) (10)

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Fig. 7. Equivalent circuit schematic with two propagation paths of square ring.

a minor effect on . especially remain approximately constant and equal to 1 when approaches 0. This is due to the fact that the proposed RLR becomes a stub loaded resonator structure that excites only two modes, which is consist with the discussions in Section II-A. From Fig. 6(a) and (b), four resonant modes can be quickly determined when and are given. Moreover, the separation degree between four modes is varied when choosing different and . Thus, the proposed RLR can exhibit a variety of filtering responses. C. Generating Mechanism of Transmission Zeros

Fig. 6. Design graphs for the proposed quadruple-mode SRLR with varied resonant modes. (a) Normalized odd-mode resonant frequencies versus and . and . (b) Normalized even-mode resonant frequencies versus

where and . and are the normalized frequencies to . and are the even- and odd-mode resonant frequencies, respectively. is the fundamental frequency of one half-wavelength resonator with electrical length . In this work, all of the electrical lengths are calculated at . Once and are determined, there exist two solutions of at the finite scope by solving (9) and the same for by solving (10). It reveals that two even modes and two odd modes can be generated by the proposed RLR. Based on the above discussion, two net-type graphs of the quadruple-mode SRLR are plotted in Fig. 6. Fig. 6(a) illustrates two normalized even modes, and with varied and . It can be seen that and are both decreased as increases from 1 to 30 when is fixed at a certain value. Similarly, the tendencies of the variation are the same when increases. The other two normalized odd modes, and , versus and are depicted in Fig. 6(b). As shown in the figure, the larger and are, the smaller and are. In addition, a wider range of is obtained as increases from 1 to 90 . Also, it can be observed that the range of variation of is small, which implies that and have

Look back at Fig. 3(b), two transmission zeros located at 4.22 and 5.19 GHz can be observed due to the transversal signal interference. As studied in [24], the square ring section of the proposed RLR provides two different propagation paths between the input and output ports. The two signal currents will then cancel each other on the output side. Therefore, one or more transmission zeros can be produced and an excellent attenuation characteristic near the passband will be achieved. An equivalent circuit of the square ring is depicted in Fig. 7, which consists of the upper and lower sections. The total -parameters of this circuit can be obtained by adding the individual -parameters of two propagation paths connecting two ports. Based on the - and -parameter operations of the TLM, the total transfer admittance can be calculated and is obtained as (11a) where (11b) (11c) As described in [23], the transmission zeros always happen is equal to zero. Thus, we obtain at the frequencies where (12) The transmission zeros can then be found by solving (12) and as follows: for

(13)

for

(14)

It is noted that (13) is based on the signal-interference technique and is the same as the condition in [25, eq. (2)]. In general, two transmission zeros ( and ), as indicated in Fig. 3(b),


are determined by (13) with spectively.

and (14) with

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, re-

III. DESIGN OF QUAD-BAND HTS MICROSTRIP FILTER A. Initial Design According to the above-mentioned analysis, the mode-splitting characteristics of the proposed quadruple-mode RLR can be controlled by the electrical length parameters of the square ring. To confirm validity and flexibility of this new resonator on constructing a multi-band filter, a quad-band HTS filter with excellent insertion-loss performances is designed by choosing appropriate values of and from Fig. 6. The detailed design steps is developed and given below. In this paper, a quad-band response operated at 2.45/3.5/5.2/5.8 GHz for wireless local area networks (WLANs) and worldwide interoperability for microwave access (WIMAX) potential applications is required to design. As revealed in Fig. 6, two of the even or odd modes of the proposed RLR can be firstly determined to the desired frequency channels by properly choosing the values of and . The other two resonant modes can then therefore be be ascertained based of the known electric lengths. Thus, four resonant frequencies, denoted as , , , and of the designed resonator will be finally obtained. In this design, is chosen to 2.4 GHz for normalization. Thus, two normalized even-mode resonant frequencies are computed as follows: and .A suitable point (solid red dot in online version) for realizing the normalized frequencies at the even mode can be quickly found in Fig. 6(a). The corresponding and are then found to be 23 and 31 , respectively. Based on the condition of with the known , is obtained as 67 . Subsequently, the normalized odd-mode resonant frequencies, and can be acquired based on the known and from Fig. 6(b). As the solid red dot indicated on the net-type design graph in Fig. 6(b), the corresponding and are found at about 1.06 and 2.46, respectively. and can therefore then be determined to be 2.53 and 5.9 GHz. As discussed above, the electrical lengths of , , and are known. Thus, the initial physical dimensions of the SRLR indicated in Fig. 1(a) can be acquired by using the ADS LineCalc tool. Based on the computed electrical lengths, the microstrip structure of the proposed RLR can be obtained. Its frequency response and the corresponding current density distributions are simulated by using the full-wave electromagnetic (EM) simulator Sonnet and are given in Figs. 8 and 9, respectively. The substrate used in this paper is MgO with a relative dielectric constant of 9.78 and a thickness of 0.5 mm. As illustrated in Fig. 8, four resonant modes, , , , and , are located at 2.52, 3.54, 5.89, and 5.81 GHz, respectively. However, it is found that there exists a frequency discrepancy in the second odd-mode resonant frequency between the computed result (@ 5.89 GHz) and the target specification (@ 5.2 GHz), as illustrated in Fig. 8. This reason is that there exist only three design parameters, , , and , of the proposed SRLR, which is in general not enough for four resonant modes

Fig. 8. Comparison of frequency responses of the quadruple-mode RLR between the theoretical results (black solid line) and the adjusted results (blue dashed line in online version).

Fig. 9. Simulated current density distributions of the quadruple-mode RLR at GHz. (b) GHz. (c) resonant frequencies. (a) GHz. (d) GHz.

controllable simultaneously. To tackle this problem, a method to adjust the second odd-mode resonant frequency and remove is needed. Fig. 9 shows the simulated current density distribution of the designed SRLR at resonant frequencies. It can be observed that the current is mainly concentrated on the two open microstrip lines and the lower horizontal segment of the square ring at the first odd-mode frequency of , as presented in Fig. 9(a). At the second odd-mode frequency of , the current is mainly concentrated on two open microstrip lines and both horizontal segments of the square ring from Fig. 9(d). For even-mode resonant frequencies, the current is mostly distributed on the open microstrip lines and vertical segments of square ring, as depicted

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Fig. 10. Basic structure of meander coupled line.

Fig. 11. Modified quadruple-mode SRLR with meander coupled-line sections.

in Fig. 9(b) and (c). Thus, based on the analysis of different current distributions between the odd and even modes, some modifications on the horizontal segments of the square ring to interfere with the current path are considered to adjust the second odd-mode resonant frequency. B. Adjustment Design Using Meander SRLR As studied in [26], the meander coupled line is a slow-wave structure and exhibits the dispersion characteristic. The phase velocity on this line structure is frequency dependant and the dispersion relation presents a nonlinear curve. Thus, the frequency interval among two different electric lengths tends to be larger or smaller than the traditional linear one. Therefore, this property can be used to adjust the interval between the second odd mode and the second even mode. In addition, the effective resonant regions in four resonant modes are distinguishing, which is also crucial for the possibility of this adjustment. The basic structure of the meander coupled line is depicted in Fig. 10. As shown in this figure, the meander coupled line is described by three parameters: the number of meander units , and the length of the horizontal and vertical sections. Additionally, this configuration is utilizing space efficiently and results in size reduction. Based on the above discussion, the modified proposed RLR installed with the meander coupled line is depicted in Fig. 11. In this figure, the straight microstrip line sections located in the horizontal direction have been replaced by the meander coupled line. Also, two open microstrip lines of the proposed resonator are folded to a hairpin shape for further compactness. The frequency interval among the two second modes can then be adjusted by tuning the parameter , , and . Meanwhile, to compensate the variations of the other three frequencies that are caused from the inserted meander coupled line, and as two additional parameters are required as some minor adjustment. To speed up the design process, the parameterization and optimization tools of Sonnet are utilized. In addition, some dis-

continuous points, such as microstrip bends and T-junctions, need to be taken into consideration. Finally, the optimized physical dimensions of the meander RLR are as follows: , , , , and (unit: millimeters). The simulated of the meandering RLR, denoted via the blue dashed line (in the online version), is depicted in Fig. 8. As shown in this figure, four excited resonant modes, denoted as , , , and , are located at 2.45, 3.50, 5.21, and 5.79 GHz, respectively, which agree with the target specifications and prove the validity of the meander coupled-line technique on adjusting the second odd-mode resonant frequency. To clarify the desired quadruple-mode RLR, a design procedure is summarized as follows. 1) Based on the target specifications, compute the normalized even-mode resonant frequencies, i.e., and . The electrical lengths of and will be obtained from Fig. 6(a). 2) According to the known and , the other two normalized frequencies, i.e., and , can therefore be found in Fig. 6(b) and the corresponding odd-mode resonant frequencies of and can be determined. 3) The meander coupled line is utilized to adjust the second odd-mode resonant frequency to meet the design specifications. 4) Ascertain the physical dimensions of the RLR using the LineCalc Tool based on the known electrical lengths. The EM simulator is utilized to optimize the circuit layout according to the target specifications. C. Final Design Based on the proposed meander quadruple-mode RLR unit cell, a quad-band filter with a second-order Chebyshev frequency response and 0.1-dB ripple level is designed for WLAN and WIMAX potential applications. Fig. 12(a) depicts the layout of the designed quad-band filter. The right folded microstrip line of the RLR in Fig. 11 is folded along the horizontal direction. Thus, two cascaded RLRs can be coupled with the pseudo-interdigital coupling structure [27], which also miniaturizes the circuit size. This quad-band HTS filter is designed with the following design specifications: center frequencies and fractional bandwidths (FBWs) are GHz, GHz, GHz, GHz, , , , and , where the subscripts , , , and denote the first, second, third, and fourth passbands, respectively. Based on these target specifications, the initial physical dimensions of the quadruple-mode RLR can be firstly determined from Section III-A and listed as follows: , , , , and (unit: millimeters). The width of all microstrip lines is set to 0.2 mm. According to the discussion in [28], a multi-band filter can be equivalent to the design of several single-band filters independently, where each passbands is designed individually. The coupling scheme of the designed quad-band filter is depicted in Fig. 12(b). Thus, four passbands of the designed quad-band filter are designed individually based on the two pairs of resonant modes of the proposed RLR. The lumped circuit element values of the low-pass prototype filter are found to be


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Fig. 13. External quality factor versus the coupled-line length mm. (The solid line: mm and the dash line:

Fig. 12. (a) Layout of the designed quad-band HTS filter with the pseudo-interdigital structure. (b) Coupling scheme of (a).

, , , and . The external quality factors and coupling coefficients can be deduced as and for the first passband, and for the second passband, and for the third passband, and and for the fourth passband. The following step is to design the external coupling to meet the desired external quality factor . To provide more choices in designing the bandwidths for four passbands simultaneously, a high-impedance parallel-coupled microstrip line (PCML) is applied to design the input/output coupling structure. Shown in Fig. 12, the PCML has three design freedoms, such as line width , coupled length , and spacing . According to [29], the of the proposed filter can be extracted from the following expression: (15) and represent the resonant frequency and the where absolute bandwidth between the 90 points of the phase

when mm.)

response. The EM simulator Sonnet is used to extract the desired for the four passbands. Fig. 13 plots a design graph of for four passbands of the proposed quad-band filter. The width of the PCML, , is set to 0.1 mm. It can be observed from Fig. 13 that for four bands will be decreased as the coupled length increases or the coupled space narrows down. In order to satisfy the required for the four passbands simultaneously, the coupled length mm and spacing mm are determined from Fig. 13. Note that more design data can be obtained by choosing different combinations of the three parameters of the PCML. In spite of this, there exist some limitations for choosing the arbitrary bandwidths of four channels because three parameters are not enough to control four target values independently. The final step is to adjust the coupling lengths and spaces between the coupled resonators to meet the desired coupling coefficients for four passbands, respectively. Similar to the determination of , the coupling coefficients can also be extracted from the simulated -parameters of the filter. As described in [29], when two synchronously tuned coupled resonators have a close proximity, the coupling coefficient can be extracted from (16) and are the lower and higher dominant resonant where frequencies of the coupled resonant frequencies, respectively. Incorporating the current distributions in Fig. 9 and the actual simulation, it is found that the coupling between the open lines and the coupling between the vertical segments of the ring section of two RLRs mainly determine the coupling strength of four passbands. Therefore, two types of coupling structure are investigated, as depicted in Fig. 14. To realize the coupling independently, some modifications of Fig. 12 are carried out. The coupling length and distance of these two structures are indicated as , , , and , respectively. is also assumed for simplification. The corresponding simulated coupling curve, between the coupled resonators in Fig. 14(a) and (b), is shown in Fig. 15(a) and (b), respectively. It can be seen that there exist

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Fig. 14. Sketch of two types of resonator coupling structures. (a) Type I. (b) Type II.

Similarly, the coupling coefficients versus the coupling length with respect to three different is shown in Fig. 15(b). As shown, , , , and are enlarging as increases when is fixed, but for a determined , , , , and decrease as increases. While it is found that and are primarily control , but affect and less and have almost no effect on . Therefore, and can be determined to 0.75 and 2 mm based on the desired , respectively. In this case, the four coupling coefficients are 0, 0.005, 0.025, and 0.011, respectively. The final configuration of the designed HTS filter, shown in Fig. 12, contains two types of coupling that are discussed above simultaneously. Therefore, the total extracted coupling coefficients can be obtained from the superposition of the corresponding coupling coefficients in two coupling types [29]. By observing the resonant responses (both the magnitude and phase) of two coupling types, it is found that the coupling coefficients present the same “sign,” except for . Therefore, the sum of each coupling coefficient for each passband is 0.073, 0.068, 0.033, and 0.047, respectively. Finally, some adjustments of , , , and are conducted to satisfy the desired coupling coefficients of four passbands exactly. Following the abovementioned design steps, the coupled length and distance of the two RLRs are ultimately optimized by Sonnet as follows: , , , , , , , and (unit: millimeters). IV. FILTER IMPLEMENTATION AND VERIFICATION

Fig. 15. Simulated coupling coefficient as a function of coupling length under different coupling distance .

three groups of curves in each graph. Fig. 15(a) depicts the coupling coefficient as a function of coupling length on the condition of three different coupling distances . Herein, , , , and indicate the coupling coefficients between the identical modes of two coupled resonators, respectively. For the fixed , these four coupling coefficients ( , , , and ) are increase monotonously as enlarges. Meanwhile, the coupling coefficients are decrease as widens when keeps unchanged. Moreover, it can be observed that and mainly influence , , and , while they affect less from Fig. 15(a). Thus, based on the values of , , and , and can be preliminary determined to 0.4 and 3.5 mm. At this time, , , , and are obtained as 0.073, 0.063, 0.008, and 0.036, respectively.

For demonstration purposes, the designed second-order quadband HTS filter was fabricated on a double-sided YBCO films deposited on a 0.5-mm-thick MgO substrate with a loss tangent of 0.62E-5 at 77 K. The film is patterned by the standard photolithography. The ion etching technology is used to etch the front-side film to form the circuit structure, and the circuit is mounted on a gold-plated metal carrier and then carefully packaged into a shield box. Fig. 16 shows a photograph of the fabricated quad-band HTS filter with the cover opened. The overall size of the filter is 11.35 mm 6.5 mm (not include the feed lines), which amounts to ( is the guided wavelength of the 50line in the substrate at the center frequency of the first passband). Compared with some referenced quad-band filters, specifically listed in Table I, the designed HTS filters exhibits some superiority in circuit compactness, except for some circuits that are implemented with dual-plane/layered [1] or grounding structures [6]. The packaged superconducting filter was cooled down to a temperature of 77 K in a vacuum cooler and measured using an Agilent N5230A network analyzer. Calibration was done inside the cooler. The I/O cables and connectors inside the cooler were previously measured at both room and low temperatures so the effect of these cables and connectors was compensated. The red dashed lines, depicted in Fig. 17 (in online version), are the simulated results of the quad-band filter. It can be observed that the four passbands are centered at 2.43, 3.53, 5.18, and 5.78 GHz, respectively, which agree well with the design specifications. In addition, three transmission zeros shown in Fig. 17 are created at 1.96, 3.25, and 5.55 GHz, respectively, which enhance the skirt selectivity


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Fig. 16. Photograph of the fabricated HTS quad-band filter with metal cover opened.

TABLE I SIZE COMPARISON BETWEEN SOME OF THE REFERENCED QUAD-BAND FILTERS AND THE DESIGNED HTS FILTER

Fig. 18. Enlarged scale in-band. (a) First passband. (b) Second passband. (c) Third passband. (d) Fourth passband. -axis: -parameters (dB), legend: same as in Fig. 17.

1.94, 3.24, and 5.59 GHz are observed, which lead to more than 30-dB band-to-band isolation. It also greatly improves the steepness of the slope close to the passband and the out-of-band rejection with the level as high as 25 dB up to 8 GHz. Fig. 18 depicts an enlarged scale of insertion loss results in four passbands. From this figure, the measured insertion losses at center frequency of each passband can be observed and are approximately 0.12, 0.12, 0.23, and 0.25 dB, respectively, which demonstrates the advantage of insertion loss by using the HTS material. It will also greatly satisfy the potential requirement in high sensitivity of the future wireless communication. Simultaneously, the unloaded quality factors of the HTS filter in the four passbands are 1069, 1046, 1192, and 701, respectively. Some deviations between the measurements and simulations are mainly due to fabrication tolerance. Besides, a cryostat is indispensable to ensure the proper operation of this HTS filter. Therefore, cost reduction of the HTS device is an important research subject for the application in real WLAN and WIMAX systems in the future. Fig. 17. Simulated and measured frequency responses of the fabricated quadband HTS filter.

and isolation between the two passbands of the HTS filter. Among them, and can be explained by the mechanism described in Section II, while is produced by the pseudo-interdigital coupling structure [30]. The measured results are illustrated as the black solid lines in Fig. 17, which are in good agreement with simulations. The measured four passbands of the fabricated HTS filter are centered at 2.44, 3.53, 5.18, and 5.79 GHz and its corresponding 3-dB FBW are 4.96%, 5.07%, 2.32%, and 3.63%, respectively. The measured return losses are better than 20, 20, 15, and 16 dB, respectively. In addition, three transmission zeros located at

V. CONCLUSIONS A compact second-order quad-band HTS filter has been designed and realized in this paper. The quadruple-mode resonant characteristics of the SRLR have been investigated and its design graphs have been drawn to guide multi-band filter design. Furthermore, the meander coupled-line technique has been utilized to adjust the resonant frequency for achieving the desired specifications. Finally, the proposed superconducting filter has been constructed with two coupled quadruple-mode RLRs and the pseudo-interdigital coupling structure has been adopted to miniaturize the filter size. Three transmission zeros have been created due to the multipath propagation and sharp passband shirts have been observed. A satisfactory agreement between the measurements and simulations has verified the validity of the proposed design principle. The designed quad-band HTS filter

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with low insertion loss and compact size is very suitable for a multiband high-sensitivity wireless system. ACKNOWLEDGMENT The authors would like to express great appreciation to the editors and anonymous reviewers for their valuable comments and suggestions, which highly improved this paper. REFERENCES [1] L.-Y. Ren, “Quad-band bandpass filter based on dual-plane microstrip/DGS slot structure,” Electron. Lett., vol. 46, no. 10, pp. 691–692, May 2010. [2] C.-M. Cheng and C.-F. Yang, “Develop quad-band (1.57/2.45/3.5/5.2 GHz) bandpass filters on the ceramic substrate,” IEEE Microw. Wireless Compon. Lett., vol. 20, no. 5, pp. 268–270, May 2010. [3] J.-C. Liu, J.-W. Wang, B.-H. Zeng, and D.-C. Chang, “CPW-fed dualmode double-square-ring resonators for quad-band filters,” IEEE Microw. Wireless Compon. Lett., vol. 20, no. 3, pp. 142–144, Mar. 2010. [4] K.-W. Hsu and W.-H. Tu, “Compact wide-stopband quad-band bandpass filter with tunable transmission zeros,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2012, pp. 1–3. [5] H.-W. Wu and R.-Y. Yang, “A new quad-band bandpass filter using asymmetric stepped impedance resonators,” IEEE Microw. Wireless Compon. Lett., vol. 21, no. 4, pp. 203–205, Apr. 2011. [6] S.-C. Liu, “Microstrip dual/quad-band filters with coupled lines and quasi-lumped impedance inverters based on parallel-path transmission,” IEEE Trans. Microw. Theory Techn., vol. 59, no. 8, pp. 1937–1946, Aug. 2011. [7] J.-Y. Wu and W.-H. Tu, “Design of quad-band bandpass filter with multiple transmission zeros,” Electron. Lett., vol. 47, no. 8, pp. 502–503, Apr. 2011. [8] J. Xu, C. Miao, L. Cui, Y.-X. Ji, and W. Wu, “Compact high isolation quad-band bandpass filter using quad-mode resonator,” Electron. Lett., vol. 48, no. 1, pp. 28–30, Jan. 2012. [9] J. Xu, W. Wu, and C. Miao, “Compact microstrip dual-/tri-/quad-band bandpass filter using open stubs loaded shorted stepped-impedance resonator,” IEEE Trans. Microw. Theory Techn., vol. 61, no. 9, pp. 3187–3199, Sep. 2013. [10] J. S. Hong, E. P. McErlean, and B. M. Karyamapudi, “A high-temperature superconducting filter for future mobile telecommunication systems,” IEEE Trans. Microw. Theory Techn., vol. 53, no. 6, pp. 1976–1981, Jun. 2005. [11] L. Gao et al., “8-GHz narrowband high-temperature superconducting filter with high selectivity and flat group delay,” IEEE Trans. Microw. Theory Techn., vol. 57, no. 7, pp. 1767–1773, Jul. 2009. [12] A. M. Abu-Hudrouss, A. B. Jayyousi, and M. J. Lancaster, “Tripleband HTS filter using dual spiral resonators with capacitive-loading,” IEEE Trans. Appl. Supercond., vol. 18, no. 3, pp. 1728–1732, Sep. 2008. [13] L. Y. Ji, J. Ma, J. Sun, L. Wang, Y. Q. Li, and B. Liu, “Design and performance of dual-band high temperature superconducting filter,” Science China, vol. 55, no. 4, pp. 956–961, Apr. 2012. [14] L.-M. Wang et al., “Quarter-wavelength stepped-impedance YBCO superconducting filresonators for miniaturized dual-band highters,” IEEE Trans. Appl. Supercond., vol. 19, no. 3, pp. 895–898, Jun. 2009. [15] H. W. Liu et al., “Dual-band superconducting bandpass filter using embedded split ring resonator,” IEEE Trans. Appl. Supercond., vol. 23, no. 3, Jun. 2013, Art. ID 1300304. [16] Y. Heng et al., “Dual-band superconducting bandpass filter using stubloaded resonators with controllable coupling and feeding structures,” IEEE Microw. Wireless Compon. Lett., vol. 23, no. 8, pp. 400–402, Aug. 2013. [17] J. Z. Chen, N. Wang, Y. He, and C. H. Liang, “Fourth-order tri-band bandpass filter using square ring loaded resonators,” Electron. Lett., vol. 47, no. 15, pp. 858–859, Jul. 2011. [18] M. T. Doan, W. Q. Che, and W. J. Feng, “Tri-band bandpass filter using square ring short stub loaded resonator,” Electron. Lett., vol. 48, no. 2, pp. 106–107, Jul. 2012. [19] H. W. Liu, B. P. Ren, X. H. Guan, J. H. Lei, and S. Li, “Compact dual-band bandpass filter using quadruple-mode square ring loaded resonator (SRLR),” IEEE Microw. Wireless Compon. Lett., vol. 23, no. 4, pp. 181–183, Apr. 2013.

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Haiwen Liu (M’04–SM’13) received the B.S. degree in electronic system and M.S. degree in radio physics from Wuhan University, Wuhan, China, in 1997 and 2000, respectively, and the Ph.D. degree in microwave engineering from Shanghai Jiao Tong University, Shanghai, China, in 2004. From 2004 to 2006, he was with Waseda University, Kitakyushu, Japan, as a Research Assistant Professor. From 2006 to 2007, he was a Research Fellow with Kiel University, Kiel, Germany. From 2007 to 2008, he was a Professor with the Institute of Optics and Electronics, Chengdu, China. Since 2009, he has been a Chair Professor with East China Jiaotong University, Nanchang, China. He has authored or coauthored over 100 papers in international and domestic journals and conferences. His current research interests include electromagnetic modeling of high-temperature superconducting circuits, RF and microwave passive circuits and systems, synthesis theory and practices of microwave filters and devices, antennas for wireless terminals, and radar systems. Dr. Liu has served as a Technical Program Committee member for many international conferences. He has been a reviewer for international journals, including the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, the IEEE TRANSACTIONS ON APPLIED SUPERCONDUCTIVITY, the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, the IEEE TRANSACTIONS ON COMPONENTS, PACKAGING, AND MANUFACTURING TECHNOLOGY, IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, and IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS. He was the recipient of the Alexander von Humboldt Research Fellowship while with Kiel University. He was supported by the Chinese Academy of Sciences under the 100 Talents Program while with the Institute of Optics and Electronics. He was a recipient of Wang Kuancheng Science Foundation Award in 2008, the Best Paper Prize of the 2005 International Conference on Communications, Circuits and Systems Proceedings in Hong Kong, the Prize of the Osaka City Mayor for the Conference of Chinese Alumni in Japan in 2005, the 100 Best Ph.D. Dissertations in Shanghai, China, in 2006, the National Distinguished Ph.D. Student Scholarship, China, in 2003, the National First-Class Guanghua Education Scholarship, China, in 2002, and the Huawei Company Scholarship, China, in 1999.


Baoping Ren was born in Ganzhou, China, in 1988. He received the B.S. degree in communication engineering and M.S. degree in communication and information system from East China Jiaotong University, Nanchang, China, in 2011 and 2014, respectively. He is currently the Academic Secretary for the Jiangxi Province Key Laboratory of RF Communications and Sensor Networks, East China Jiaotong University. His current research interests include RF and microwave passive components and systems, high-temperature superconducting circuits, and metamaterials and all their applications.

Xuehui Guan (M’11) received the B.S. degree in communication engineering from Jiangxi Normal University, Nanchang, China, in 1998, and the Ph.D. degree in electromagnetic fields and microwave techniques from Shanghai University, Shanghai, China, in 2007. Since June 2007, he has been an Associate Professor with the School of Information Engineering, East China Jiaotong University, Nanchang, China. In 2012, he was a Senior Researcher Associate with the School of Electrical and Electronic Engineering, City University of Hong Kong, Kowloon, Hong Kong. Since June 2013, he has been a Visiting Scholar with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore. His current research interests include microwave passive circuits and systems, high-temperature superconducting circuits, substrate integrated waveguide (SIW) components, synthesis technique of microwave filters, and planar antennas for wireless communications.

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Pin Wen was born in Nanchang, China, in 1987. He received the B.S. degree in communication engineering from East China Jiaotong University, Nanchang, China, in 2012, and is currently working toward the M.S. degree at East China Jiaotong University. His current research interests include antenna theory and design and superconducting filter design.

Yan Wang was born in Benbu, China, in 1989. She received the B.S. degree in communication engineering and M.S. degree in communication and information system from East China Jiaotong University, Nanchang, China, in 2011 and 2014, respectively. She is currently an Engineer with Railway Communications, Railway Design Institute, Shanghai, China.