The pulse generator is based on a Class-S digital pulse amplifier and a Step. Recover Diode circuit. The antenna subsystem is based on a bow- tie topology.
QUETZAL: Qualified Ultra-wideband Testbed For Reduced Data-rates And Location A. Molfulleda, M. N´ajar, P. Miskovsky, J. A. Leyva, L. Berenguer, C. Ibars and M. Navarro
Publication: Vol.: No.: Date:
in Proc. TRIDENTCOM 2006, Barcelona (Spain) March 1-3 2006
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QUETZAL: Qualified Ultra-wideband Testbed For Reduced Data-rates And Location A.
Mollfulleda1, M. Nájar1,2, P. Miskovsky1,2, J. A. Leyva1, L. Berenguer3, C. Ibars1, M. Navarro1 e-mails: {antonio.mollfulleda, montse.najar, pavel.miskovsky, joananton, lluis.berenguer, Christian.ibars,monica.Navarro}@cttc.es
1
Centre Tecnològic de Telecomunicacions de Catalunya; Av. Del Canal Olímpc s/n, 08860 Castelldefels, Bcn (Spain) Universitat Politècnica de Catalunya (UPC); C/Jordi Girona 1-3, D5, Campus Nord, 08034 Barcelona (Spain), 3 ITOWA Investigation Total Ware; C/Faraday 159, 08224 Terrasa Barcelona (Spain)
2
Abstract—This work presents an UWB testbed for reduced data rates with capabilities of measuring range and location. The transmitter uses Time Hopping spread spectrum codes to reduce the peak to average ratio in the power spectral density. The pulse generator is based on a Class-S digital pulse amplifier and a Step Recover Diode circuit. The antenna subsystem is based on a bowtie topology. The approach for the receiver is based on a filter bank reducing the complexity of the traditional RAKE receiver. Normalized Minimum Variance algorithm is proposed for estimating Time of Arrival exploiting the transmitted pulse train. Positioning estimation is tackled with the Extended Kalman Filter with TOA bias tracking allowing high accuracy even in Non Line of Sight scenarios. Index Terms—Ultra-wideband, step recovery diode, class-S amplifier, bow-tie antenna, STFT, filter bank, location, testbed
I. INTRODUCTION
I
mpulse Radio (IR) techniques have been widely used over the last three decades [1]. Time-domain reflectometry, radar, imaging and military systems have covered most of the attention of these techniques. Recently, the Federal Communications Commission (FCC) has released the UWB emission spectral mask [2], limiting the radiated power in the band between 3.1 GHz and 10.6 GHz. Following this report, a number of emerging commercial applications have revived significant research and development of techniques for generating narrow pulses and UWB signals. The transmission quality of any communication system is determined by the received signal to noise ratio (SNR), which in FCC compliant UWB systems, is strongly related to the efficient usage of the spectrum mask. The main problem of the transmission of IR signals are spectral lines [1], which force to reduce the transmitted power in order to keep the radiated signal well bellow the mask limitations. Using spreading codes in which more than one pulse per symbol is transmitted the peak level of the spectral lines is reduced. Two types of codes can be This work is partially supported by the European Union under project IST-2002-507525 (NEWCOM), by the Spanish Government under contracts FIT-330200-2004-281 (PLANETS MEDEA+), FIT-330201-2004-11 (QUETZAL) and TEC2005-08122-C03-01 (jointly financed by FEDER) and by the Catalan Government SGR 2005-00996 and SGR2005-00690.
1-4244-0106-2/06/$20.00 ©2006 IEEE
used in traditional IR systems: time-hopping spread spectrum (TH-SS), where codes randomize the pulse position inside a TH interval; and direct sequence spread spectrum (DS-SS), where the pulse sign is randomized. With respect to the receiver, the traditional approach for multi-path channels is the RAKE receiver. In dense multi-path environments this solution captures only a small fraction of the channel energy. A filterbank approach is presented as a solution that captures more channel energy, improving the performance of the RAKE receiver. Besides transmission quality issues, one of the main advantages of UWB technology is the potential accuracy that can be achieved in applications that require an estimation of the location, a consequence of the short transmitted pulse duration. Determining the location of a User Equipment (UE) consists of two steps: estimation of signal measurements and position computation based on the measurements. The first step involves estimation of the signal parameters from the received signal: time of arrival (TOA), time difference of arrival (TDOA) or angle of arrival (AOA). The second step combines multiple timing or angles of arrival measured from a convenient number of Base Stations (BS). There are two primary reasons for inaccuracies observed in location systems, due to the propagation conditions in the wireless channel: multipath propagation, and non-line-of-sight (NLOS) condition. Both situations yield a biased estimation of the first arrival, which bears information related to the position of the mobile terminal. In particular, due to NLOS, the first arrival suffers stronger attenuation than later arrivals and therefore wrong timing information is obtained. In the multipath situation, late signal arrivals induce an offset of the maximum of the impulse response, thus biasing timing estimates, if timing estimation algorithms, which assume a simple frequency-flat channel are used. The resulting positioning systems can only provide biased position estimation. The scheme proposed in [2], consists in deriving high-resolution timing estimation using the parametric Normalized Minimum Variance (NMV) over the Discrete Fourier Transform (DFT) of the channel estimates. In this way, the first low power estimated arrivals are robustly estimated in severe multipath situations. In this paper we present the architecture of an UWB testbed that captures all channel energy in order to evaluate the data transmission and location capabilities of an UWB system. The
IR transmitter presented in Section II uses TH-SS and DS-SS codes to reduce the effects of the spectral lines. The design of the antenna subsystem is also described. Next, the receiver architecture, which is based on a filterbank, is shown. Section IV is devoted to location estimation; the TOA estimation algorithm is briefly presented and the position computation is carried out by the Kalman tracker, which deals with the bias of the measures caused by the NLOS situation. The paper ends with the conclusions obtained from the testbed design.
through those capacitors introduces additional current, which contributes to the rapid removal of stored charge in the PNjunction of the transistors. The RC circuits formed by R1C1 and R2C2 form a low pass filter that limits the maximum Pulse Repetition Frequency (PRF). The constant time of these RC networks must be RC 1 10 f , where fp is the maximum achievable PRF. Note that the output impedance of the pulse amplifier is 50:, which is required by the SRD pulse generator. Finally, to work at frequencies up to 40MHz, conventional RF transistors have been used. p
II. TRANSMITTER ARCHITECTURE The complete impulse radio transmitter consists of a digital device such as CPLD or FPGA, two digital pulse amplifiers (DPA), two identical Gaussian monocycle pulse generators, one for positive pulses and one for negative pulses, an inverter/noninverter block, a power combiner, and the antenna. Figure 1 shows a block diagram of the IR transmitter. Based on the data bits, received through a serial port, the digital device performs PPM modulation and applies TH-SS codes. The DS-SS code selects one of the two channels for pulse generation. The DPA drives the pulse generators, which are based on Step Recovery Diodes (SRD). The inverter/non-inverter block changes the pulse sign in one of the fingers, so that in one channel positive pulses are generated whereas in the other pulses are negative. Finally the power combiner joints both signal paths to feed the antenna. Inverter/ Non-Inverter DPA
Pulse Generator
x(1) Power Combiner
CPLD DPA
Pulse Generator
x(-1)
Figure 1. Block diagram of IR transmitter using TH/DS-SS codes.
For the pulse generators, a Gaussian monocycle is preferred as pulse waveform because its spectrum can be easily fitted below the FCC mask. The Pulse Repetition Frequency (PRF) of conventional SRD circuits is normally limited to 10 MHz [3][4]. This fact limits the application of SRD technology to UWB, since the PRF required to reach the PSD specified by the FCC mask for 3-Volt pulses is about 60MHz. In this section we present a pulse generator capable to operate at up to 40 MHz. To achieve this operating frequency, a high-speed digital pulse amplifier is required to drive enough current for the SRD circuit. A. Digital Pulse Amplifier The digital pulse amplifier is based on a Class-S amplifier driving a medium power bipolar transistor (Q3), which acts as a switch [6]. The latter is the responsible for generating enough current to drive the SRD pulse generator. Figure 2 shows the circuit of the pulse amplifier. Transistors Q1 and Q2 are configured to form a two-position switch. The complementary pair Q1 and Q2 ensures rapid switching and adequate voltage to maintain the ON/OFF state of the output transistor Q3. The Zener diode Z1 and base resistance are used for accommodating the voltage levels in DC mode. The base capacitors are introduced to speed-up the switching time. The voltage variation
Vcc
C1
Q1 Z1
R1
Vcc R3 Q3
C2 Q2 R2
50
-Vcc
Figure 2. Digital Pulse Amplifier.
B. SRD Pulse Generation The key property of a SRD diode to generate sub-nanosecond falling or raising edges is the very fast switching from forward to reverse modes [3]. While the diode is forwarded-polarized, electric charge is stored in the PN-junction of the diode. The total stored charge depends on the average current and the recombination time of the diode according to QF
I F W
(1)
where QF is the stored charge in the forward mode IF is the average current flowing through the PN-junction and W is the recombination time of the diode. Figure 3 shows the circuit configuration for Gaussian monocycle pulse sharpening using SRD diodes. At the beginning of operation, the voltage source Vin is set to ground level. In this mode the SRD-1 operates in forward polarization through the bias circuit. The total stored charge in the PNjunction is given by (1). When the voltage source Vin rises (it can take a few nanoseconds), reverse current flows through the diode removing the stored charge. During this discharge period the diode keeps the low impedance state and consequently its voltage remains close to ground level. When the stored charge is completely removed, the SRD-1 switches abruptly to a highimpedance state. In this moment the input voltage is transmitted to the load generating a rising edge of duration equal to the switching time of the diode, which can be lower than 100ps. In the same way that SRD-1 is used to configure the rising edge of the pulse, SRD-2 is used to generate the falling edge of a Gaussian pulse. Whereas SRD-1 is operating on low impedance state, SRD-2 operates in forward polarization through the bias circuit and the short-circuited stub. Again electric charge is stored in PN-Junction of SRD-2. In the moment in which SRD-1 switches to a high impedance state, the rising edge is propagated to SRD-2. During the rise time, reverse current flows through SRD-2 removing the stored charge, which switches to high impedance state when the charge
is removed. This causes the falling edge of the Gaussian pulse. Following the Gaussian pulse generator, a passive shapingnetwork formed by a short-circuited stub is used to generate a Gaussian monocycle pulse [4]. The impulse response of the shaping network is hSN t 0.5G t 0.5G t W P where WP must be equal to one half of the pulse width. To obtain the optimum Gaussian monocycle in terms of amplitude, WP should be the half of the pulse coming from SRD-2. Besides shaping the monocycle waveform, the shaping network generates a reflected wave, which is propagated through SRD-2 back to SRD-1. As a result, additional pulses appear in the output port. To solve this problem a 6dB-attenuator has been placed between the diodes so that this reflection is attenuated to an appropriate level. The result is a low ripple Gaussian monocycle pulse generator.
switching time of about 70ps. Figure 5 shows a picture of the pulse generator. In Figure 6 a measurement of the positive and negative pulses is depicted. The pulse width is 185ps, whereas the peak-to-peak amplitude is 2.2V. Contrasting simulations and measurements, it has been found that the unbalanced response of the generated pulses is caused by the inductive effect of the SRD package.
Figure 5. Picture of the developed transmitter. L2
L1
TL Zs
50: SRD-2 Stub
SRD-1 Vin
6dB Attenuator
RL
The measured spectrum using TH-SS and DS-SS codes is depicted in Figure 7. The peak-to-average ratio measured in the generated PSD is reduced to 10 dB. Note that the 10-dB bandwidth of the generated pulse is about 9GHz. 1.2
Inverted Pulse Non-Inverted Pulse
1 0.8
Figure 3. SRD Gaussian pulse generation circuit.
Bottom view
Amplitude (mV)
C. Inverter / Non- Inverter Circuit The key point of the inverter block lies in the transitions between balanced and unbalanced propagation modes. The pulse inversion is achieved by introducing a transition between unbalanced to balanced mode and, subsequently,, a transition from balanced to unbalanced but changing the reference plane at the output. This idea was introduced in [5] using transitions between microstrip and coplanar slotlines. However, the application of these transitions is limited to pulses of 1ns width. Recently, in UWB transitions between microstrip and DoubledSided Parallel-Strip have been developed [8], which allows the inversion of UWB pulses. The Inverter/Non-Inverter circuit has two different paths: one performs the pulse inversion whereas the other is the NonInverter, as shown in Figure 4. The Non-Inverter branch line has the same length as the Inverter branch. This is necessary to maintain both ways synchronized and same strip losses.
0.6 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 -1 -1.2 -300
-200
-100
0
100
200
300
400
500
time(ps)
Figure 6. Measurement of pulse generation (185 ps and 2.2 V).
Top view
Figure 7. Generated PSD using TH-SS and DS-SS codes.
Figure 4. Inverter / non-inverter Scheme.
D. Measurement Results on Pulse Generation The whole pulse generator has been fabricated in microstrip technology using a TACONIC RF30-0300 substrate, with 0.76mm thickness and a relative permittivity Hr =3. The SRD diodes are from Micrometrics MSD700-19-1, which have a
E. Antenna Design The main concern of the antenna subsystem is to achieve a small size antenna having omni-directional radiation pattern and covering the FCC UWB mask frequency range. The antenna subsystem is integrated in both, transmitter and receiver circuits. To meet the first requirement, the use of planar technology is almost mandatory. Directivity and frequency range requirements might be fulfilled by properly choosing the antenna topology. The testbed implements a modified bow-tie topology developed on double-sided technology, which is preferred for
alternative receiver architecture based on analog, quasiorthogonal filter banks has been studied. 0
simulated measured
-5
S11 [dB]
balanced feeding. The geometry consists in rounding the perimeter of a conventional 90º bow-tie antenna, which leads to a wider bandwidth [7]. The size of the whole antenna and the size of the feeding point define the lower and upper frequency limits of operation, respectively. The second part of the antenna subsystem is the matching network. We have considered a binomial transformer consisting of seven cascaded quarter-wavelength sections at 6.85 GHz center frequency. In the inset of Figure 5 we can also see the binomial transformer. Note that the matching network is also developed using doubled sided parallel striplines. To connect the antenna subsystem to the previous stage of the UWB system, the transition between double-sided and microstrip is also used. The antenna subsystem has been also fabricated on a TACONIC RF30-0300 substrate. The inset of Figure 5 shows the antenna subsystem. Previously, its performance was simulated using conventional software for electromagnetic analysis and design using FDTD. Figure 8 shows simulated and measured results for the input return losses magnitude. Simulation results predict return loss figures lower than –9.9dB within the 3.1GHz – 10.9GHz band. However, measurements show return losses below 11.7dB within the 3.6 GHz-10.9 GHz frequency range. Although this bandwidth is narrower than expected, measured input losses have been improved by 2dB. Also, co-polar and cross-polar radiation patterns at 3 GHz, 5 GHz and 9 GHz are shown in Figure 9. As seen from this figure, at 9 GHz the co-polar radiation is dominant at low frequencies (3 and 5 GHz), and almost equal to the cross-polar radiation at higher frequencies (9 GHz). As expected the radiation pattern suffers some distortion at different frequencies, similar to classical dipole structures, but globally preserves good radiation characteristics. It has been verified that the use of double-sided stripline for antenna feeding gives rise to low cross-polar radiation at higher frequencies [9], which should not cause problems as long as the same antenna is used at the receiver side. One of the testbed activities consist on verifying the performance of this antenna in a real environment. This will be done by a set of measurements placing the receiver antenna in different positions.
-10 -15 -20 -25 -30 0
1
2
3
4
5
6
7
8
9
10 11
frequency [GHz] Figure 8. Antenna, picture and measurement results.
Figure 9. Radiation pattern, co-polar (solid), cross-polar (dotted)
The proposed receiver architecture is shown in Figure 10. The received signal is split into M branches and filtered by a bank of filters with impulse responses h1* t , h2* t , L hM* t . The signal is then sampled at t=T. Mathematically, we express the discrete-time signal as ci { x t , h t
³ x t h t dt , i * i
1, 2, L , M
(3)
If the filter bank is an orthonormal basis of the signal space where x t lies in, then the latter may be represented as x t
M
¦ c h t i i
(4)
i 1
For practical purposes, we wish to represent the received signal with as few coefficients as possible, using filters h1 t , h2 t , L , hM t that can be easily implemented with analog hardware. Possible orthogonal or nearly orthogonal filter bank implementations are the Short Time Fourier Transform (STFT) and the Discrete Wavelet Transform (DWT) [11].
III. RECEIVER ARCHITECTURE The traditional receiver for spread spectrum signals in multipath environments is the Rake receiver, which collects the energy of N signal paths, N being the number of Rake fingers. The amount of energy captured by the Rake receiver greatly depends on the amplitude distribution of the incoming paths. For outdoor UWB applications, the received energy is spread among a large number of paths with significant energy, especially in non-line of sight environments [10], making the Rake approach rather impractical. Moreover, implementation issues such as the need for very accurate synchronization and for the availability of a pulse template at the receiver constitute additional disadvantages of this approach. In view of the impairments of the Rake receiver, an
Figure 10. Filter bank UWB receiver.
For a window function J t , the STFT is given by f
STFT nT , lB
³ x t J t dt * n ,l
f
(5)
where J n,l t J t nT e j 2S lBt is a translated and modulated version of the window function. The STFT is characterized by having a constant time and frequency resolution, given by the window function. In terms of implementation issues, bandpass filters J n ,l t may be difficult to implement for higher frequencies since they require a higher fractional bandwidth. A filter bank receiver based on calculating the pulse Fourier coefficients was proposed in [12]. It is easily seen that such receiver allows an interpretation in terms of the STFT. Given a bandpass basis function h t , the DWT is given by f
WT a, l
³ x t h t dt * a ,l
(6)
f
where h t 1 a h t l a is a scaled and translated version of the basis function. Unlike the STFT, the DWT has variable time and frequency resolution, and its implementation can be done with constant Q filters. a,l
IV.
LOCATION
A. TOA Estimation The NMV TOA estimation exhibits a good performance in bias and variance compared with the TOA estimation based only on the maximum of the channel response estimate. The frequency domain signal observations at time n=1…N, from a channel composed of L multipath arrivals follows the model:
ª¬ y wo ; n
L
¦ a n PeW l
l
y w1 ; n L
y wM 1 ; n º¼
T
The state vector s’(k) and the state matrix D are defined as, ªs k º ªA 0 º sc k « » D « » ¬ 0 B¼ ¬b ( k ) ¼ where the components of the vector s(k) represent the position of the mobile terminal and the speed in two-dimensional Cartesian co-ordinates at discrete time k and the components of the vector b(k) are the time measurement bias for each BS. A is the state matrix defined for continuous movement [15] and B E I is the state matrix which defines the time measurement bias at each BS as a random walk [16]. The disturbance transition vector is defined as T w k > 0 w v w b @ , where w v and w b are the speed and bias noise vectors with covariance matrix Q v (k ) and Q b ( k ) , respectively. TOA measurements are non-linear in the position state variables, z k
g s k b k v k
v n
l 1
where al is the amplitude of the l-th propagation path, eW l is the steering vector for an arrival at timing W l , P is a diagonal matrix containing the DFT of the received pulse and v n is the noise in the estimated channel. The timing estimation of all the multipath channel rays may be computed as the maxima of the expression [13]: f H R 1f S W WH 2 W fW R fW where fW PeW and R ¦ y n y n is the correlation matrix of the estimated channel. Normally, the first maximum in (8) is selected as the timing bearing position information. In IR UWB communications a single data symbol is associated to several consecutive pulses, each located in its own frame. The redundant information introduced by the pulse repetition permits a robust estimation of the first TOA as the minimum of the TOAs estimated at each frame. The maximum number of available pulses for the parameter estimation depends on the channel time coherence, defined as the expected period of time over which the channel response is essentially invariant. Figures 11 and 12 compare the Root Mean Square Error (RMSE) and the standard deviation of the TOA estimation considering one and five pulses in LOS and NLOS, respectively. N
H
n 1
where z k is the measurement vector obtained from the different BS and v k is the measurement noise with diagonal covariance matrix C(k ) , assuming that the measures are uncorrelated. 4.5 Multiple pulses (RMSE) One pulse (RMSE)
4 3.5
Error (m)
y n
B. Positioning Estimation The Extended Kalman Filter (EKF) proposed for location purposes allows tracking the position and speed of the mobile, yielding an accurate prediction algorithm. Also the tracking of the bias due to NLOS is possible by increasing the dimension of the state vector by adding TOA bias as additional parameters to be estimated [14]. The transition equation defined for continuous movement, is linear: sc k 1 Dsc k w k
Multiple pulses (RMSE + Std deviation) One pulse (RMSE + Std deviation)
3 2.5 2 1.5 1 0.5 00
5
10
15
20
25
Eb/No (dB) Figure 11. Ranging error in LOS.
30
Figure 14. Tracking (Eb/No=25)
14
10
Multiple pulses (RMSE + Std deviation)
Error (m)
12
Multiple pulses (RMSE) One pulse (RMSE)
V. CONCLUSIONS
One pulse (RMSE + Std deviation)
8 6 4
2 00
5
10
15
20
25
30
Eb/No (dB) Figure 12. Ranging error in NLOS.
For positioning, the scenario considered for simulations is a typical indoor office with 5 base stations with one sensor each one. The trajectory is a walk trajectory with speed around 5 Km/h over an area of 30 meters. The ranging is estimated from the minimum NMV TOA corresponding to 5 impulse responses. This ranging estimation is biased because of the NLOS condition and the multipath channel. After the ranging estimation, the Extended Kalman filter combines all measurements and makes accurate estimation of the mobile location. Figure 13 shows the results of the classical Kalman filter, which track the position and the speed, and the proposed one, which track the position, the speed, and the TOA, bias in each base station. It can be observed that a reduction of the positioning error is achieved with bias tracking. The results obtained prove that UWB can offer good accuracy in indoor geolocation. Figure 14 shows the simulated scenario with the Base Stations positions, the real trajectory and the tracking with the proposed Kalman filter with bias tracking.
The implementation of an ultra-wideband testbed for low-rate communications, ranging measurements, and positioning has been described. The main concern of the testbed is to provide a platform capable of capturing all channel energy in order to estimate location capabilities of IR UWB systems. This testbed is being developed in the context of recent FCC regulation. Increased transmission range is achieved by maximizing the transmitted power below the mask limitations. To this end, the transmitter uses TH-SS codes in order to reduce the spectral lines caused by an impulse radio signal. The developed transmitter uses an SRD-based circuit and a passive shaping network to generate a Gaussian monocycle. The measured pulse width is about 200ps. A Class-S digital pulse amplifier using conventional RF transistors has been designed to drive SRD pulse generator, which can work at up to 40MHz clock frequency. The designed antenna subsystem is based on a bowtie topology, which is shown to have a good radiation pattern. The matching network to feed the antenna uses a seven-section binomial transformer. Measurement results show a good matching in the whole frequency band. With respect to the receiver, we consider a filter bank approach, which reduces the complexity of a traditional RAKE receiver and can potentially capture all channel energy. Further research on this type of receiver is required before the final implementation. Ranging based on NMV TOA estimation, exploiting the pulse train defined in the IR transmission, has been evaluated in LOS and NLOS scenarios. From ranging measurements, the proposed EKF with TOA bias tracking improves significantly the performance of the location estimate in NLOS scenarios. ACKNOWLEDGMENT
9
8 7
Error (m)
The authors wish to acknowledge to Jose González de Abestru and Jordi Romeu for their help and comments on the antenna design and measurements.
Kalman RMSE Kalman- bias RMSE Kalman RMSE + Std deviation Kalman-bias RMSE + Std dev
6 5
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Eb/No (dB) Figure 13. Positioning error
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