Single-Layer Wideband Circularly Polarized Patch ... - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

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Single-Layer Wideband Circularly Polarized Patch Antennas for Q-Band Applications Tao Zhang, Yan Zhang, Lina Cao, Wei Hong, and Ke Wu

Abstract—In this communication, dual-fed wideband circularly polarized (CP) patch antennas based on the single-layer printed circuit board (PCB) process and the substrate integrated waveguide (SIW) technology are proposed. Firstly, a patch antenna fed by an SIW coupler is developed to achieve a measured 3 dB axial-ratio (AR) bandwidth of 21% (from 38 GHz to 47 GHz). Then, by introducing second-order inductive windows, an improved CP patch antenna is developed. The measured results show that the 1 dB gain bandwidth is expanded from 3 GHz to 9 GHz while the overlapped bandwidth of AR and reflect coefficient is maintained. Finally, a 2 2 CP array is developed with a dual-polarized patch array. By carefully selecting the resonant frequency of the patches, a wide gain bandwidth is achieved. Moreover, by compensating the major error that deteriorates the AR bandwidth with a specially-designed coupler, the measured 3 dB AR bandwidth reaches as wide as 22.8%. The measured peak gain is 12.35 dBi. Index Terms—Dual-polarized array, patch antenna, substrate integrated waveguide, wideband circularly polarized.

I. INTRODUCTION In 2010, a millimeter wave communication standard named Q-LINKPAN was launched in China [1]–[3], which includes both the short-range high data rate communications (PAN) and long-range high data rate communications (LINK). In 2013, the spectrum allocation for Q-LINKPAN was issued by the Ministry of Industry and Information Technology of the People's Republic of China with 5.9 GHz (42.3 GHz–47 GHz, 47.2 GHz–48.4 GHz) for PAN and 3.6 GHz (40.5 GHz–42.3 GHz, 48.4 GHz–50.2 GHz) for LINK. For the LINK application of Q-LINKPAN, wideband circularly polarized (CP) antennas are usually used for polarization diversity. However, to design a low cost, wideband millimeter wave CP antenna is a challenge. In [4]–[7], the travelling-wave antennas [4], [5], the wideband circular polarizer [6] and the metal-topped via fence [7] are introduced to achieve wide AR bandwidths. Furthermore, the sequential rotation (SR) technique [8], [9] and the dual fed method [10] have been introduced to enhance the AR bandwidth. However, structural complexity makes all the above designs difficult to implement on a single-layer printed circuit board (PCB). In this communication, the dual fed technique is adopted to achieve wideband AR bandwidth based on the single-layer PCB process. Moreover, in order to suppress loss at millimeter wave bands, the SIW technology is employed. For the element design, a patch antenna fed by an SIW coupler is developed, which achieves an AR bandwidth of 21%. However, the gain bandwidth is narrow. To resolve this issue, second-order inductive Manuscript received February 16, 2014; revised August 06, 2014; accepted October 20, 2014. Date of publication October 24, 2014; date of current version December 31, 2014. This work was supported in part by the 863 High Tech. Project of China under Grant 2012AA01A50602, by the NSFC of China under Grant 61302019, and in part by the NNISRP (No. 201110046). T. Zhang, Y. Zhang, L. Cao, and W. Hong are with the State Key Laboratory of Millimeter Waves, Southeast University, Nanjing 210096, China (e-mail: [email protected]; [email protected]; [email protected]). K. Wu is with the Poly-Grames Research Center, Department of Electrical Engineering, École Polytechnique de Montréal, Montréal, QC H3C3A7, Canada (e-mail: [email protected]). Color versions of one or more of the figures in this communication are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2014.2365036

Fig. 1. Geometry of the original CP element. (a) Top view of the overall structure. (b) Detailed view of the SIW coupler. (c) Detailed bottom view of the CPW , , , , to SIW transition: , , , , , , , , , , , , , , (all in mm).

windows are introduced to the developed element, which expands the 1 dB gain bandwidth from 7% to 21%. For the array design, due to the large width of SIW, the dual-fed technique in [10] and the SR technique [8], [9] are difficult to realize on a single layer PCB. In this case, the dual-polarized patch array with short feed lines and simple feed layout in [11] is adopted. However, no bandwidth information is presented in [11] when the array is used as a CP one. This communication reveals its bandwidth property and presents a wideband CP array. First, the two modes of the two-element series-fed array are analyzed, which leads to a wide CP gain bandwidth of 18.2%. Then, by analyzing and compensating the errors of the orthogonal radiating fields, the measured 3 dB AR bandwidth reaches 22.8%. The peak gain of the array is 12.35 dBi. II. ANTENNA ELEMENTS The substrate is Rogers 5880 (permittivity: 2.2, loss tangent: 0.0009, thickness: 0.508 mm). The width of the feed SIW is selected based on mode transmission region [12], and is selected as 3.75 mm. the A. The Original Antenna Element Fig. 1 shows the geometry of the original CP antenna element, which combines the conventional dual-fed technique and the SIW technology. As shown in Fig. 1(a), a 90 SIW coupler is adopted to orthogonally excite the square patch antenna. When Port 1 is excited and the other port is matched, the antenna will generate left-hand CP waves. For the convenience of measurement, the two ports of the coupler are bended to opposite sides of the substrate, which leads to larger size. Moreover, the SIW to co-planar waveguide (CPW) transition [13] is adopted to

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Fig. 2. Axial ratio and T/R efficiency of the original CP antenna element.

Fig. 3. Reflect coefficient and boresight gain of original CP antenna element.

fix the coaxial connector (not shown). The detailed geometry of the transition is shown in Fig. 1(c). As shown in Fig. 1(a), the quarter-wave to alleviate the matching line is filleted with a filleting radius of parasitic radiation. As demonstrated in Fig. 2, the measured 3 dB AR bandwidth reaches 21% (38 GHz–47 GHz). Fig. 3 shows the reflect coefficient and boresight gain. Due to the limitation of measurement facilities, the magnitudes (in dB) of the two orthogonal radiating fields are measured with linearly polarized horn antenna and then added together to estimate the radiation patterns and gains [7]. The same measurement technique is used for the following two antennas. It can be seen that the measured reflect coefficient is below 10 dB from 37 GHz to 48.8 GHz. However, the gain bandwidth is relatively narrow, which could be explained as follows. Rather than return to the exited port, the return wave from the patch travels to the other port and gets absorbed by the loading resistor, which leads to gain sacrifice. Thus, the gain bandwidth is restricted by the narrow bandwidth of the patch. To explain this issue more intuitively, the transmission/reflection T/R efficiency adopted in [14], defined as the ratio of the total power consumed in the antenna system (excluding the matching load) to the input power , is displayed in Fig. 2. It could be observed that away from the center frequency, the T/R efficiency drops severely, which leads to the narrow gain bandwidth. Due to the loss in the antenna system, the measured T/R efficiency is generally larger than simulation. The same reason leads to similar phenomenon in the measurement of the following two antennas.

Fig. 4. Radiation patterns of the original CP antenna element. (a) in xoz plane, at 40 GHz, (b) in yoz plane, at 40 GHz, (c) in xoz plane, at 42 GHz, (d) in yoz plane, at 42 GHz, (e) in xoz plane, at 46 GHz, (f) in yoz plane, at 46 GHz.

In Fig. 4, the measured and simulated radiation patterns are presented. It can be seen that the patterns are not axial or symmetric, which is caused by the parasitic radiation from the substrate modes and the microstrips. This issue is especially severe for a single CP element at such high frequencies. The same issue also deteriorates the patterns of the improved antenna element in the following part. B. The Improved Antenna Element In this part, the bandwidth of the patch antenna is expanded by second-order inductive windows to improve the gain bandwidth of the CP antenna. The full-wave model and its equivalent circuit are shown in Fig. 5. The tuning procedure with the circuit model based on Advanced Design System (ADS) is depicted in Fig. 6. It can be seen that the S11 curve concentrates after adding the inductive windows and finally displays double-resonance property as shown in Fig. 7. The S-parameters of the patch antenna with transition is simulated with HFSS, de-embedded to Plane A (shown in Fig. 5(a)), and then exported to ADS. , , The parameters are as follows: , . Then by comparing the S-parameters of a single inductive window with that of its equivalent circuit on the Smith Chart, the values of the two inductors could be mapped to the real size of the inductive windows. The result in Fig. 7 shows that the

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Fig. 7. Reflection coefficients and power transmission rate matched patch antenna.

of the

Fig. 5. Patch antenna with inductive windows. (a) Full-wave model. The dimensions are denoted with the same variables as those in Fig. 1 and thus are not marked here, (b) circuit model.

Fig. 8. Geometry of the improved CP antenna: , , , , , , mm).

, , ,

, , (all in

Fig. 6. Reflect coefficients seen from Plane A to Plane E (simulated by ADS from 39 GHz to 48 GHz): seen from (a) Plane A, (b) Plane B, (c) Plane C, (d) Plane D, and (e) Plane E.

power transmission rate of the matched patch antenna is higher than 80% from 39 GHz to 48 GHz. Fig. 8 demonstrates the final geometry of the improved CP antenna. Fig. 9 shows the T/R efficiency of the improved CP antenna element. Compared to Fig. 2, the T/R efficiency shows a much flatter frequency characteristic. As frequency increases, the T/R efficiency becomes slightly higher, which compensates the directivity drop shown in the same figure. The directivity drop may be caused by substrate modes such as surface waves or leaky waves at higher frequency bands. Over the whole operating band, the measured T/R efficiency of the antenna is larger than 80%, higher than [14]. As demonstrated in Fig. 10, the measured 1 dB boresight gain bandwidth is expanded from 3 GHz (41.5 GHz to 44.5 GHz) to 9 GHz (39 GHz to 48 GHz). The measured result is generally lower than simulations, which may be caused by the inaccurate loss tangent at Q bands. Also shown in Fig. 10, from 38.5

Fig. 9. T/R efficiency, directivity and AR of the improved CP antenna element.

GHz to 48 GHz, the measured reflect coefficient is below 10 dB. As displayed in Fig. 9, the measured 3 dB AR bandwidth is 19.88% (38.5 GHz–47 GHz), similar to the original element. In Fig. 11, the radiation patterns are displayed. Over the whole operating band, the antenna displays stable radiations in the normal direction.

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 1, JANUARY 2015

Fig. 10. Reflect coefficient and boresight gain of improved CP element.

Fig. 12. Geometry of the 2 2 CP antenna array. (a) Overall structure. (b) Detailed view of the dual-polarized array. The ideal excitation is also denoted: , , , , , , , , , , , , , . , , (all in mm). The dimensions of the coupler is denoted , , . the same with Fig. 1(b) (in mm):

Fig. 13. Photographs of the proposed antennas.

errors of the two orthogonal radiation fields, which will be analyzed in the following parts. The basic operating mechanism of the dual-polarized array used as a dual-fed CP array is similar to [11], and this communication focuses primarily on its bandwidth properties. Two power dividers are adopted to divide the two power flows into four. The dual-polarized array is fed with proper excitations indicated in Fig. 12(b). SIW to CPW transitions [13] are used to connect the array and the power dividers. Photographs of the proposed antennas are shown in Fig. 13. A. Analysis on the Two-Element Series-Fed Array and Gain Bandwidth Consideration

Fig. 11. Radiation patterns of the improved CP antenna element. (a) In xoz plane, at 39 GHz. (b) In yoz plane, at 39 GHz. (c) In xoz plane, at 43 GHz. (d) In yoz plane, at 43 GHz. (e) In xoz plane, at 48 GHz. (f) In yoz plane, at 48 GHz.

III. 2

2 ANTENNA ARRAY

Fig. 12 shows the geometry of the 2 2 CP array, which consists of a dual-polarized array fed by a compensating coupler. The coupler is used to both generate orthogonal signals and compensate the amplitude

The essential part of the array configuration shown in Fig. 12(b) is actually a two-element series-fed array [11]. The reflect coefficient in Fig. 14(a) illustrates that the array has two resonances and Fig. 14(b), (c) reveal that at the first resonance, fields on the two patches are out of phase and thus cancels boresight radiation. While at the second resonance, they are in phase, which enhances boresight radiation. As analyzed above, the first mode is actually a non-boresight radiating one and thus our antenna design should utilize the second mode. However, it is not suitable to place the second resonance at the center of our operating band as is done in usual cases. Due to the influence of the non-boresight radiating mode, directivity of the series-fed array suffers from a severe drop at lower bands also shown in Fig. 14(a). In this case, to diminish such effect, the resonant frequency of the second mode should be selected lower than center frequency. However, offsetting the resonant frequency too low will lead to higher reflect coefficient at the higher frequencies of the operating band. Thus, a proper

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Fig. 16. Simulated amplitude ratios of the compensating coupler. The electrical field at the peak frequency is shown in the inset.

Fig. 14. Two modes of the two-element series-fed array. (a) Reflect coefficient. (c) Field distribution at . (b) Field distribution at

Fig. 17. Reflect coefficient and boresight gain of the CP array.

Fig. 16, the frequency-dependency of the coupler could be tuned to agree with the amplitude error and will compensate it. The final performance of the coupler is shown in Fig. 15. Fig. 15. Amplitude and phase performance of the ideally excited 2 2 CP array and the compensating coupler. The phase difference of the two outputs of the coupler has subtracted 90 .

resonant frequency of the radiating mode should be carefully selected. In this design, after optimizing the whole antenna, it is selected at 41.6 GHz, 1.4 GHz lower than the center frequency. B. Analysis on the Axial Ratio Bandwidth Property of the 2

2 Array

As shown in Fig. 12(b), the array is first excited with ideal orthogonal signals. Then the amplitude error and phase error of the two orthogonal radiating fields are shown in Fig. 15. From 36 GHz to 47 GHz, the . However, the amplitude error is highly phase error is within frequency-dependent and the absolute value is large. C. Design of the Compensating Coupler In this part, the SIW coupler is specially designed to compensate the amplitude error. As shown in Fig. 16, the amplitude ratio of the coupler reaches a peak at a certain frequency (denoted as peak frequency). Based on the electrical field distribution at the peak frequency shown in mode the figure inset, this could be ascribed to the emergency of [15]. By controlling the width of the multi-mode region as shown in

D. Performance of the 2

2 Array

In Fig. 17, the measured and simulated boresight gains are presented. It is shown that by properly selecting the resonant frequency of the boresight radiating mode, the simulated center frequency of the achieved gain bandwidth could be optimized to coincide with the desired one. As a result, a simulated 3 dB gain bandwidth of 19.76% is achieved. The measured 3 dB gain bandwidth is 18.2% (40 GHz–48 GHz) with a peak value of 12.35 dBi. The discrepancy between the measurement and simulation may be caused by permittivity variance and fabrication errors. In Fig. 18, the T/R efficiency is displayed. The measured results reveal that the T/R efficiency is higher than 78% over the whole operating band. Also shown in Fig. 18, the measured 3 dB AR bandwidth reaches 22.8% (37 GHz to 46.5 GHz). In Fig. 19, the radiation patterns are demonstrated. The maximum radiation directions in both planes are found to be slightly frequency dependent. This is because the fundamental part of the array is actually a series-fed array. IV. CONCLUSION This communication proposes an original CP antenna element, an improved element and a 2 2 array for Q-band applications on a single

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the 1 dB gain bandwidth is expanded from 7% to 21%. For the array design, the gain-bandwidth and AR-bandwidth properties of the dual-polarized patch array [11] are revealed. The measured 3 dB gain and AR bandwidth are respectively 18.2% and 22.8%. These antennas possess the advantages of wide bandwidth, low cost, and easy integration with other planar circuits, while one drawback is the relatively large size occupation. ACKNOWLEDGMENT The authors would like to thank Dr. P. Chen for his helpful suggestions on the measurement of the antennas.

REFERENCES Fig. 18. T/R efficiency of the CP array.

Fig. 19. Radiation patterns of the CP array. (a) in xoz plane, at 40 GHz, (b) in yoz plane, at 40 GHz, (c) in xoz plane, at 42 GHz, (d) in yoz plane, at 42 GHz, (e) in xoz plane, at 47 GHz, (f) in yoz plane, at 47 GHz.

layer PCB. For the original CP element, by feeding a patch antenna with an SIW coupler, the overlapped bandwidth of impedance and axial ratio reaches more than 20%, while the gain bandwidth is narrower. By introducing second-order inductive windows to the improved element,

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