Temperature Issues in Saliency-Tracking-Based ... - IEEE Xplore

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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 47, NO. 3, MAY/JUNE 2011

Temperature Issues in Saliency-Tracking-Based Sensorless Methods for PM Synchronous Machines David Díaz Reigosa, Member, IEEE, Fernando Briz, Senior Member, IEEE, Michael W. Degner, Senior Member, IEEE, Pablo García, and Juan Manuel Guerrero, Member, IEEE

Abstract—High-frequency carrier signal excitation has been widely investigated for the position sensorless control of permanent-magnet synchronous machines (PMSMs) near and at zero speed. Selection of the injected signal characteristics (shape, frequency, magnitude, etc.) is a tradeoff between performance criteria of the sensorless control (stability, accuracy, robustness, bandwidth, etc.) and minimization of its adverse effects (additional losses, noise, vibration, etc.). The increased losses due to the injected signal can be of importance in PMSMs since they can produce a significant increase of the temperature in the rotor magnets. This can adversely impact the normal operation of the machine and can eventually result in the irreversible demagnetization of the magnets. This paper analyzes the impact that excitation using high-frequency signals for sensorless control of PMSMs has on the machine’s temperature. The machine design, as well as the type of injected high-frequency signal, will be shown to strongly influence the machine’s thermal behavior. Analytical models will be developed to explain this behavior, with experimental results being used to verify the analysis. Index Terms—Carrier signal excitation, permanent-magnet (PM) synchronous machines (PMSMs), temperature.

I. I NTRODUCTION

D

ESIGN and control of permanent-magnet (PM) synchronous machines (PMSMs) have been the focus of significant research effort for more than 30 years [1]–[3], [15]–[21], [23]–[25], [27] due to their dynamic performance, high power density, and high efficiency. Torque and/or position control of PMSMs requires measurement or estimation of both the magnet polarity and the rotor position. Position sensors are normally used for position feedback. The use of position sensors, though, requires space and cabling and adds a nonnegligible cost. To overcome the use of position Manuscript received September 24, 2010; revised November 23, 2010; accepted December 26, 2010. Date of publication March 10, 2011; date of current version May 18, 2011. Paper 2010-IDC-364.R1, presented at the 2010 IEEE Energy Conversion Congress and Exposition, Atlanta, GA, September 12–16, and approved for publication in the IEEE T RANSACTIONS ON I NDUSTRY A PPLICATIONS by the Industrial Drives Committee of the IEEE Industry Applications Society. This work was supported in part by the Research, Technological Development and Innovation Programs of the Spanish Ministry of Education–ERDF under Grant MEC-ENE2007-67842-C03-01 and the Ministry of Science and Innovation under Grant MICINN-10-CSD200900046. D. D. Reigosa, F. Briz, P. García, and J. M. Guerrero are with the Department of Electrical, Computer, and Systems Engineering, University of Oviedo, 33204 Gijón, Spain (e-mail: [email protected]; [email protected]; [email protected]; [email protected]). M. W. Degner is with Ford Motor Company, Dearborn, MI 48121-2053 USA (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIA.2011.2126033

sensors, sensorless control techniques have been proposed and studied during the past two decades [1]–[12], [15], [22], [24], [26], [27]. Sensorless control methods can be divided into back-EMF [22] and saliency-tracking-based techniques [1]–[12]. Back-EMF methods have been demonstrated to be fully capable in the medium- to high-speed range. However, their performance (accuracy, robustness, bandwidth, etc.) deteriorates as the speed decreases due to the diminishing magnitude of the back-EMF signal and eventually fails in the low-speed range. In addition, for the particular case of PMSMs, back-EMF methods do not provide magnet polarity information at standstill, which makes start-up problematic [2]. To overcome the limitations of the back-EMF methods, saliencytracking-based sensorless methods have been proposed [1]–[6], [10]–[12]. These methods inject a high-frequency signal (normally a voltage) that interacts with rotor-position-dependent saliencies (asymmetries), modulating the resulting highfrequency stator currents, from which the rotor position can be estimated. The saliency-tracking-based methods provide a viable option for sensorless position/speed control. These signal excitation techniques have been applied to PMSMs for both initial rotor position estimation (including magnet polarity detection) [5]–[7] and very low speed control [1]–[6], [10]–[12]. There are, however, several sources of error that can limit the accuracy of these techniques. Secondary saliencies caused by saturation, nonideal behavior of the inverter (distortion of the injected carrier signal), etc., have been reported as the most important sources of error [10], [11], [24], [26], [27]. Other sources of error include the equivalent high-frequency resistance [15], variations in the equivalent high-frequency inductance due to saturation [11], measurement errors [10], etc. The carrier signal excitation has been created by either modification of the pulsewidth modulation pattern [7]–[9] or the superposition of a high-frequency signal on the fundamental excitation [1]–[6], [10]–[12]. Although significant research has been performed on the selection of the carrier signal to obtain satisfactory sensorless control, little research has been done to evaluate the impact that each solution (type and parameters of the carrier signal) has on the normal operation of the machine. The additional losses created by the carrier signal can have a significant impact on the rotor temperature, which can be of tremendous importance in PMSMs. The stator and rotor losses in PMSMs depend on the type of carrier signal, including its magnitude and frequency, as well as on the machine design. The magnet losses are particularly important in surface PMSMs (SPMSMs) where the

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REIGOSA et al.: TEMPERATURE ISSUES IN SALIENCY-TRACKING-BASED SENSORLESS METHODS FOR PMSMs

magnets are subjected to higher flux harmonics than interior PMSMs (IPMSMs) [19]–[21]. The additional losses caused by the excitation of a highfrequency signal result in an increased temperature, which can have several adverse effects on the normal operation of the machine. Elevated magnet temperatures decrease the torque capability, which may or may not be acceptable depending on the particular application. It has also been shown [15] that increased magnet temperature can produce significant errors in the estimated position. Finally and worst of all, excessive heating of the magnets can result in irreversible demagnetization of the rotor magnets. This paper analyzes the effects that excitation using a high-frequency signal for position estimation purposes has on the magnet temperature in PMSMs. Both IPMSMs and SPMSMs will be included in the analysis. The physical principles and formulation are presented first. Experimental results to confirm the theoretical predictions are then provided. II. C ARRIER S IGNAL E XCITATION M ETHODS AND T HEIR I MPACT ON THE T EMPERATURE IN PMSMs Many types of high-frequency periodic signals have been proposed for use in sensorless control. This paper will focus on three of these: rotating voltage vector [5], [11], [15], pulsating voltage [5], [11], [15], and square-wave carrier voltage [13], [14]. Selection of the high-frequency signal (type, magnitude, and frequency) is made based on performance criteria of the sensorless control. This section reviews the principles of these methods and presents preliminary results showing the impact that the various carrier signals have on the motor temperature. Section IV will more thoroughly study the motor temperature effect. A. Review of Carrier-Signal-Excitation-Based Methods The high-frequency three-phase ac machine models normally used for the study of saliency-based sensorless methods assume a purely inductive behavior for the machine (1) [1]–[6], [10]–[12]. One limitation of this approach is that it does not include the losses due to the use of a carrier signal excitation r Vdqsc

=

r Lrdqs pIdqsc .

r Rdqr

Fig. 1. Block diagram showing the implementation of carrier-signalexcitation-based sensorless control.

shown in the block diagram of the implementation in Fig. 1 [5], [11], [13]–[15]. The most common types of carrier signal are the rotating (3) [5], [11], [15] and pulsating (4) [5], [11], [15]. Recently, square-wave excitation has also been proposed [13], [14], and the square-wave excitation can be expressed as a Fourier series (5) s = Vc ejωc t Vdqsc r Vdqsc

r Vdqsc

= Vc cos(ωc t) ∞ 4Vc = π

h=1,3,5,...

(3) (4) 1 sin [(2h − 1)ωc t + π] . (5) 2h − 1

Asymmetries intrinsic to the rotor design result in differences between the high-frequency electric parameters in the d- and q-axes, particularly in the inductive terms in (2). When a carrier signal voltage excitation is injected, the resulting carrier current is modulated by the machine’s saliencies. The carrier current can be shown to be (6) (shown in a stationary reference frame) when a rotating carrier signal voltage (3) is injected [5], [11], [15] and (7) (shown in an estimated rotor position synchronous reference frame) when a pulsating carrier signal voltage (4) is injected [5], [11], [15]. For the case of square voltage excitation (5) [13], [14], the resulting current consists of a series of harmonic components of the form shown by (7) for an individual harmonic, which are the result of the harmonic series shown in (5) isdqsc = Ipc ej(ωc t+ϕpc −π/2) + Inc ej(−ωc t+2θr +ϕnc +π/2) irdqsc = 2 IΣZ ej(ϕΣZ ) + IΔZ ej(2Δθ+ϕΔZ ) sin(ωc t)

(6) (7)

(1)

In [15], a more detailed high-frequency model was introduced that accounts for these losses through the inclusion of stator and rotor high-frequency resistances (2). These resistances attempt to account for the copper, eddy-current, and hysteresis losses in both the stator and rotor. It was also shown in [15] that the high-frequency resistance and inductance of the machine depend on the carrier signal frequency r r r r r r = Rdqr Idqsc + Rdqs Idqsc + Lrdqs pIdqsc Vdqsc

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(2) r Rdqs

where is the rotor high-frequency resistance matrix, is the stator high-frequency resistance matrix, and Lrdqs is the high-frequency inductance matrix. Superindex “r” denotes a rotor synchronous reference frame. The high-frequency carrier signal is superimposed on the fundamental excitation voltage used for torque production, as

where ωc is the carrier frequency, ϕpc and ϕnc are the positive and negative carrier current component offsets, ϕΣZ and ϕΔZ are the mean and differential impedance component offsets, θr is the rotor position, Δθ is the phase error between the estimated and the injection axes, and Ipc and Inc are the magnitudes of the positive and negative carrier currents, which, neglecting resistive terms, can be approximated as [11], [12], [15] Ipc ≈

ΣL Vc ωc ΣL2 − ΔL2

Inc ≈

ΔL Vc . ωc ΣL2 − ΔL2

(8)

Similarly, IΣZ and IΔZ are the magnitudes of the mean and differential impedance carrier currents, which, neglecting resistive terms, can be approximated as [11], [12], [15] IΣZ ≈

ΣL Vc 2 2ωc ΣL − ΔL2

IΔZ ≈

ΔL Vc . (9) 2 2ωc ΣL − ΔL2

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TABLE I R EVIEW OF P REVIOUSLY P UBLISHED C ARRIER S IGNAL PARAMETERS

For the case of squarewave voltage excitation (5) [13], [14], the resulting current consists of a series of harmonic components of the form shown by (7) for an individual harmonic, which are the result of the harmonic series shown in (5). It is interesting to point out that, while the rotor position information is contained in the components Inc and IΔZ , respectively, these components normally account for a minor portion (typically a few percent) of the overall carrier signal current. On the other hand, the components Ipc and IΣZ , which do not contain rotor position information, are responsible for almost all of the adverse effects due to the carrier signal excitation. Generally speaking, the carrier signal voltage Vc is preferred to have a magnitude as small as possible in order to minimize the resulting current and, consequently, its adverse effects. The minimum magnitude will be primarily determined by accuracy issues (the signal-to-noise ratio) [12]. For the selection of the injected signal frequency ωc , the maximum frequency corresponds to half of the switching frequency (Nyquist frequency) [34]. Below this limit, higher frequencies are normally preferred since they provide larger spectral separation with the fundamental excitation, which makes filtering easier [15]. Reviewing the literature on the subject, a large variety of carrier signal frequencies and magnitudes has been used, with Table I showing some examples. Unfortunately, not all of the reviewed references provided complete information on the carrier signal used [2], [3], [28]–[32]. In all cases, the criteria for selection of the carrier signal were based primarily on sensorless control performance, with limited published results analyzing and explaining the tradeoffs between the adverse effects of carrier signal excitation and the sensorless control performance. B. Impact of the Carrier Signal on the Machine’s Temperature The temperature increase caused by the various carrier signal excitations proposed in the literature was experimentally measured using several different PMSMs. Table II summarizes the results (details on the machine designs and signal injected are given in Section IV). The temperature increases shown in Table II were measured on the stator frame after having the machine operate in steady state long enough to reach a stable thermal condition, injecting the carrier signals without fundamental excitation. These large temperature increases in

TABLE II I NCREASE OF T EMPERATURE D UE TO THE C ARRIER S IGNAL VOLTAGE E XCITATION

the stator frame indicate that there are parts within the machine experiencing similar or larger temperature increases. It can be observed from Table II that, depending on the machine design and the type/parameters of carrier signal, temperature increases range from 3 ◦ C, which would have minimal interference with the regular operation of the machine, to 65 ◦ C, which would significantly interfere with the operation of the machine. III. L OSSES IN PMSMs Normal operation of PMSMs produces losses in both the stator and rotor, which can produce a significant increase in the machine’s temperature. Adequate thermal design is therefore of tremendous importance in guaranteeing that the machine will not surpass safe temperature limits during normal operation. This is particularly important in PMSMs, compared to other types of ac machines like induction and reluctance machines, since excessive temperatures can significantly reduce the magnet strength and even cause permanent demagnetization in extreme circumstances. Losses are primarily produced by the fundamental excitation and the switching harmonics of the inverter [17]–[21]. The carrier signal excitation adds an additional loss mechanism that must also be considered. The stator copper losses produced by the carrier signal can be expressed as the sum of the losses produced by all the current harmonics created by the carrier signal PCu = 3/2

∞

2 Rh(T ) Idqh

(10)

h=1

where Rh(T ) is the high-frequency resistance of the stator phase [15], [16] and Idqh is the magnitude of the induced carrier current harmonic. The stator iron, hysteresis, and eddy-current losses can be approximated using [17] Piron = Peddy + Physt ∞ |hBh |2 Peddy = Ke f 2 h=1

Physt = Kh f B

∞ k 1+ ΔBh B

(11)

(12)

h=1

where f is the frequency, B is the flux density peak value, Bh is the flux density peak value of the hth harmonic, ΔBh is the


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TABLE III S TATOR W INDING L OSSES

TABLE IV M OTOR PARAMETERS

Fig. 2. Variation of eddy-current loss with rotor speed at rated current without carrier signal injection in an SPMSM (from [21]).

flux reversal associated with a minor loop, and k, ke , and kh are constants. The losses in the rotor lamination can be expressed similarly to the stator iron losses (11) and (12), while the magnet eddycurrent losses can be expressed as the sum of the eddy-current losses induced by the carrier signal and/or other harmonics induced by the fundamental excitation (13) [20], [21]. The variation of Peddy_mag for a three-phase SPMSM was reported in [21], with the result being shown in Fig. 2. The parameters of the SPMSM are as follows: a magnet pole arc of 131.4 electrical degrees, 36 stator slots, and 42 poles. It is noted that the eddycurrent losses shown in Fig. 2 are only created by fundamental dependence harmonics since the carrier signal injection was not considered in [21] Kh Jh2 f 2 (13) Peddy_mag = where Kh is a constant and Jh is the current sheet induced in the PMs by the carrier signal or other harmonics. It can be concluded that the stator losses due to the carrier signal excitation increase with its magnitude (11)–(13) and also with its frequency (12) and (13). For the rotor side, it can also be concluded from (12)–(14) that magnet and core losses due to the carrier signal excitation increase with its magnitude and frequency. For the purposes of this paper, which focuses on the low speed operation of the machine, where carrier signal based sensorless control is applicable, the losses in the rotor due to harmonics created by the fundamental excitation that are asynchronous with the rotor are considered negligible. At higher speeds these losses can be significant, depending on the machine design, but at low rotor speeds they do not contribute significantly to the temperature of the rotor. (See Fig. 2 and [21]). It was already shown in Section II (see Table II) that the magnitude of the losses due to the carrier signal excitation depends strongly on the type of signal being injected. Table III summarizes the analytically derived expressions for calculating stator winding losses for each form of carrier signal excitation. It can be observed that the losses for the case of rotating carrier signal excitation are twice those for the case of sinusoidal pulsating signal excitation for the same amplitude of carrier current excitation. This can be explained by recognizing that a pulsating carrier voltage can be expressed as two oppositely rotating carrier signal voltages, each with half the magnitude of

the overall pulsating signal (14). Each of these rotating carrier signal voltages has 1/4 the loss of the single rotating carrier signal excitation, which results in 1/2 the loss in total

r = Vc cos(ωc t) = Vdqsc

Vc jωc t Vc −jωc t e + e . 2 2

(14)

The winding losses using squarewave carrier signal excitation are the same as for sinusoidal pulsating carrier signal excitation when both have a frequency equal to half of the switching frequency. However, sensorless control based on a sinusoidal pulsating carrier signal excitation [5], [11], [15] cannot be performed at this frequency. IV. E XPERIMENTAL R ESULTS In this section, experimental results for different forms of carrier signal excitation, both without and with fundamental current in the machine, are presented. Experimental results using the machines shown in Table IV are provided to illustrate and support the discussion. The stator frame temperature was measured in all the cases since it was easily accessible. It is noted that the cooling system of the test machines (see Table IV) is natural convection, and the stator windings are distributed. In addition, the magnet temperature was also measured for motor #2, and details on this are provided later in this section. Although the magnet temperature is not necessarily equal to the stator frame temperature, the stator temperature was considered an acceptable indicator of the induced losses in the machine [16]. PMSMs can be classified into two major groups: SPMSMs and IPMSMs. For the type of analysis presented in this paper, it is convenient to further subdivide SPMSMs into two subcategories: machines where the magnets cover the total surface of the rotor (magnet pole arc = 180 electrical degrees) and machines where the magnets do not cover the total rotor surface (magnet pole arc < 180 electrical degrees). A. Carrier Signal Excitation Without Fundamental Excitation This section evaluates the effect of the carrier signal on the machine temperature when no fundamental excitation is

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TABLE V C ARRIER S IGNAL PARAMETERS

Fig. 4. Measured stator temperatures, motor #2. Rotating carrier signal excitation: (o) 2500 Hz and (∗ ) 500 Hz. Pulsating carrier signal excitation: (♦) 2500 Hz and () 500 Hz. Square-wave carrier signal excitation: (x) 5000 Hz.


present. The types of carrier signal excitation analyzed were the rotating (3), pulsating (4), and square wave (5). The magnitudes and frequencies used for the carrier signal excitation are listed in Table V, where Vc and Ipc , IΣZ are the magnitude (peak value) of the carrier signal voltage and the induced carrier signal current and fc is the carrier frequency. The carrier signal current magnitude was set to the same value in all cases to make comparison easier. It was selected based on sensorless control performance criteria, i.e., to guarantee stable sensorless control with an accuracy in the estimated position in the range of one to two mechanical degrees. It has been previously demonstrated that, for the same magnitude of excitation, rotating- and pulsating-carrier-signal-based sensorless controls have similar performance [12]. It should be noted that different frequencies were used for the case of square-wave excitation and for the case of rotating and pulsating excitation, to be consistent with previous published works (see Table I). As was stated in Section III, for the same carrier signal current magnitude, a lower carrier signal frequency is preferred from a loss perspective. This is confirmed by the results in Figs. 3–5 and Table I. It can be observed that higher carrier signal frequencies produce larger temperature increases. It can also be observed that the induced losses using a square-wave carrier signal excitation of 5 kHz are higher than the induced losses by the pulsating excitation using 2.5 kHz (the highest frequency that has been proposed in the literature), confirming the conclusions in Section III. SM P SM arc = 180: The temperature rise for this case due to a pulsating carrier signal excitation is expected to be approximately one-half that produced by a rotating carrier


signal (3), (4), (10)–(12). This analysis is confirmed with the experimental results obtained for motor #1. As can be observed from Fig. 3 and Table II, the increase of temperature for a carrier signal frequency of 2500 Hz is 65 ◦ C with a rotating carrier signal and 32 ◦ C with a pulsating carrier signal. Similar conclusions can be reached for a 500-Hz carrier signal frequency (see Table II). SM P SM arc < 180: For machines with a magnet pole arc of less than 180 electrical degrees, the rotor losses will depend on the excitation axis for the pulsating carrier signal. When the axis is aligned with the magnet pole pieces, the losses are expected to be higher in the magnets. When the excitation axis is in the interpolar space, the losses would be expected to be lower in the magnets. In Fig. 4 and Table II, the experimental results obtained for motor #2 are shown with the pulsating carrier signal injected in the d-axis. As can be observed, the rotating carrier signal produces higher temperature than the pulsating carrier signal. Figs. 6 and 7 show the experimental results obtained for motors #1 and #2, respectively, when a pulsating carrier signal is injected in the d-axis, q-axis, and an axis that is 45 electrical


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Fig. 8. Simplified thermal model of the machine.

A simple explanation of this behavior can be derived using a simplified thermal model of the machine (see Fig. 8) Tm = Tr + Rθ Plosses Fig. 6. Measured stator temperature, motor #1. Pulsating carrier signal excitation: ωc = 2500 Hz and IC = 8%IRATED . (x) q-axis excitation, (o) d-axis excitation, and excitation axis at 45 electrical degrees from d/q-axis.

Fig. 7. Measured stator temperature, motor #2. Pulsating carrier signal excitation: ωc = 2500 Hz and IC = 8%IRATED . (x) q-axis excitation, (o) d-axis excitation, and excitation axis at 45 electrical degrees from d/q-axis.

degrees phase shifted from the d- and q-axes. The experimental results show that, for motor #1 (magnet pole arc = 180 electrical degrees), the increase in temperature is approximately the same, independent of the axis of excitation. This can be explained by the fact that the magnets cover the total rotor surface and the losses in the magnet are independent of the excitation axis. From Fig. 7, it can be observed that the increase of temperature for motor #2 (magnet pole arc = 150) depends on the excitation axis, with the increase of temperature when the carrier signal is injected in the d-axis (aligned with the magnet pole pieces) being higher than that when it is injected in the q-axis (aligned with the laminated steel between magnets). If the excitation axis is between the d- and q-axes [1], the increase of temperature is between the increases of temperature when the carrier signal is injected in the d- or q-axis but is closer to the temperature for d-axis excitation. Based on this observation, pulsating q-axis excitation would be preferred in terms of machine losses when the magnet pole arc is less than 180 electrical degrees. However, it should be noted that the q-axis injection produce higher acoustic noise and higher torque ripple than the d-axis injection.

(15)

where Plosses represents the power losses caused by the carrier signal excitation, Tr is the room temperature, Tm is the surface temperature of the machine, and Rθ is the thermal resistance of the machine. The thermal resistance of motor #2 was experimentally measured to be ≈ 0.81 ◦ C/W. The difference in the effective highfrequency resistance of the rotor for motor #2 from the d-axis to the q-axis was measured to be ≈ 1.1 Ω, i.e., high-frequency resistance of the magnets, aligned with the d-axis, relative to the laminated steel, aligned with the q-axis. Assuming that the difference in losses between injecting the pulsating signal in the d-axis and injecting the pulsating signal in the q-axis can be modeled using an equivalent circuit like that developed in (2), the temperature difference obtained using the simplified thermal model is ≈ 7.1 ◦ C. This compares favorably to the measured temperature difference of ≈ 6.5 ◦ C. To confirm the experimental results shown in Fig. 7 and to verify the assumption that the rotor and magnet losses are different depending on the axis of excitation when the magnet pole arc is less than 180 electrical degrees, the magnet temperature of motor #2 was measured with rotating carrier signal and pulsating carrier signal (d- and q-axis) excitations. The magnet temperature was measured with a camera that provides an accurate measurement of the surface temperature (see Fig. 9), and details of the measurement process can be found in [16]. Since the window diameter is not large enough to obtain a complete thermal image of the magnet surface, several thermal images, taken with the machine rotating, were composed to obtain the complete thermal image of a piece of PM [see Fig. 9(b)]. The results presented in Fig. 9(c)–(e) show the temperature distribution during the heating process. It can be observed from Fig. 9(c) that rotating carrier signal excitation produces a relatively uniform magnet temperature increase. Contrary to this, a temperature gradient, i.e., a non-uniform temperature distribution, can be observed for the case of pulsating carrier signal excitation [Fig. 9(d) and (e)]. It should be noted that the magnet temperature for the rotating carrier signal injection [Fig. 9(c)] is higher than for the pulsating carrier signal injection [Fig. 9(d) and (e)] and that while injection in the d-axis results in rather uniform temperature increase of all the magnet surface [Fig. 9(d)], for the case of q-axis injection

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TABLE VI S TEADY-S TATE T EMPERATURES (ROOM T EMPERATURE ≈ 25 ◦ C), ωR = 4 Hz, IC = 8%IRATED , I = 100%IRATED

Fig. 10. Increase of temperature. Rotating carrier signal excitation: ωc = 2500 Hz, IC = 8%IRATED , ωR = 4 Hz, and I = 100%IRATED .

Fig. 9. (a) Window in motor #2. (b) Scheme of the thermal image composition. (c)–(e) Thermal image composition of rotating carrier signal excitation, d-axis pulsating carrier signal excitation, and q-axis pulsating carrier signal excitation, respectively. ωc = 2500 Hz, and IC = 8%IRATED . The thermal images were obtained after 15 min of continued carrier signal excitation.

the temperature increase is not uniform, being slightly higher in the left border of the magnet in Fig. 9(e). IMPSM: Since the magnets in this type of machines are covered by the laminated steel and are not directly adjacent to the air gap, the eddy currents induced in the PMs by the carrier signal excitation will be lower than that in SPMSMs. From Fig. 5, it can be observed for this particular IPMSM that all of the saliency-tracking-based techniques tested result in a lower temperature when compared to the SPMSMs tested. Even so, the temperature increases seen in the IPMSM may still not be acceptable depending on its particular application. B. Fundamental Excitation and Carrier Signal Excitation When the machine has fundamental excitation, the combined effects of the carrier signal excitation and the fundamental excitation on temperature need to be considered. It should be noted, however, that superposition of these two excitations does not apply since the temperature increase

when carrier signal excitation and fundamental excitation are present simultaneously is different from the temperature increase obtained when the effects of carrier signal excitation and fundamental excitation acting alone are added together. When the fundamental excitation is present, the increase of temperature due to the carrier signal excitation is smaller than that observed when the carrier signal excitation alone was present. Table VI summarizes the steady-state temperatures reached in the three test machines for just fundamental excitation. For these experiments, the machines were operating in the steady state at 4 Hz and 100% rated load. Once their temperatures reached steady state, the carrier signal was added to the fundamental excitation, with the increase in temperature as a function of time for a rotating carrier signal being shown in Fig. 10. Table VI shows the measured steady-state increase in temperature for each of the studied carrier signal excitations. As can be observed, the increase in temperature when the carrier signal is present, in addition to the fundamental excitation, is lower than that when only the carrier signal excitation is present. Comparing the experimental results from Table II and Table VI, it can be observed that the increase in temperature induced by the carrier signal excitation is lower when it is superimposed on the fundamental excitation than that when it is applied without fundamental excitation. This behavior can be explained by the increase of the thermal conductance of the frame with temperature [35].


V. C ONCLUSION Carrier signal excitation for saliency-tracking-based sensorless control of PMSMs can have a significant impact on the machine’s temperature, including the rotor magnets. Analysis of this impact has been presented in this paper. The analysis covered different forms of carrier signal excitation and different machine designs. It was concluded that, for the same magnitude of carrier signal current, pulsating carrier signal excitation has lower losses than either rotating carrier signal or squarewave carrier signal excitation. Higher frequencies also result in larger overall losses and, consequently, larger temperature increases. This conclusion is of great importance since the use of higher frequencies is, in general, beneficial in terms of sensorless control performance. It has also been shown that the IPMSMs are more attractive in terms of machine losses than the SPMSMs for their use with saliency-tracking-based sensorless methods. Experimental verification has been used to confirm the results.

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David Díaz Reigosa (S’03–M’07) was born in Spain in 1979. He received the M.E. and Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 2003 and 2007, respectively. In 2007, he was a Visiting Scholar with the Wisconsin Electric Machines and Power Electronics Consortium, University of Wisconsin, Madison. He is currently an Associate Professor in the Department of Electrical, Computer, and Systems Engineering, University of Oviedo. His research interests include sensorless control of induction motors, permanentmagnet synchronous motors, and digital signal processing. Dr. Reigosa was the recipient of the IEEE Industry Applications Society Conference Prize Award in 2007. From 2004 to 2008, he was a recipient of a fellowship from the Personnel Research Training Program funded by the Regional Ministry of Education and Science of the Principality of Asturias.

Fernando Briz (A’96–M’99–SM’06) received the M.S. and Ph.D. degrees from the University of Oviedo, Gijón, Spain, in 1990 and 1996, respectively. From June 1996 to March 1997, he was a Visiting Researcher at the University of Wisconsin, Madison. He is currently an Associate Professor in the Department of Electrical, Computer, and Systems Engineering, University of Oviedo. His research interests include control systems, power converters and ac drives, sensorless control of ac drives, magnetic levitation, diagnostics, and digital signal processing. Dr. Briz was a recipient of the 2005 IEEE T RANSACTIONS ON I NDUSTRY A PPLICATIONS Third Place Prize Paper Award and three IEEE Industry Applications Society Conference Prize Paper Awards in 1997, 2003, and 2007.

Michael W. Degner (S’95–A’98–M’99–SM’05) received the B.S., M.S., and Ph.D. degrees in mechanical engineering, with a focus on electric machines, power electronics, and control systems, from the University of Wisconsin, Madison, in 1991, 1993, and 1998, respectively. His Ph.D. dissertation was on the estimation of rotor position and flux angle in electric machine drives. In 1998, he joined the Ford Research Laboratory, Dearborn, MI, where he worked on the application of electric machines and power electronics in the automotive industry. He is currently the Senior Technical Leader of the Electric Machine Drive Systems Department, Hybrid Electric Vehicle and Fuel Cell Vehicle Laboratory, Ford Research and Advanced Engineering, Ford Motor Company, where he is responsible for the development of electric machines, power electronics, and their control systems for hybrid and fuel cell vehicle applications. His research interests include control systems, machine drives, electric machines, power electronics, and mechatronics. Dr. Degner was a recipient of the 2005 IEEE T RANSACTIONS ON I NDUS TRY A PPLICATIONS Third Place Prize Paper Award and has been the recipient of several IEEE Industry Applications Society Conference Paper Awards.

Pablo García received the M.S. and Ph.D. degrees in electrical engineering and control from the University of Oviedo, Gijón, Spain, in 2001 and 2006, respectively. In 2004, he was a Visiting Scholar with the Wisconsin Electric Machines and Power Electronics Consortium, University of Madison, Wisconsin. He is currently an Assistant Professor in the Department of Electrical, Computer, and Systems Engineering, University of Oviedo. His research interests include sensorless control, diagnostics, magnetic bearings, and signal processing. Dr. García was a recipient of a fellowship from the Personnel Research Training Program funded by the Spanish Ministry of Education for the period 2002–2006.

Juan Manuel Guerrero (S’00–A’01–M’04) received the M.E. degree in industrial engineering and the Ph.D. degree in electrical and electronic engineering from the University of Oviedo, Gijón, Spain, in 1998 and 2003, respectively. Since 1999, he has been with the Department of Electrical, Computer, and Systems Engineering, University of Oviedo, where he has occupied different teaching and research positions and is currently an Associate Professor. From February to October 2002, he was a Visiting Scholar at the University of Wisconsin, Madison. From June to December 2007, he was a Visiting Professor at Tennessee Technological University, Cookeville. His research interests include parallel-connected motors fed by one inverter, sensorless control of induction motors, control systems, and digital signal processing. Dr. Guerrero was a recipient of an award from the College of Industrial Engineers of Asturias and León, Spain for his M.E. thesis in 1999, the IEEE Industry Applications Society Conference Prize Paper Award in 2003, and the University of Oviedo Outstanding Ph.D. Thesis Award in 2004.

Temperature Issues in Saliency-Tracking-Based ... - IEEE Xplore

Temperature Issues in Saliency-Tracking-Based ... - IEEE Xplore

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