Using Reverse-Blocking IGBTs in Power Converters for Adjustable ...

IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 42, NO. 3, MAY/JUNE 2006

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Using Reverse-Blocking IGBTs in Power Converters for Adjustable-Speed Drives Christian Klumpner, Member, IEEE, and Frede Blaabjerg, Fellow, IEEE

Abstract—A new semiconductor power device that is urgently needed particularly in power converter topologies, the reverse blocking insulated gate bipolar transistor (RB-IGBT), has been realized by adding minor changes to the structure of a standard IGBT to make it capable of withstanding reverse voltage. However, the switching behavior of the device’s intrinsic diode during reverse recovery is not as good as a discrete IGBT and series diode implementation. This paper analyzes the use of this device in three power converter topologies that may benefit from it, namely: 1) the matrix converter, 2) the two-stage direct power converter (DPC), and 3) the three-level voltage source rectifier. A commutation method to override the poor reverse-recovery characteristic of the RB-IGBT intrinsic diode in a two-stage DPC is proposed. A loss analysis shows that by using RB-IGBTs the efficiency of the two-stage DPC becomes similar to a two-level voltage source converter. Index Terms—Bidirectional switch, power converter, reverse blocking, reverse recovery, semiconductor losses.

I. I NTRODUCTION

M

OST OF TODAY’S industrially manufactured power converters for adjustable-speed drives (ASD) are based on unidirectional switches, which consist of an insulated gate bipolar transistor (IGBT) and an antiparallel fast-recovery diode (FRD). However, some converter topologies require bidirectional alternating current (ac) switches that are able to control current (ON state) and withstand voltage (OFF state) in both directions. Such devices can be created by embedding an IGBT in a diode bridge [Fig. 1(a)] or by connecting two unidirectional switches in antiseries in a common collector (CC) [Fig. 1(b)] or a common emitter (CE) [Fig. 1(c)] configuration. Alternatively, the newly developed reverse-blocking (RB) IGBTs, which can withstand voltage in both directions and hence are an ideal switch for current source converters, offer a convenient way to implement a bidirectional switch: two RB-IGBTs in an antiparallel connection [Fig. 1(d)] or in a hybrid connection where a true RB-IGBT and a discretely implemented Paper IPCSD-05-101, presented at the 2003 Industry Applications Society Annual Meeting, Salt Lake City, UT, October 12–16, and approved for publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Industrial Power Converter Committees of the IEEE Industry Applications Society. Manuscript submitted for review July 1, 2003 and released for publication February 13, 2006. This work was supported by the Innovation Post.Doc. program of the Danish Research Council and by Danfoss Drives A/S under Contract 2013-01-0045. C. Klumpner is with the School of Electrical and Electronic Engineering, University of Nottingham, Nottingham NG7 2RD, U.K. (e-mail: [email protected]). F. Blaabjerg is with the Institute of Energy Technology, Aalborg University, DK-9220 Aalborg East, Denmark (e-mail: [email protected]). Digital Object Identifier 10.1109/TIA.2006.872956

Fig. 1. Possible configuration of bidirectional switches. (a) Diode-embedded IGBT. (b) CC-IGBTs. (c) CE-IGBTs. (d) Antiparalleled RB-IGBTs. (e) Hybrid bidirectional switch (RB-IGBT antiparalleled with an IGBT and a series diode).

RB-IGBT are antiparallel [Fig. 1(e)]. (The importance of the last arrangement will be explained later.) The embedded switch is the most convenient to control, but it has high conduction losses and in some applications [1]–[3] requires snubbers for safe commutation. The CC/CE configuration is easy to implement since it requires only a change of connections between standard silicon chips on a ceramic substrate to create a new power module [3]. Conduction losses are still high (one IGBT and one diode), but snubberless commutation techniques make it suitable to use in a wide range of applications. Configurations using the RB-IGBT device decrease conduction losses and reduce the number of power devices to only two per bidirectional switch as compared with four (CC/CE) or five per embedded switch. However, since the reverse voltage blocking capability is obtained by changing the structure of a normal IGBT, the switching performance of its intrinsic diode is poor compared with that of a discrete implementation. This paper begins by analyzing the particularities of an industrially manufactured RB-IGBT and by comparing its switching behavior against a discrete implementation (diode + IGBT). The application of the RB-IGBT to several converter topologies is then analyzed to show that in some cases the intrinsic diode disadvantage does not influence its performance, whereas in other cases careful analysis is required to determine the conditions where RB-IGBTs are more effective to use. A new commutation method for a two-stage direct power converter (DPC) using RB-IGBTs in the rectification stage is proposed and analyzed by simulation and experiments. Finally, the semiconductor losses in the investigated converter topologies are determined when using RB-IGBTs compared to discrete alternatives. The result of these analyses is that RB-IGBTs can decrease the losses and the number of power components required compared with a voltage-source converter (VSC) operating under similar conditions.

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Fig. 2. Overlap commutation of two RB-IGBTs that causes a hard turn-on situation in the incoming switch, whereas the intrinsic diode of the outgoing RB-IGBT experiences reverse recovery.

II. A NALYSIS OF THE RB-IGBT Recently, some semiconductor manufacturers have started producing RB-IGBTs by changing the structure of a normal nonpunch-through (NPT) IGBT to improve its reverse voltage blocking capability. However, the resultant intrinsic diode in the device has a poorer switching behavior than a standard FRD. This creates concern during overlap commutation when the load current is force commutated from an outgoing RB-IGBT to an incoming RB-IGBT. Under these conditions, the incoming device is turned ON while the outgoing RB-IGBT is still on. The outgoing RB-IGBT will experience reverse recovery, and the reverse recovery peak current and the turn-off time will depend on the switching behavior of the intrinsic diode of the RB-IGBT device. The amount of charge carriers in the junction and its capacitance, and the di/dt of the current through the incoming switch, influence the reverse recovery current and the switching time. Since the first two parameters cannot be changed for a given device, the only way to influence the reverse-recovery behavior is by changing its turn-on gate resistance. To assess this condition, it is necessary to analyze the forced commutation process between two RB-IGBTs. This is shown in Fig. 2, where a positive load current Iload has to be commutated from a smaller (IN1) to a higher (IN2) source potential. Overlap commutation is required to provide continuity of the inductive load current; hence, the incoming device must be turned ON before the outgoing device is turned OFF. Although the outgoing device is still on, the load current will immediately start commutating to the incoming device because the intrinsic diode of the outgoing device becomes reverse biased. The reverse recovery current IRR depends on the di/dt of the current through the incoming device, which depends on the turn-on behavior of the IGBT and on the gate resistance. This is seen as a circulating current from the voltage source side and causes turn-on losses in both the outgoing (diode) and the incoming (IGBT) devices. Fig. 3 presents a comparison of the turn-on behavior of a normal diode [Fig. 3(a)] with the intrinsic diode of an RB-IGBT [Fig. 3(b)]. As compared with a normal FRD with similar ratings, the reverse recovery current is 2.6 times higher. Note that the gate resistance has been increased compared to the data sheet to decrease the di/dt during turn-on to 200 A/µs as compared to 400 A/µs as noted in the data sheet. This decreases the reverse recovery current but increases the turn-on time, as can be seen.

Fig. 3. Hard turn-on behavior of an RB-IGBT when it switches with di/dt = 200 A/µs against (a) a normal fast-recovery (DSEI 30-12) diode: IC_pk = 21 A, IRR = 6 A; and (b) the intrinsic diode of the complementary RB-IGBT (IXRH 50N100) device: IC_pk = 31 A, IRR = 16 A. Test conditions: Rg = 100 Ω, Vg-on = 15 V, Vg-off = −15 V, VCE = 250 V, Iload = 15 A. Scale: VCE = 50 V/division, IC = 5 A/division, Iload = 10 A/division, Time = 200 ns/division.

Switching against a diode with higher reverse-recovery current has the following disadvantages. • The first is increased current stress in the incoming IGBT. Usually, the maximum admissible current is about twice the rated current of the device, whereas the reverserecovery current may reach perhaps 20%–50% of the rated current. • Increased turn-on losses of the incoming device, since they are influenced by the reverse recovery current. • Increased turn-off losses of the outgoing device, which is the intrinsic diode of the RB-IGBT. One method to reduce the reverse recovery current peak is to decrease the di/dt of the IGBT during turn-on by increasing its gate resistance or by using an active gate driver. However, this also increases the turn-on time. Unfortunately, reducing the reverse recovery peak current and increasing the turn-on time have opposing effects on the incoming IGBT turn-on losses. Hence, since the collector current through the incoming switch is the sum of the load current and the reverse-recovery current, for load currents close to the rated value, reducing the reverserecovery current has a smaller effect than increasing the turnon time and therefore turn-on losses will increase. Furthermore, the influence of a slower di/dt on diode reverse-recovery losses depends heavily on the rate of change of reverse recovery current versus the rate of change of the turn-on time. The application areas of RB-IGBTs are dictated by the cost of power semiconductors and by the amount of total losses, that is, conduction and switching. Since it is hard to obtain information about the cost of the device, we may only estimate that production cost will be smaller since only one silicon chip is needed instead of the two required in the discrete implementation, together with fewer connections. Furthermore, IXYS currently only produces 50-A current-rated and 600–800–1000–1200-V voltage-rated RB-IGBT devices [4]. Thus, at this stage, it is difficult to generalize the device switching performance to other current ratings to get an idea about the switching losses of a given converter topology at different power levels. The preceding discussion identifies that the RB-IGBT device has a lower voltage drop in the ON state but can have higher

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Fig. 4. Matrix converter-fed induction motor drive where the bidirectional switches are implemented with (a) discrete RB devices (CC-connected IGBTs) and (b) antiparalleled RB-IGBTs.

turn-on losses under some conditions. Hence, it appears to be better suited for converter topologies where the switching losses are low or where the turn-on of one device does not occur against the turn-off of another device. Some converter topologies that may benefit from using the RB-IGBT can be identified on this basis, namely: • the matrix converter [1]–[4]; • the bidirectional power flow current source converter, which may be able to provide sinusoidal input currents and a sinusoidal output voltage while requiring the minimum of 12 semiconductor power devices [4]; • the three-level voltage-source rectifier (VSR), which needs bidirectional switches to connect with the middle point of the split direct current (dc)-link capacitor [5][6]; • the two-stage DPC in the rectification stage [7]–[10]. III. M ATRIX C ONVERTER W ITH RB-IGBT S A matrix converter [1]–[4] is able to provide sinusoidal input currents and bidirectional power flow and does not need bulky dc-link capacitors with limited lifetime in its main power path. It consists of nine bidirectional switches that can connect any of the input phases to any of the output phases. The topology of a matrix converter using bidirectional switches made of antiseries CC-connected IGBTs with antiparalleled diodes is presented in Fig. 4(a). The main disadvantages of the matrix converter are listed as follows: 1) the voltage transfer ratio is limited to 0.866; 2) it has a low immunity to power-grid disturbances; and 3) it normally requires a higher number of power semiconductor devices, that is, 18 IGBTs and 18 diodes. One alternative is to use antiparalleled RB-IGBTs to build the bidirectional switch, to eliminate the need for FRDs, and to reduce the on-state voltage drop. The resulting topology is presented in Fig. 4(b). Bernet et al. [2] investigated the use of RB-IGBTs to build a matrix converter and estimated the losses of a discrete RB device implementation versus the use of RB-IGBTs. In that paper, the reverse recovery losses in the intrinsic diode were realistically considered but the increased turn-on losses in the incoming RB-IGBT due to a higher peak current caused by a poor reverse recovery were neglected since the turn-on/off losses were assigned to the IGBT device and not to a pair of IGBTs and fast recovery/intrinsic diodes. Nevertheless, the work concluded that, depending on the switching frequency, the two alternatives to implement the bidi-

Fig. 5. Three-level front-end ASD topologies with (a) a Vienna rectifier, (b) bidirectional switches implemented with diode-embedded IGBTs, and (c) bidirectional switches implemented with antiparalleled RB-IGBTs.

rectional switch give a different shape of losses. The conduction losses in a matrix converter built with CC-connected IGBTs are higher than with RB-IGBTs, whereas the switching losses increase much faster for the built RB-IGBT matrix converter. This means that at low switching frequencies, a matrix converter built with RB-IGBTs will be more efficient than the one built with CC-connected IGBTs. At a given switching frequency, the losses in the two converters become equal, whereas above this the situation reverses. IV. T HREE -L EVEL VSR W ITH RB-IGBT S Multilevel inverters when used as rectifiers have the advantage of a lower switching voltage ripple across each input phase boost inductance. This allows for a smaller input filter size and lower voltage ratings of the switching devices. For example, a three-level rectifier will halve the size of the boost inductance compared to a two-level rectifier, and since it can use switching devices rated at half the voltage compared to a twolevel rectifier, it may also be able to switch faster. Furthermore, where regenerative operation is not needed, some switching devices can be removed from the circuit, although bidirectional switches are always needed to connect the boost inductor to the dc-link capacitor middle point. Two topologies have been proposed that require the smallest number of controlled switching devices. 1) The Vienna rectifier [5], [6], shown in Fig. 5(a), consisting of 3 IGBTs and 18 diodes, where six of them may be normal diodes. This topology has the disadvantage that the conduction path for the line currents consists of either two diodes, or two diodes and an IGBT.

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Fig. 6. Scheme of a three-level rectifier leg using RB-IGBTs revealing that the reverse recovery process in the intrinsic diode of the RB-IGBT never takes place (a) when Iin > 0, T+ , Cdc+ , and D+ form a boost converter and (b) when Iin < 0, T− , Cdc− , and D− form a boost converter.

2) The three-level VSR [6], shown in Fig. 5(b), where the bidirectional switch is implemented by a diode-embedded switch. This topology requires 18 FRDs and three IGBTs. The line current path consists of either one diode or two diodes and an IGBT. Conduction losses for this topology are reduced as compared with a Vienna rectifier. For both these topologies, at controlled device turn-on, the current through each input phase boost inductor is switched from the diode that clamps the input to the positive/negative dc-link terminal, to the device string that connects to the middle point. Hence, the devices used to build the center bidirectional switches will rarely/never experience reverse recovery, which means that using devices with poor reverse recovery characteristics will not increase the switching losses. This allows the third topology shown in Fig. 5(c) to be proposed, using RB-IGBTs as the center connect devices since the poor reverse recovery characteristics of their intrinsic diode will not increase the rectifier switching losses. Fig. 6 shows this in more detail for a one-phase leg of a three-level rectifier. Assuming the rectifier operates with a unity power factor, for the case of a positive input current when Vin > 0 [Fig. 6(a)], RB-IGBT T+ , capacitor Cdc+ , and diode D+ form a boost converter, whereas the intrinsic diode inside the RB-IGBT is always reverse biased and never experiences reverse recovery. Diode D+ experiences reverse recovery whenever RB-IGBT T+ turns on, but since this diode can be chosen with an appropriate switching behavior, and the switching behavior of the RB-IGBT is similar to a standard IGBT, the switching losses will be similar to the situation where the bidirectional switch is implemented with discrete components. Fig. 6(b) shows the negative input current situation where RB-IGBT T− , capacitor Cdc− , and diode D− form another boost converter. Once again, the intrinsic diode inside the RB-IGBT never experiences reverse recovery. In this case, the reduction of conduction losses compared to the topology shown in Fig. 5(b) is not so important. This is because with a voltage transfer ratio slightly higher than unity, the duty cycle of the bidirectional switch is very small compared to the diode that connects to the positive/negative dc-link terminals, and the input current flows mainly through the diodes. The real advantage is the reduction of the number of switching devices to 12, that is, six RB-IGBTs and six FRDs, which is similar to a two-level voltage source inverter (VSI), but requires only 50% of its voltage rating. It should be noted that no special commutation strategy is required for the bidirectional switches, as is the case for a matrix converter;

Fig. 7. Two-stage DPC ASD topologies consisting of (a) a three-phase to twophase matrix converter linked to a VSI, (b) same where the rectification stage is a 15-IGBT topology reversible current source rectifier, and (c) same as (a) where the bidirectional switch is implemented with antiparalleled RB-IGBTs.

hence, RB-IGBTs that form the bidirectional switch can be switched simultaneously. V. T WO -S TAGE DPC W ITH RB-IGBT S IN THE R ECTIFICATION S TAGE The topology of a two-stage DPC, proposed in [7] and analyzed in [8] and [9], is shown in Fig. 7(a). Functionally, it is identical to a conventional matrix converter controlled by an indirect modulation approach (which now also has a physical meaning), and requires the same number of switches as a matrix converter implemented with unidirectional switches (IGBTs and diodes). A more economical topology can be derived using only 15 IGBTs [9], as seen in Fig. 7(b). Zero-current switching (ZCS) of the rectification stage is also possible [8] when the inversion stage produces a zero-voltage vector. Furthermore, only a diode and a capacitor mounted in the link between the two stages are needed to clamp inductive load currents [8]. Finally, the current rating for rectification stage devices can be customized by taking into account the requirements for power flow at each switching level. For example, for unidirectional power flow, the upper and lower IGBTs can be removed from the rectification stage of the topology shown in Fig. 7(b), [9]. The configuration can also be extended to a multiple drive system by connecting several inversion stages to the rectifier output link [10]. RB-IGBTs can be incorporated into the rectification stage as shown in Fig. 7(c). The advantage offered by this solution is a reduction of the conduction losses. The poor diode recovery behavior of the RB-IGBT is of less concern here than in a matrix converter because it is possible to switch the rectification stage at zero current [8]. Furthermore, if the switching losses of the DPC rectifier stage are compared to a VSR, this stage always

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Fig. 9. Overlap commutation between two RB-IGBTs via an auxiliary current source that avoids a hard turn-on situation in the incoming switch by controlling the reverse-recovery process of the intrinsic diode of the outgoing RB-IGBT. Fig. 8. Two-stage DPC waveforms. (a) Link voltage. (b) Potentials of the twolink poles. (c) Turn-on (positive side) and turn-off (negative side) voltages for the rectification stage.

switches the smallest magnitude line-to line-voltage, whereas the VSR always switches the dc-link voltage (which must be at least as high as the peak line-to-line input voltage). This influences the losses much more than the fact that the current handled by the VSR is always smaller because it processes only active power. Moreover, there are only two commutations in the rectification stage per switching period, as it has to switch between only two of the input lines to achieve sinusoidal input currents. It is useful to calculate the average voltage seen by the switching devices in the rectification stage in the case of unity power factor demand and maximum obtainable dc-link voltage during the switching period. One of the dc-link poles will be permanently connected to the largest magnitude input phase potential. The other dc-link pole, corresponding to the other threephase to one-phase switch group, will switch between the other two input phases (these both have the same polarity). Hence, the line-to-line voltage switched by the rectification stage is small. This may be seen in Fig. 8(a), where the voltage in the dc-link is shown as it is obtained by switching between two of the input line-to-line voltages. In Fig. 8(b), the potential against the ground of the two dc-link poles is presented, showing that only one of the poles is switching between two input lines at any switching period. Fig. 8(c) shows the switching voltage across devices that experience hard switching in the rectification stage. The average switching voltage may be accurately estimated by integrating the line-to-line voltage corresponding to the phase voltages of the lines that are switched for a 30◦ interval (1), i.e.,

Vsw-av


π/6 6 = Vline-pk sin (α) π 0  √ √  3 6 6Vph = 1− π 2 = 0.632Vph .

and approximately 280 V, giving an average of about 140 V. Hence, the expectation is that although the switching losses will increase when using RB-IGBT, it is possible to maintain low-percentage switching losses in the rectification stage of a two-stage DPC because of the low switching voltage stress. However, there is one potential problem with a higher reverse recovery current in the rectification stage as it switches. This is because with an RB-IGBT the reverse-recovery current exceeds the load current and is seen as a circulating current by the input filter capacitors. Such a periodical short circuit of the input filter capacitors has a higher potential to excite oscillations in the input LC filter, and this will contribute to the degradation of the input current quality. Hence, a method to limit/eliminate the reverse recovery peak current is desirable. VI. R EVERSE R ECOVERY C ONTROL OF THE I NTRINSIC D IODE IN THE RB-IGBT Despite its poor reverse recovery behavior, it has been shown that it is possible to use RB-IGBTs in converter topologies where the intrinsic diode does not experience reverse recovery, such as a three-level rectifier. For matrix converters, it may be possible to employ a resonant soft-switching method [11] that will minimize the turn-on losses. For a two-stage DPC, it has been shown that the switching losses in the rectification stage are small either because of the lower average switching voltage or because ZCS is used for the rectification stage [8]. However, it has been reported that with ZCS of the rectification stage, the switches that are normally subject to a hard turn-on may possibly experience partial reverse recovery after ZCS takes place because the carriers in the junction have not been fully removed. A. Controlled Commutation via an Auxiliary Current Source

(1)

For the case of a 230-Vrms phase-to-neutral voltage, the average switching voltage is Vsw-av = 145.4 V, which is much smaller than for the previously presented three-level VSR, which needs 600 V in the dc-link and switches half of this voltage (300 V). This is confirmed by Fig. 8(c), where the switching voltage can be seen to vary almost linearly between zero

The forced turn-on commutation process of an IGBT caused by the reverse recovery of the outgoing diode was described in Section II. Another commutation strategy will now be proposed to compensate for the poor reverse recovery behavior of an RB-IGBT. One method to control the switching process to achieve soft switching is to control di/dt using a snubber or an auxiliary circuit. One good example is the resonant converter technology where an auxiliary circuit delivers a constant current to cover the load current and to achieve soft commutation [11]. Fig. 9 shows how this can be applied to the circuit in Fig. 2. A constant current source, which is normally short-circuited, is

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Fig. 10. Simplified circuit to simulate the proposed method to control the RBIGBT soft turn-on in the rectification stage of a two-stage DPC.

connected to the RB-IGBT emitters by opening Saux . Since the current delivered by the current source is higher than the load current, it forces the current that flows through the outgoing switch to decrease and the output potential to increase from IN1 to IN2. When it reaches IN2, the incoming switch can be activated and the current source short-circuited, ending the soft commutation process. Implementing this auxiliary circuit is not an easy task, and the cost of the converter and its complexity will increase. With an ASD, the load can act as a current source, and it is also known that if the control signals of a VSI that supplies a motor load are disabled, the freewheeling diodes can provide a conduction path for the load currents. This enables the dclink current to reverse direction, irrespective of the operating regime of the motor, and the dc-link current will equal the highest magnitude of the phase currents. Since an induction motor’s stator currents are always higher in magnitude than about 40%–60% of the rated current (because of magnetizing currents), it can be concluded that a two-stage converter inherently has the capability to always provide a negative current source into the dc-link on demand. The new idea proposed here is to use an RB-IGBT in the positive dc-link current path of the two-stage DPC topology rectification stage, which corresponds to motoring operation, and to use the inversion stage as the auxiliary current source circuit, delivering negative dc-link current when the command signals for the inversion stage IGBTs are disabled. Fig. 10 shows the circuit used to simulate the proposed soft turn-on method. The test circuit consists of two RB-IGBTs that connect each input voltage to the dc link (PN), a clamp circuit (diode and a capacitor), and an asymmetric H-bridge that supplies a 15-A current source. The asymmetric H-bridge is emulating a three-phase inversion stage feeding an inductive load. In the first situation shown in Fig. 11, the signals for the H-bridge inverter are continuously on, causing continuous dclink current on both the inversion and the rectification stages. Hard turn-on commutation takes place, and this is clearly seen in the outgoing (IRR = 25 A) and incoming switch (Ion-pk = 40 A) current waveforms. In the second situation shown in Fig. 12, the command signals of the H-bridge are disabled, causing the freewheeling

Fig. 11. Simulation of the commutation between two RB-IGBTs connected in the rectification stage of a two-stage DPC topology causing a hard turnon situation. Diode: IRR = 25 A. Transistor: Ion-pk = 40 A. Time scale: 5 µs/division.

Fig. 12. Simulation of the commutation between two RB-IGBTs connected in the rectification stage of a two-stage DPC topology controlled by the freewheeling of the inversion stage. Diode: IRR = 15 A. Time scale: 5 µs/division.

of the inverter diodes and the reversal of the dc-link current just before the rectification stage commutates. This forces a decrease of current through the outgoing switch as well as an increase of the dc-link voltage to the level in the clamp circuit, which is supposedly higher than any of the input lineto-line voltages. Hence, although the incoming device is turned ON , it will not conduct because its intrinsic diode remains reverse biased. In fact, it will only conduct when the command signals for the inversion stage switches are enabled again and the dc-link current becomes positive. Fig. 12(a) shows that by controlling the reverse recovery process with a current source (the freewheeling inversion stage), it is possible to reduce the reverse recovery current peak to about 15 A, whereas the turnon current peak in the incoming device is completely eliminated. During freewheeling, a certain amount of energy flows into the clamp capacitor, which is used during fault conditions to limit the dc-link voltage below the maximum voltage the devices can withstand. In a way, it can be considered that the poor reverse recovery behavior of the intrinsic diode is hidden by moving the turn-on losses that normally take place in the rectification stage into the inversion stage by having to turn on at a higher dc-link (clamp) voltage, when the IGBTs in the inverter resume operation. However, correction of the output

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Fig. 14. Two-stage DPC with hybrid bidirectional switches in the rectification stage.

To control the reverse recovery of RB-IGBTs that handle the negative direction of the dc-link current, a convenient auxiliary current source as has just been described cannot be implemented. Hence, the two-stage DPC has to use hybrid bidirectional switches in the rectification stage, consisting of an antiparallel connection of an RB-IGBT and a discrete implemented RB device made up of a series of connection of an IGBT and a diode, as shown in Fig. 1(e). The topology of a two-stage DPC drive using hybrid bidirectional switches in the rectification stage in this way is shown in Fig. 14. It provides low conduction losses in motoring operation as well as soft turn-on commutation of the RB-IGBTs, whereas in the rectification stage the standard IGBTs and diodes provide low switching losses in regenerative operation. With this topology, it is possible to obtain soft turn-on of the RB-IGBT using the following switching sequence for the rectification stage. Fig. 13. Experimental evaluation of (a) the standard uncontrolled versus (b) the proposed controlled reverse recovery of the RB-IGBT intrinsic diode. The setup is similar to Fig. 10: Vin1 = 350 V, Vin2 = 20 V, Cclamp = Vin1 , Iload = 15 A, Rg = 22 Ω. Scale: IC (10 A/division), VCE (200 V/division), time (1 µs/division).

voltage due to an undesired voltage state associated with the freewheeling state of the inversion stage is necessary. This principle has also been experimentally tested on a very similar setup to the one shown in Fig. 10, and the results are shown in Fig. 13. The only difference is that the main voltage source Vin1 replaces the clamp capacitor. The signal to disable the gate pulses for the two IGBTs in the inverter stage is produced by a monostable synchronized with the turn-on slope of the incoming RB-IGBT. A slight unalignment between the enable_INV signal and the VCE appears due to the propagation delay in the gate driver and the IGBTs. Fig. 13(a) shows the reverse recovery of the intrinsic diode inside the outgoing RBIGBT, which is reflected in a larger current peak (IRR is approximately 13 A) and a larger width (approximately 250 ns) that is almost cleared when the controlled reverse recovery is activated as Fig. 13(b) shows. There are some parasitic oscillations in the current waveform, most likely caused by the fact that the physical implementation of the test bench was not carefully realized and some parasitic inductance/capacitance exist. Furthermore, when the dc-link current in the asymmetric H-bridge reverses, it still causes reverse recovery in the clamp circuit FRD. B. Controlled Reverse Recovery of RB-IGBTs in a Two-Stage DPC Topology It is seen that this method may be applied only for RBIGBTs that control the positive direction of the dc-link current.

1) The nonconductive outgoing IGBT from the hybrid bidirectional switch is turned OFF [Fig. 15(b)] to disable a possible short circuit of the inputs. 2) The command signals of the inversion stage are disabled [Fig. 15(c)]. The load currents will flow through the freewheeling diodes, forcing the dc-link current to reverse polarity. The dynamic response of this process is slow, dictated by the motor short-circuit impedance, so that it behaves similar to a current source. This provides the reverse recovery current for the intrinsic diode of the outgoing device, whereas the excess current will flow into the clamp circuit and raise the dc-link voltage to the clamp capacitor voltage level. 3) The incoming switch is turned ON [Fig. 15(d)], but the intrinsic diode remains reverse biased since the dc-link voltage equals the clamp circuit voltage, which is normally higher than any of the line-to-line voltage. Hence, the dc-link current continues to flow into the clamp capacitor, which not only protects the converter but now also plays an important role in the commutation process. 4) The gate signals for the inversion stage resume [Fig. 15(e)], and the dc-link current becomes positive again, making the diode in the clamp circuit to turn off and the incoming switch to enter conduction. 5) The nonconductive incoming switch is turned ON, reestablishing a four-quadrant behavior of the hybrid bidirectional switch. In this way, controlled commutation of the RB-IGBT is obtained with small reverse recovery currents and low voltage drop during motoring operation.

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sampling time, it is then necessary to divide the result by the loss simulation time. To calculate the conduction losses, it is necessary that the current through each device and the parameters of the device determined from the manufacturer data sheet are integrated into the loss model. An intermediary temperature characteristic, equally distant to the standard 25 and 125 ◦ C forward characteristics, has been used for each device to obtain a result closer to real operating conditions. The conduction losses are modeled as

Pcond-IGBT =

1

T
sim

  (VCE-sat +rd-IGBT IC )IC dt

Tsim

(2)

0

Pcond-FRD =

1

T
sim

  (VAK-f +rd-FRD IK )IK dt

Tsim

(3)

0

Fig. 15. Controlled commutation of the two-stage DPC rectification stage using a hybrid bidirectional switch. (a) Initial state. (b) Turning off the nonconductive switch. (c) Disable inverter command; load currents flow through the freewheeling diode providing IRR . (d) Controlled reverse recovery ends; all the energy from the load go into the clamp circuit; the incoming device is turned on (still reverse biased). (e) Command of the inversion stage is enabled; the incoming RB-IGBT will now conduct.

VII. L OSS C OMPARISON The power level for evaluating the impact of using RB-IGBT devices was chosen to be 4 kW, which is representative of the low power range. Simulation models for the converter topologies were developed using SIMCAD-Powersim Technologies Inc. However, since this software package uses power semiconductors with ideal characteristics, it is necessary to build another model for the losses to add to the simulation model that is used to investigate the behavior of the power converters. The converter behavior simulation is started first, running until the converter reaches its steady state. Then, the loss model calculator is enabled for a time duration that is a multiple of both the input and the output time periods. Since each model loss calculator integrates the energy loss in each

where VCE-sat and VAK-f are the voltage drops across the IGBT/diode when they are forward biased, IC and IK are the currents that flow through, and rd-IGBT and rd-FRD are the dynamic resistances of the IGBT/diode. The switching losses have been determined in the following way. A pair of switching devices (IGBT and diode mounted in a module) has been chosen to fit the requirements of each topology under study, because usually for a pair of switching devices the manufacturer is able to provide experimental data that show the turn-on, the reverse-recovery and turn-off energy losses as a function of current. The curves given by the manufacturer are converted into an analytical model, and the turn-on and turnoff times are then determined. A commutation detector block included into the simulation model decides whenever a hard turn-on or a hard turn-off occurs, latching the voltage and the current through the device that experiences hard switching and calculating the switching loss energy and integrating this over the loss simulation time. In the data sheet, the switching voltage is constant. The following dependence [1], [12] between the switching loss energy and the current to be switched can be approximated as Esw-on = Aon (IC-on )Bon

Esw-off = Aoff IC-off

(4)

where Aon , Bon include also the contribution of the reverserecovery losses and Aoff are the coefficients determined after applying a curve-fitting algorithm to the data sheet curves. This gives the possibility to estimate the commutation times as a function of the current to be switched, i.e., ton = = toff =

2Esw-on {VCE IC }data sheet 2Aon (IC-on )(Bon-1 ) {VCE }data sheet Aoff = const. {VCE }data sheet

(5)

This allows estimation of the energy loss each time a hard commutation takes place, depending on the actual voltage

KLUMPNER AND BLAABJERG: USING REVERSE-BLOCKING IGBTs IN POWER CONVERTERS FOR ADJUSTABLE-SPEED DRIVES

815

Fig. 16. Distribution of semiconductor losses for the rated power (4.5 kW) in a few converter topologies. REC: rectification stage; INV: inversion stage; conduction losses refer to the IGBT/FRD device and switching losses to their type (ON = turn-on, OFF = turn-off).

across and the current through the device at the switching moment, i.e., Psw-on =

Tsim 1  VCE-on IC-on ton [IC-on ] 2Tsim t=0

(6)

Psw-off =

Tsim 1  VCE-off IC-off toff . 2Tsim t=0

(7)

Losses are integrated separately, which makes it easier to obtain the loss distribution depending on the type of the semiconductor device (i.e., normal diode, FRD, IGBT), the energy loss type (i.e., conduction, turn-on, turn-off), and the converter stage (i.e., rectifier, inverter). Equivalent operating conditions have been considered, which means that the DPC topologies have a lower output voltage and a higher load current and that each converter topology has a different switching frequency to ensure a similar switching stress of semiconductors. The total losses of the converters also include the inverter stage, which, in the case of a dc-link voltage source, switches at 8 kHz. In Fig. 16, the loss distribution of the various converter topologies compared with a diode rectifier VSI topology is shown. The two-level VSC is included in the comparison as it is currently the most used active front-end solution. Other topologies included in this comparison are those in Figs. 4(a), 5(a) and (c), and 7(a) and (c). The matrix converter with RBIGBTs is not included in this analysis because its switching losses will be prohibitive. This can be readily understood if it is recognized that its switching stress is equivalent to a

two-stage DPC topology where all the diodes in both rectifier and inversion stages would experience a poor reverse recovery behavior. Although in the loss distribution it was considered that the two-stage DPC with RB-IGBTs would have similar reverse recovery behavior as a real FRD due to the lack of device data at this power level, its turn-on losses are small (1.5 W) and, as it was shown, alternative techniques to provide ZCS or to control the commutation may be used to further decrease this so that the inaccuracy due to lack of device parameters will not significantly change the result. It is seen that the two-level VSC has the lowest semiconductor losses since it is best fitted for the purpose of the unidirectional switch, but it requires the largest boost inductance that will also cause the largest additional copper and core losses, higher than in a three-level rectifier or in a DPC. It is seen that the semiconductor losses in the two topologies that use RBIGBTs have loss levels similar to those of the two-level VSC, which leads to the conclusion that they are realistic competitors.

VIII. C ONCLUSION Although the switching behavior of RB-IGBTs is not as good as in a discrete implementation, there are already a number of power converter topologies that may benefit from it, and these have been analyzed in this paper. The most important benefit—the reduction of the conduction losses—is illustrated by a loss distribution analysis of a few potential converter topologies for ASD applications, which clearly show that by using RB-IGBTs the overall losses become competitive with the

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two-level VSC. The use of RB-IGBT in a hybrid bidirectional switch in connection with the newly proposed commutation technique was shown to allow the turn-on process of a twostage DPC rectification stage to be controlled. R EFERENCES [1] M. Bland, P. Wheeler, J. Clare, and L. Empringham, “Comparison of calculated and measured losses in direct AC–AC converters,” in Proc. IEEE PESC, 2001, vol. 2, pp. 1096–1101. [2] S. Bernet, T. Matsuo, and T. A. Lipo, “A matrix converter using reverse blocking NPN-IGBT’s and optimized pulse patterns,” in Proc. IEEE PESC, 1996, vol. 1, pp. 107–113. [3] C. Klumpner, F. Blaabjerg, and P. Nielsen, “Speeding-up the maturation process of the matrix converter technology,” in Proc. IEEE PESC, 2001, vol. 2, pp. 1083–1088. [4] A. Lindemann, “A new reverse blocking capability,” presented at the Eur. Conf. Power Electronics Applications, 2001, Paper PP00389, CD-ROM. [5] P. Bialoskorski and W. Koczara, “Unity power factor three phase rectifiers,” in Proc. IEEE PESC, 1993, pp. 669–674. [6] J. Minibock and J. W. Kolar, “Comparative theoretical and experimental evaluation of bridge leg topologies of a three-phase three-level unity power factor rectifier,” in Proc. IEEE PESC, 2001, vol. 3, pp. 1641–1646. [7] J. Holtz and U. Boelkens, “Direct frequency converter with sinusoidal line currents for variable speed AC motors,” IEEE Trans. Ind. Electron., vol. 36, no. 4, pp. 475–479, Nov. 1989. [8] L. Wei and T. A. Lipo, “A novel matrix converter topology with simple commutation,” in Conf. Rec. IEEE-IAS Annu. Meeting, 2001, vol. 3, pp. 1749–1754. [9] J. W. Kolar, M. Baumann, F. Schafmeister, and H. Ertl, “Novel threephase AC–DC–AC sparse matrix converter,” in Proc. IEEE APEC, 2002, vol. 2, pp. 777–791. [10] Klumpner and F. Blaabjerg, “A new cost-effective multi-drive solution based on a two-stage direct power electronic conversion topology,” in Conf. Rec. IEEE-IAS Annu. Meeting, 2002, vol. 1, pp. 444–452. [11] M. Bland, L. Empringham, J. Clare, and P. Wheeler, “A new resonant soft switching topology for direct AC–AC converters,” in Proc. IEEE PESC, 2002, vol. 1, pp. 72–77. [12] F. Blaabjerg, J. K. Pedersen, and U. Jaeger, “Evaluation of modern IGBTmodules for hard-switched AC/DC/AC converters,” in Conf. Rec. IEEEIAS Annu. Meeting, 1995, vol. 2, pp. 997–1005.

Christian Klumpner (S’00–A’01–M’02) was born in Resita, Romania, in 1972. He received the B.Sc. degree in electromechanical engineering from the University of Resita, Resita, Romania, in 1995, and the M.Sc. and Ph.D. degrees in electrical engineering from the “Politehnica” University of Timisoara, Timisoara, Romania, in 1996 and 2001, respectively. From 1996 to 1997, he was with Bee Speed, Timisoara, Romania. Between 1998–2000, he was a Guest Researcher at the Institute of Energy Technology, Aalborg University, Aalborg, Denmark, working on matrix converters, under the auspices of the Danfoss Professor Program. From 2001 to 2003, he was a Research Assistant Professor at the Institute of Energy Technology, Aalborg University, continuing the research in direct power conversion under the auspices of the Innovation Post-Doc Program supported by the Danish Research Agency and Danfoss Drives A/S. Since October 2003, he has been a Lecturer in the School of Electrical Engineering, University of Nottingham, Nottingham, U.K. His research interests include power electronics and ac drives, with special focus on direct power conversion. Dr. Klumpner was the recipient of the Isao Takahashi Power Electronics Award for outstanding achievements in power electronics at the International Power Electronic Conference (IPEC2005) organized by Institute of Electrical Engineers of Japan (IEEJ) in Niigata, Japan.

Frede Blaabjerg (S’86–M’88–S’90–SM’97–F’03) was born in Erslev, Denmark, on May 6, 1963. He received the M.Sc.E.E. degree from Aalborg University, Aalborg, Denmark, in 1987, and the Ph.D. degree from the Institute of Energy Technology, Aalborg University, in 1995. From 1987 to 1988, he was with ABB-Scandia, Randers, Denmark. He joined Aalborg University as an Assistant Professor in 1992, became an Associate Professor in 1996, and a Full Professor in power electronics and drives in 1998. In 2000, he was a Visiting Professor at the University of Padova, Padova, Italy. He was a part-time Program Research Leader in wind turbines at the Research Center Risoe, Denmark. In 2002, he was a Visiting Professor at Curtin University of Technology, Perth, Australia. He is involved in more than 15 research projects with industry, among which is the Danfoss Professor Program in Power Electronics and Drives. He is the author or coauthor of more than 350 publications including the book Control in Power Electronics (Academic Press, 2002). His research interests include power electronics, static power converters, ac drives, switched reluctance drives, modeling, characterization of power semiconductor devices and simulation, power quality, wind turbines, and green power inverter. Dr. Blaabjerg is a member of the European Power Electronics and Drives Association. He was a member of the Danish Technical Research Council in Denmark from 1997 to 2003 and as the Chairman from 2001 to 2003. He was the Chairman of the Danish Small Satellite program and the Center Contract Committee that supports collaboration between universities and industry. He became a member of the Danish Academy of Technical Science in 2001 and of the Academic Council in 2003. From 2002 to 2003, he was a member of the Board of the Danish Research Councils. In 2004, he became the Chairman of the program committee Energy and Environment. He is also a member of the Industrial Drives, Industrial Power Converter and Power Electronics Devices and Components Committee of the IEEE Industry Applications Society. He is an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, IEEE TRANSACTIONS ON POWER ELECTRONICS, Journal of Power Electronics, and the Danish journal Elteknik. He was the recipient of the 1995 Angelos Award for his contribution in modulation technique and control of electric drives, of an Annual Teacher Prize from Aalborg University also in 1995, of the Outstanding Young Power Electronics Engineer Award from the IEEE Power Electronics Society in 1998, of the C. Y. O’Connor Fellowship from Perth, Australia, in 2002, of the Statoil Prize for his contributions in power electronics in 2003, and of the Grundfos Prize in acknowledgement of his international scientific research in power electronics in 2004. He has also received five IEEE Prize Paper Awards during the last six years.