speed up the transient time, which contributes to the higher contrast ratio and luminous efficiency [13]. The most acclaimed ERC [1] is discussed for comparison ...
A Cost Effective PDP Sustainer using Two-Winding Transformer with Hybrid Operation W.S. Kim, S.Y. Chae, B.C. Hyun, D.Y. Lee* and B.H. Cho School of Electrical Engineering and Computer Science, Seoul National University San 56-1, Shillim-dong, Kwanak-ku Seoul, 151-744, Korea *
Consumer IC Design Center, International Rectifier, EL Segundo, CA, USA
Abstract- In this paper, a new PDP sustainer using transformer energy recovery networks is proposed. The twowinding transformer gives the proposed topology several merits, such as a lower number of devices, simple gate driver, ZVS-on of the main switches, ZCS-off of the resonant switches and diodes and high energy recovery efficiency. The turns ratio realizes the voltage-applied overdriving technique as well as the currentinjected overdriving technique. The hybrid driving technique using both methods is researched, which is useful for a faster transition time, hence providing higher contrast ratio and luminous efficiency. The proposed circuit is verified with simulation and experimental results of a 42-inch PDP.
I.
INTRODUCTION Figure 1. Proposed Energy Recovery Circuit
The plasma display panel (PDP) is the representative candidate of emissive displays with high contrast ratio, large screen size, wide viewing angle and rapid response time. The dielectric structure of the AC-PDP brings about the intercapacitance between the sustaining(X) and scanning electrodes(Y) and thus results in an energy loss of CPVS2 during the charging and discharging transients. The excessive surge currents give rise to EMI and increase the device stress of the power switches. A rise in the sustain voltage (VS) for the high definition and single scan driving results in an increase of capacitive loss. To solve these problems, various types of energy recovery circuits (ERC) have been proposed by adopting the zero voltage transition (ZVT) networks [1-11]. However, they have drawbacks such as a complex configuration, high cost and incongruity for large size PDPs with a reasonable operating frequency. In Fig. 1, a new ERC using transformer networks is proposed. In comparison to the earlier approaches, twowinding transformer networks are utilized to reduce the number of devices. The gate driver for the ZVT network is simplified and does not require clamping diodes. Designing the turns ratio gives controllability for the proper resonant energy booster, hence the surge current is reduced by achieving zero voltage switching (ZVS) of the main switches. The hybrid driving technique using both voltage applied mode and current injection mode can easily be implemented to speed up the transient time, which contributes to the higher contrast ratio and luminous efficiency [13]. The most acclaimed ERC [1] is discussed for comparison with the This work is supported by Samsung SDI in Korea ad
1-4244-0714-1/07/$20.00 ©2007 IEEE.
proposed circuit. The operating principle and mode analysis are explained with the energy recovery scheme, and the hybrid operation principle is compared with earlier boost-up methods. The experimental results obtained from a 42-inch PDP verify the characteristics of the proposed ERC. II. TOPOLOGY COMPARISON WITH PRIOR APPROACHES It is worthwhile to look at the main widely referred topology. The left side layout schematic (Y-board) of ERC is shown in Fig. 2. The panel capacitor (CPDP) and the resonant inductor (LR) form a series resonant network with auxiliary capacitors (CSS1 and CSS2) charged to 1/2VS. The induced energy to the panel is softly recovered to CSS2 and is used in the next transition time. However, in practice, there exists non-idealities, such as the parasitic resistance, forward diode voltage drop and wire inductance. Parasitic effects prevent the panel voltage from reaching the input voltage (VS) or ground. It causes the non-ZVS operation of the main switches, and surge current, EMI and poor energy recovery efficiency become important concerns. The proposed ERC consists of full-bridge switches (SXS, SXG, SYS and SYG), ER switches (SXR, SXF, SYR and SYF) and diodes (DXR, DXF, DYR and DYF), and twowinding transformers (TX_X, TX_Y). Using the transformer instead of 4 auxiliary capacitors, the proposed circuit has different energy recovery mechanism and circuit configuration. ZVS operation is possible, and reducing the number of devices solves the cost problems. A detailed comparison is given as follows.
fasdf
294
TABLE I COMPARISON OF NUMBER OF DEVICES RESONANT SOURCE
RESONANT INDUCTOR
SWITCH
DIODE
CONV. ERC
8
8*
4
CAPACITOR (6)
PROP. ERC
8
4
0**
TRANSFORMER(2)
* Include the clamping diode, ** Use the leakage inductor
III. MODE ANALYSIS
(a)
The description of each mode and the key waveforms are shown in Fig. 3 and 4. Only the first half cycle is explained because the other half cycle is the same. Mode 1[t0 ~ t1]: Before t0, SYG and SXG are in the on-state and the voltage across the panel capacitance (VP) is zero. At t0, SYG turns off and the resonant mode starts by turning on SYR. At this time, current starts to flow through the secondary side of TX_Y and DYR conducts, and the secondary voltage of the transformer (VS) is applied to the primary side (VTX_Y_P) in proportion to the turns ratio (= nVS). SYR –VTX_Y_P – LR – CPDP – SXG form the resonant path with (1-n)VS. VCPDP goes to VS in a resonance manner.
(b)
Figure 2. Layout comparison the proposed ERC with the conventional one (a) Conventional ERC, (b) Proposed ERC
A. Resonant energy source In Fig. 2, the resonant energy source of the ERC, CSS1 and CSS2, is substituted by a two-winding transformer. Two capacitors divide the sustain voltage in half and is used for the series resonance between LR and CPDP. In the proposed circuit, the primary side voltage of the transformer, VS, is transferred to the secondary side in proportion to the turns ratio and is used as the resonant energy source.
t ⎡ ⎤ − R sin ωt ) ⎥ (1) VCPDP = ((1 − n))VS − VON ) ⎢1 − e τ (cos ωt + 2ω LR ⎢⎣ ⎥⎦
B. Resonant and energy recovery path During the falling transition time of the panel voltage (VCPDP), the resonant path of the conventional circuit is organized by CPDP-LR-DF-SF-CSS2. The capacitive energy is recovered to the CSS2 and used for the next rising transition. In the proposed circuit, SF-TX_Y_P-CPDP configures the resonant path and recover the energy to the sustain power supply through the transformer secondary side (SF-TX_Y_S-DF). Furthermore, the recovered energy capability can be controlled by the transformer turns ratio.
− (1 − n) VS e τ sin ω t ω LR t
I LR _ Y =
(2)
where R = total parasitic resistance τ = (2 LR / R ) ,
ω = (1/ LR CPDP ) − ( R / 2 LR ) 2 n = turns ratio of Transformer Mode 2 [t1 ~ t2]: At t1, VCPDP clamps to VS and the inductor current (ILR_Y) starts to flow through the body diodes of SYS. At this time, the ZVS-on condition of SYS is achieved. Discharge energy is supplied from the source voltage (VS). After SYS turns on, the remaining ILR_Y linearly decreases. The ZCS-off condition of SYR is achieved when ILR_Y goes to negative.
C. Gate driver of ZVT network In the conventional circuit, all of the ZVT switches (SR and SF) have a floating ground. Using the clamping diode (DR_CLAMP) or two high side drivers solves this problem, but creates a complex and expensive circuit composition. On the other hand, a common hi-lo driver allows for a simple gate driver of the ZVT switches in the proposed circuit.
VCPDP = VS
D. Number of devices Using the transformer recovery network, the main benefit is a decrease in the number of devices. Table I shows the comparison of the number of devices. The body-diodes of the ZVT switches remove the clamping diodes in the conventional circuit (DR_CLMAP and DF_CLAMP). The leakage inductor of the transformer alternates the resonant inductor (LR). In the conventional circuitry, two capacitors can configure the voltage divide source. In practice, a parallel connection of several capacitors is required to reduce resr. A detailed analysis is shown in the subsequent section with a hybrid operation.
I LR _ Y = I LR _ Y (t1 ) −
(1 − n)VS (t − t1 ) LR
(3) (4)
Mode 3 [t2 ~ t3]: Discharge of the panel continues and power is transferred from the source.
295
VCPDP = VS
(5)
I LR _ Y = 0
(6)
Mode 5 [t4 ~ t5]: After VCPDP goes to zero, SYG has the ZVSon condition. SYF achieves the ZCS-on condition after the remaining inductor current (ILR_Y) linearly decreases to zero. VCPDP = VS I LR _ Y = I LR _ Y (t4 ) +
(9)
(1 − n)VS (t − t 4 ) LR
(10)
The opposite side circuit (X-board) operates with the same modes as above. IV. HIGH FREQUENCY CAPABILITY
AND HYBRID OPERATION
In a practical situation, there exists the parasitic inductance and resistance on the board and in the device leads. As reported in [12], the parasitic inductance on the resonant path results in the limitation of transition time. In addition to the limitation of inductance and operating frequency with CPDP, resr of the capacitive resonant source (CSS2) creates a serious problem. The voltage ripple of CSS2 is described as (11). ∆vCSS 2 = 2 I LR _ PEAK sin ω t × resr −
2 I LR _ PEAK
ωC
cos ωt
(11)
where ILR_PEAK = peak inductor current
There are two main factors on determining the voltage ripple of the capacitor, capacitance and resr. By considering the operating circumstances of the PDP, such as operating frequency, large intercapacitance and high peak current, the main factor in defining the type and required number of capacitors is not the capacitance but resr. The bode plot of different types of capacitors and the voltage ripple of CSS with respect to the resr, are shown in Fig. 5. A capacitance of 10uF is large enough for the proper operation, but resr of one capacitor is not suitable for the operation. As a result, a parallel configuration of multiple capacitors is required. Another demerit caused by the capacitive source is the frequency restriction. In the normal operating frequency range, capacitor has resistive characteristic.
Figure 3. The mode analysis of the proposed ERC
Figure 4. The key waveforms of the proposed ERC
Mode 4 [t3 ~ t4]: At t3, SYF turns on and SYS turns off, and the falling transition starts. The resonant source is again (1-n)VS and VCPDP clamps to zero. VCPDP = VS − ((1 − n))VS − VON ) × t −t ⎡ ⎤ − 3 R sin ω (t − t3 )) ⎥ ⎢1 − e τ (cos ω (t − t3 ) + 2ω LR ⎢⎣ ⎥⎦
I LR _ Y = −
− (1 − n) VS e ω LR
t −t3
τ
sin ω (t − t3 )
(7)
(8)
Figure 5. Caracteristic of different types of capacitor and ripple voltage variation w.r.t. resr
296
As shown in Fig. 5, an expensive capacitor type with superior high frequency characteristics must be used. In the proposed topology, the transformer is used for the resonant source, and thus it has an advantage in high frequency operations. There is another advantage in using the transformer resonant network. As mentioned above, there exists a limitation of transition time due to the parasitic inductance. To boost up the speed, the current injection mode type [7-10] is easily used because it doesn’t require a special structure or additional devices. It can achieve ZVS of the main switches as well as fast transition time at the cost of conduction loss. Using the transformer turns ratio, the resonant source voltage level and build-up current (IBUILD_UP) can be changed as well as shown in (12) and (13). For the high frequency operation, a hybrid operation mode can be applied to the proposed circuit. A proper ratio between the injected current and the turns ratio can make a faster rising time possible with less loss and current stress of the devices. t ⎡ ⎤ − R sin ω t ) ⎥ + VCPDP = ((1 − n))VS − VON ) ⎢1 − e τ (cos ω t + 2ω LR ⎢⎣ ⎥⎦ Ibuild _ up
e
ωCPDP
−
(a)
(b)
t
τ
sin ω t
(12)
− (1 − n) VS e τ sin ω t + ω LR t
I LR _ Y =
I build _ up e
−
t
ω
(cos ωt −
R sin ωt ) 2ω LR
(c)
(13)
Figure 6. The hybrid operation in proposed ERC; (a) Hybrid mode before t0, (b) Equivalent circuit during hybrid mode and (c) VCPDP and ILR_Y with three different methods.
V. EXPERIMENTAL RESULTS
switches are turned on. Using a sufficient turns ratio (4:10), the resonant voltage rises to the VS, and hence the full ZVS operation is achieved as shown in Fig. 7(b). The ZVS-on condition gives a lower total consumed power with an increase in the number of pulses as shown in Fig. 8. The hybrid operation experimental results (dots) are shown with the simulation results (lines) in Fig. 9. The difference between the simulation and experimental result is considered as the core loss of the transformer. The hybrid operation is compared with the current injection method and voltage applied method respectively. With considering the practical limitation of reducing the total resonant inductance and transition time, the hybrid operation with a turns ratio of 4:9 shows the lowest loss in the available transition time of 200 to 350nS.
To confirm the operation of the proposed circuit, an experimental system was fabricated and the device parameters are as shown in Table II. 150nH of leakage inductance is used for the resonant inductor with a 200 kHz operation condition. Fig. 7(a) and (b) show the pane voltage (VX and VY) and the inductor current (I LR_Y). In Fig. 7(a), the rising voltage waveform could not reach to the VS due to the parasitic effects and hard voltage transition that appears after the main TABLE II DEVICE PARAMETERS Main and
IRFP90N20 (200V, 90A, Rds_on =0.04Ω )
ER switch Diode
DSEK60-06 (600V, 60A, Vforward = 1.7V)
Regenerative
EI40, n = 0.5 (4turns : 8turns), 0.44 (4:9), 0.4
Transformer
(4:10), 0.375 (3:8), 0.25 (2:8), Litz winding (14/16,
VI. CONCLUSION A new two-winding transformer-type PDP sustainer is proposed. Compared with the conventional one, the proposed topology has the merits of simple structure, reduced number of devices, full ZVS-on of the main switches and ZCS-off of the ZVT switches. The hybrid driving technique results in lower loss than from using only the current injection mode or the voltage applied-mode. Hence, it is advantageous in terms of
2 parallel winding technique ), k ≥ 0.997 Resonant
Leakage inductor of the RT (150nH for 200kHz)
Inductor Driving IC
IR2110 (hi-lo driver)
Panel
42inch PDP (SD type)
297
faster transition time, thus providing higher contrast ratio and luminous efficiency.
Figure 9. Hybrid operation with different driving methods
(a)
REFERENCES [1] [2]
[3]
[4]
[5] (b) Figure 7. Experimental waveforms of the proposed ERC (a) VX, VY and ILR_Y with non-ZVS condition (4:8), (b) VX, VY and ILR_Y with ZVS condition (4:10)
[6]
[7]
[8]
[9]
[10]
[11]
[12]
[13]
Figure 8. Comparison the Input current with number of sustain pulses
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