Design of a Compact Dual-Polarization Receiver for Pseudo-Correlation Radiometers at L-band. I. Ramos-Perez, X. Bosch-Lluis, A. Camps, J. F. Marchan-Hernandez, R. Prehn and B. Izquierdo. UPC Campus Nord, Buildings D3-D4, tel. +34+934017362, e-mail:
[email protected] E-08034 Barcelona, Spain Abstract—This paper describes the receiver that has been designed for the PAU instrument [1]. The design’s main challenge has been the integration in a single 7 cm x 11 cm x 3 cm box of a receiving unit consisting of 4 channels (two per polarization) and merging two different sub-systems: an L-band radiometer (PAU-RAD) and a GNSS-Reflectometer (PAU-GNSSR) [2].
I. INTRODUCTION A general description of the PAU instrument, its objectives and topology is provided in [1]. Before entering in the design details of the front-end, it is necessary to set the specifications for each instrument, and thus determine how a single receiver could be designed to satisfy the following objectives: PAU-RAD: •
Operating frequency: maximum sensitivity for sea surface salinity (SSS) estimation is at L-Band (1-2 GHz), in particular the 1400 - 1427 MHz band is protected for passive observations.
•
Polarimetric radiometer: the instrument’s outputs must be the four Stokes parameters [3].
•
Input signal level: -110 dBm.
•
Number of antennas: 4x4 antenna array.
•
Signal quantification: minimum 8 bits, to achieve > 20 dB side lobe level, and main beam efficiency (MBE) > 95 % [4].
•
Automatic calibration of gain and phase, and capability to synthesize a beam with suitable properties at variable incidence angles. PAU-GNSS-R:
•
Operating frequency: 1.57542 GHz (L1 of GPS signal).
•
Continuous data acquisition to track the coarse acquisition (C/A) code.
•
Input signal level: -133 dBm (spread spectrum).
•
Number of antennas: central 2x2 antenna array.
•
Signal quantification: 1 bit, enough to work the C/A code (sign bit).
•
Automatic phase calibration to synthesize a beam with suitable properties pointing to the specular reflection direction.
A trade-off between the above mentioned specifications was necessary to successfully develop a single receiver shared by both PAU-RAD and PAU-GNSS-R instruments. In order to study the design of the receiver, this article is organized as follows. In Section II, the main restrictions of the design are briefly determined, the operation frequency, the receiver’s topology and dimensions. The use of a GPS front-end as downconverter due to size requirements and other components is presented in Section III. Then, the actual receiver implementation is shown in Section IV, and some measurement results obtained from assembled device are presented in Section V. II.
DESIGN RESTRICTIONS
A. Frequency of Operation The receiver operating frequency is defined by the L1 signal of the GPS signal (1.5742 GHz), which is also suitable for SSS estimation. On one hand the PAU-GNSS-R works with a spread spectrum signal that, due to the scattering on the sea surface, is at least 23 dB below the PAU-RAD one (thermal noise). For this reason, thanks to the 30.1 dB correlation gain, PAU-GNSS-R can detect the GPS signal when the correct C/A code is applied. On the other hand, from PAU-RAD point of view, the noise signal that we want to detect is at least 23 dB above the GPS signal so the radiometric error induced is minimum, and it only occurs in the directions of specular reflection which are known. Therefore it is possible that both PAU-RAD and PAU-GNSS-R can share the same receiver. B. Receiver’s Topology Focusing on the receiver’s structure, the instrument stability required to measure the sea salinity with an L-band radiometer leads to a Dicke or noise injection topologies to compensate for gain fluctuations. However, PAU-GNSS-R requires continuous data acquisition, and the input cannot be chopped. Therefore, a new pseudo-correlation topology –to authors’ knowledge- has been devised. The receiver has two radiometers, one for each polarization (vertical and horizontal), and each one has two channels: one with the antenna and the reference noise signals in phase (1), and the other one 180º out of phase (2). The reference noise signal is introduced by the 100 ohm resistor of a Wilkinson power splitter used to divide the input signal [1]. Afterwards each chain is individually amplified, down-converted and amplified again to allow a 8-bit quantification for later digital processing (calibration, beamforming, and cross-correlation).
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Since the noise introduced by the amplifiers is uncorrelated (apart from the cross-talk), it vanishes when the output signals are cross-correlated (3).
Figure 1. Receiver’s topology block diagram
S H ,V + SR S acH1,V = A + n1 ⋅ g11/ 2 2
(1)
S H ,V − S R S acH ,2V = A + n2 ⋅ g1/2 2 2
(2)
S
H ,V out
(S ) + (S ) = H ,V A
2
R
2
2
g1 g 2
(3)
C. Array’s Dimensions One of the goals of the array design is to cancel the antenna cross-polar component. A symmetry pattern can be observed around the central point of the array, both in vertical and horizontal dimensions.
III.
USE OF GPS FRONT-END AS DOWN-CONVERTER
Taking into account that the working frequency is L1 of GPS and that many components were already designed for its commercial applications, some of them have been used for the implementation of this receiver. One that helped to fulfill size requirements has been the use of a GPS front-end as downconverter. Usually these devices provide the output at intermediate frequency (IF) digitized to 2 bits (sign and magnitude) enough to recover the C/A code, used in GPS applications. First, it was considered to determine the signal’s power using 1 bit as in [5], but with this coarse sampling the array pattern presented side lobes at a level of only 5 dB, , totally unacceptable for radiometric applications. For this reason it was determined that a quantification of 8 bits was necessary to obtain a level of 20 dB of side lobe levels [4]. A commercial GPS down-converter circuit was used. The one selected was the Zarlink GP2015 down-converter that has an analog output test pin that will be sampled at 8 bits. This solution forced to manually set the AGC in order to obtain a linear response, as seen in Fig 3. For input levels from -85 dBm to -60 dBm there is a linear behavior, with a gain of 52 dB, and with a maximum linearity error of 0.25 dB. A factor to bear in mind is the high noise temperature dependence that presents the down-converter. To minimize this effect the datasheet recommended an input gain higher than 20 dB, therefore the noise figure is limited by the first element of the signal conditioning stage, according to Friis’formula. At the receiver’s output the signals are centered around 4.309 MHz with a 2.6 MHz bandwidth, to make the A/D conversion easier.
The receiver size (11 cm x 7 cm x 3 cm) is limited by the separation between antennas, which is found to be 0.63λ to achieve a MBE around 95%.
Figure 3. Output power versus input power for front-end GP2515 with manually set AGC for obtaining a linear behaviour at 25ºC
Figure 2. Arrangement of the 4 x 4 receivers inside the array
At this point of the design the problem dealing with the limited surface available arises.
The second key point in terms of surface availability is the implementation of the Wilkinson power splitter. At the operating frequency (1.57542 GHz) and using transmission lines the area required would have been unacceptably large approximately 5 cm (corresponding to λ/4). Therefore, an implementation with lumped elements has been selected to minimize the occupied area.
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IV.
RECEIVER IMPLEMENTATION
In this section the structure of PAU receiver is defined. This will be determined for the design parameters of each module to implement: receiver’s topology, calibration purpose, isolation among desired signals, signal conditioning stages, wire’s type etc. The block diagram corresponding to the receiver design is presented in Fig. 4. The receiver was carefully designed to preserve symmetry, minimize cross-talk and interconnection routes, and maintain equal delays for all ways, being divided in two parts depending on the operation frequency, RF or IF. For space reasons the two stages (RF and IF) have been implemented one above other using different substrate according to the frequency.
source. It is necessary to compensate different phases and amplitudes among channels. To implement this structure two switches model RSW-2-25P have been used, one for each radiometer, increasing the isolation between correlated noise and antenna signal. Once selected, the signal is input into a Wilkinson power splitter, dividing the signal in two paths. These signals are amplified to adjust the antenna input power of -110 dBm, to the down-converter linear behavior power margin. In order to minimize the noise figure of the receiver, the first amplifier is a low noise amplifier (LNA), followed by a band-pass filter centered to 1.575 GHz, and next, a second amplifier to obtain the necessary gain, see Fig 4. The total RF gain is indicated in (4), obtaining an input power level for down-converter of -76.65 dBm, inside the linear behavior. GRF = L1 + L 2 + A3 + L 4 + A5 ≈ 33.35 dB
(4)
As it has been determined previously, a gain higher than 20 dB is needed to the down-converter input, to minimize the dependence of the noise factor, so that the receiver noise factor is given by the RF stage, see (5), corresponding to a noise figure of 5.3 dB. FeqRF = L1 + L1 ⋅ ( F 2 − 1) + L1 ⋅ L 2 ( F 3 − 1) ≈ 3.4 (5)
Another point to be taken into account is the high isolation among signal conditioning stages necessary to obtain a proportional output to Ta–Tph, being Ta the antenna temperature, and Tph the physical temperature of the Wilkinson power splitter resistor. For this reason, these stages are isolated obtaining a cross-talk of -40 dB.
Figure 4. PAU receiver block diagram
Where: L1: antenna loss + switch RSW-2-25P + two dc block (DC), L2: Power loss (DS52-0004) + own architecture + DC, F2: Noise figure of power splitter F3: Noise figure of LNA (AM50-0002) A3: LNA gain L4: Band-pass loss filter A5: Amplifier gain A. RF Stage The RF stage is implemented with ROGER 4003 substrate to minimize losses. Its structure consists of two parts, first one a selection and then an amplification stage. The selection stage is controlled externally by the Control Unit and switches between antenna acquisition and calibration purposes. Usually, this stage is connected to antenna acquisition, necessary for GNSS-R application. To carry out the calibration process two signals are needed: uncorrelated noise, generated internally by a matched load at each input radiometer, to compensate instrumental biases, and a two level correlated noise, same for all receivers, generated externally from a common noise
B. IF Stage The IF stage is implemented with FR4 substrate. The output of each RF channel is interconnected by means of semiflexible cable to the input of each downconverter. This moves the information from 1.57542 GHz to 4.3 MHz amplifying the RF signal approximately 52 dB. To put the downconverter into operation, a common external TTL signal of 10 MHz is necessary. This signal is used to internally synthesize its different local oscillators through a coaxial cable. The output of each down-converter is amplified by a video amplifier, NE592D, with fix gain and differential output in order to cancel the common mode error. Due to component tolerances is necessary to carry out a relative independent calibration to each channel by means of the voltage divider, Radj. It is located between the down-converter and the video amplifier (Fig 4). The goal of the voltage divider adjusting gain to obtain a 110 mV rms signal at the receiver output, necessary for the ADC input. The gains of each stage should have a maximum error of 1 dB to make possible the correction in the digital process part. The output of each video amplifier is connected to an adaptation network to adapt its output impedance to the twister pair (RJ45 grade 5 cable) one. This connector is shared by four differential outputs at 4.3 MHz, two RF switch control inputs and two power supplies. Due to the amount of cables two RJ45 connectors are needed. This method minimizes considerably the number of cables connected to the receiver, and at the same time gets a high insulation against possible interference.
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V. RECEIVER INTEGRATION AND MESUREMENT RESULTS A photograph of the receiver assembled without box is shown in Fig .5. Both stages are assembled for minimize area, (top) IF stage and (bottom) RF stage. The receiver is introduced into a metal box of (11 x 7 x 3 cm) to protect against undesired signals, shown in Fig. 6.
Figure 5. PAU receiver without box
Figure 6. PAU receiver with box
6 K to 400 K. At last, the measured sensibility is 109.2 µV/K. Fig. 9 shows the scheme assembly used, and Fig. 10 shows different responses for each input stimulus.
Figure 9. Schematic to test receiver’s sensibility.
Table I summarizes the main measured results outstanding the noise figure and receiver gain. TABLE I.
SUMMARY OF MEASURENT RESULTS
Parameter
Theoretical value
Measured value
GRF
< 20 dB
33.35 dB
NFRF
< 5.3 dB
4.65 dB
Crosstalk CHX
< -40 dB
< -43 dB
-
From -85 dBm to -60 dBm
-
0.25 dB
Downconverter linear behaviour Downconverter Max lin error
GReceiver
-
121 dB ± 1 dB
In order to check the correct operation of the receiver two tests have been made, one for GNSS-Reflectometer and another for radiometer instrument. The first one consists on pointing the receiver to the sky looking for satellites obtaining satisfactory results, shown in Fig. 7 and Fig. 8.
Figure 7. Crrelation between a satellite signal and training signal
Figure 8. Satellite # 15, located on 2/12/04
The second is determining the sensibility of the receiver. This is normally done with a Thot and Tcold calibration. In our case, due to low receiver sensibility, it is necessary to use another method. This one consists of introducing different level noise signals at the device input by mean of a variable resistor. With this method it can only be measured a temperature range from 307 K to 400 K because of the thermal noise introduced by the variable resistor, instead of the expected normal range from
VI.
Figure 10. Output responses for each input stimulus.
CONCLUSIONS
The elementary receiver for the PAU instrument has been designed, manufactured and measured obtaining satisfactory results. The receiver has been designed to be shared by the two instruments PAU-RAD and PAU-GNSS-R completely, choosing a common topology and operation frequency. To fulfill size requirements the downconverter function is implemented with a front-end of a GPS receiver chip. In addition, other elements already developed for commercial GPS applications, such as LNAs, power splitters, filters, etc have been employed in the receiver construction. ACKNOWLEDGMENT This work, conducted as part of the award “Passive AdVanced Unit (PAU): A Hybrid L-band Radiometer, GNSSReflectometer and IR-radiometer for Passive Remote Sensing of the Ocean” made under the European Heads of Research Councils and European Science Foundation EURYI (European Young Investigator) Awards scheme in 2004, was supported by funds from the Participating Organisations of EURYI and the EC Sixth Framework Programme. REFERENCES [1] Camps, A., J. F Marchan-Hernandez, I. Ramos-Perez, X. Bosch-Lluis, “New radiometer concepts for ocean remote sensing: Description of the PAU (Passive Advanced Unit)”, proceedings of the IGARSS 2006, Denver, Colorado, July 31-August 4, 2006. [2] Marchan-Hernandez, J. F. , I. Ramos-Perez, X. Bosch-Lluis, A. Camps, R. Prehn, “FPGA-based implementation of DDM-generator for GPSreflectometry,” proceedings of the IGARSS 2006, Denver, Colorado, July 31-August 4, 2006. [3] Bosch-Lluis, X., A. Camps, J. F. Marchan-Hernandez, I. Ramos-Perez, R. Prehn, “FPGA-based implementation of a polarimetric radiometer with digital beamforming,” proceedings of the IGARSS 2006, Denver, Colorado, July 31-August 4, 2006. [4] Bosch-Lluis, X., “Disseny i implementació en FPGA d´un radiòmetre polarimètric de pseudo-correlació amb digital beamforming i autocalibració,” Final project, Universitat Politècnica de Catalunya, Barcelona, Spain, 2005. [5] Martin-Neira, M., Pironen, P., Camps, A., Patent “Digital rms-voltage detector”, Application number: 03290648.9; Date: 28/3/2002; Published on 1/10/2003, in bulletin 2003/40.
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