Design Consideration of SAW/BAW Band Reject Filters ... - IEEE Xplore

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IEEE TRANSACTIONS ON ULTRASONICS, FERROELECTRICS, AND FREQUENCY CONTROL, VOL. 64, NO. 9, SEPTEMBER 2017

Design Consideration of SAW/BAW Band Reject Filters Embedded in Impedance Converter Yulin Huang, Student Member, IEEE, Jingfu Bao, Member, IEEE, Gongbin Tang, Student Member, IEEE, Qiaozhen Zhang, Student Member, IEEE, Tatsuya Omori, Member, IEEE, and Ken-ya Hashimoto, Fellow, IEEE

Abstract— This paper discusses design of surface acoustic wave/bulk acoustic wave (SAW/BAW)-based band reject filters composed of the impedance converters, where capacitive elements are replaced with SAW/BAW resonators. First, basic properties of the unit cell are studied. It is shown how basic properties of a unit cell change with the design. It is also shown that when two notches caused by the resonators are placed in proximity, two synergy effects occur: 1) an extra matching point appears on one side of the transition band. This makes the insertion loss at the point smaller and the transition band steeper and 2) the dip level becomes deeper, and the total rejection level improves. Then, two resonators are fabricated, measured, and combined with inductors in circuit simulator to demonstrate functionality of the basic cell design. Finally, the wide rejection band filter is designed by cascading multistages, and effectiveness of the device configurations is demonstrated. Index Terms— Band reject, filter, impedance converter.

I. I NTRODUCTION

W

ITH the development of the mobile phone local thermal equilibrium (LTE) technology, frequencies have been increased close to TV channels, which results a large number of potential transmitters in this band. To suppress strong interfering signals which desensitize the receiver and to mitigate the spurious output signals from the transmitter side, low loss, and high-selective band reject filters are highly desired in RF front-ends. Currently, various research has been reported on band reject filters, including distributed structures [1], [2] and waveguide structures based on evanescent mode cavities [3], [4]. Manuscript received November 2, 2016; accepted June 4, 2017. Date of publication June 8, 2017; date of current version August 28, 2017. This work was supported by the National Natural Science Foundation of China and the China Academy of Engineering Physics under Grant U1430102. The work of G. Tang supported by the Japanese Government (MEXT) for the scholarship through the Super Global University Project. (Corresponding author: Yulin Huang.) Y. Huang is with the Department of Electronic Engineering, University of Electronic Science and Technology of China Chengdu 611731, China, and Graduate School of Engineering, Chiba University, Chiba 263-8522, Japan (e-mail: [email protected]). J. Bao is with the Department of Electronic Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China (e-mail: [email protected]). G. Tang, Q. Zhang, and K. Hashimoto are with the Department of Electronic Information and Electrical Engineering, Shanghai Jiao Tong University, Shanghai 200240, China, and also with the Graduate School of Engineering, Chiba University, Chiba 263-8522, Japan (e-mail: [email protected]; [email protected]; [email protected]). T. Omori is with the Graduate School of Engineering, Chiba University, Chiba 263-8522, Japan (e-mail: [email protected]). Digital Object Identifier 10.1109/TUFFC.2017.2713395

Fig. 1. Impedance converters (L-matching network). (a) Step up (Z 2 > Z 1 ). (b) Step down (Z 1 < Z 2 ).

However, they suffer the frequency dependence and large size in the UHF band. Lumped element filters based on surface acoustic wave/bulk acoustic wave (SAW/BAW) resonators occupy a relatively small area and have high attenuation in its stopband due to the high quality factors [5]–[8]. Hartmann et al. [6] produced some of the first publications on SAW notch filters [6]. In 1990, Gopani et al. [7] embedded a Two-Pole Waveguide Coupled Resonator in an all-pass network to implement a notch filter. Enhancement of the stopband width was described in [8], where multiple cascaded SAW resonators are series connected. In these filters, SAW/BAW resonators act as capacitive elements of an all-pass filter in combination with inductors at frequencies far from their resonances, while their resonances are used to create notches. Thus, these filters can be regarded as multiple-stage-cascaded impedance converters shown in Fig. 1, where capacitive elements are replaced by SAW/BAW resonators. Impedance converters are quite often used in RF circuits and modules such as the output stage of power amplifiers [9]. Thus use of SAW/BAW resonators may embed the band reject function in RF modules. Furthermore, since inductors are also quite often used for impedance matching in RF circuits including SAW/BAW filters and duplexers, they may be also used for the same purpose. This paper discusses design of an SAW/BAW-based band reject filter composed of the impedance converters. First, basic properties of the unit cell are studied. It is shown that when two notches are placed in proximity, two synergy effects occur: 1) an extra matching point appears on one side of the transition band. This make the insertion loss at the point smaller and the transition band steeper and 2) the dip level becomes deeper, and the total rejection level becomes better. Then, functionality of the basic cell design is demonstrated using two SAW resonators fabricated on 42-LT. The filter

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HUANG et al.: DESIGN CONSIDERATION OF SAW/BAW BAND REJECT FILTERS EMBEDDED IN IMPEDANCE CONVERTER

Fig. 3.

Performance of L-network with various inductor quality factor.

given by

Fig. 2. Basic L-matching network including SAW/BAW resonators. (a) Step up (Z 2 > Z 1 ). (b) Step down (Z 1 < Z 2 ). (c) Step up (Z 2 > Z 1 ). (d) Step down (Z 1 < Z 2 ).

operation is examined on a circuit simulator in combination with built-in inductors. Finally, the wide rejection band filter is designed by cascading multistages, and effectiveness of proposed design procedure is examined. II. L-M ATCHING N ETWORK W ITH SAW/BAW R ESONATORS Fig. 1 shows the basic cells of impedance converters called the L-matching network, which are also used in the ladder topology band reject filters [10], [11]. In Fig. 1, Z 1 and Z 2 are the input and output impedance, Z s and Z p are the impedance of serial and parallel arms, respectively. Since this structure is asymmetric, the input and output impedance are different in general. Here, Z 1 and Z 2 are set to be pure real and define r as Z 1 /Z 2 . Assume Z s and Z p are pure imaginary, and Z 1 is close to the serial arm. Then the transmission could be lossless when the following conditions are satisfied: (1) Im[Z s ] = ±Z 2 r (1 − r ) −1 −1 −1 Im[Z p ] = ±Z 2 r − 1 (2) where the double signs are in the same order, and r should be smaller than unity so that X s and B p are real. The conditions can be satisfied by setting Z s as inductive while Z p is capacitive and vice versa. Next, let us consider inclusion of SAW/BAW resonators in Z s and Z p . Fig. 2 shows four possible configurations. The parallel resonance in Z s and the series resonance in Z p are used to create notches within the passband of the L-matching network. The configurations Fig. 2(c) and (d) will not be further discussed since they seem less practical. Namely, their notch frequencies are influenced by inductance values, which are not so accurate. First, basic properties of the L-matching network are discussed. The transmission coefficient S21 of the network is

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S21 =

2 Ri Ro−1 (Ri + Z s )(Ro−1 + Y p ) + 1

.

(3)

When the conditions given by (1) and (2) are satisfied, (3) reduces to 1 |S21 |max ≈ (4) Z 1-1 1 + 2 Rs + Z22 G p where Ri = Re[Z 1 ], Ro = Re[Z 2 ], Rs = Re[Z s ], and G p = Re[Y p ]. Since dielectric and ohmic losses are not significant in present SAW/BAW resonators, Rs and G p will be predominantly determined by finite Q of inductors. To study the performance of the filter in passband, SAW/BAW resonators in series and parallel arms are replaced with capacitors, C0s and C0 p , respectively, because the SAW/BAW resonators act as capacitors far away from its resonance and anti-resonance frequencies. Fig. 3 shows the |S21 | of structure in Fig. 2(a) with the inductor quality factor Q L increasing from 25 to 100. The horizontal axis is the frequency deviation from the matching frequency f c , which satisfies the conditions given by (1) and (2). The result indicates that the insertion loss of passband decreases rapidly with the increase of Q L , and the decrease becomes small (less than 0.1 dB) when Q L is larger than 50. Fig. 4 shows the simulated |S21 | of the structures shown in Fig. 2. The Q factor of the inductors was set at 50 based on the conclusion of Fig. 3. In Fig. 4, the left and right sides show results for the structures Fig. 4(a) and (b), respectively. The upper two blocks show variation of the |S21 | characteristic with r when C0s /C0 p is chosen as a parameter. It is seen that with a decrease of r , the passband width becomes narrow and the minimum insertion loss becomes large. This is because when r is small, the impedance matching is possible for a narrow frequency range and becomes sensitive to the inductor Q. The lower two blocks show variation of the |S21 | characteristic with C0s /C0 p when r is fixed. For both configurations, the insertion loss becomes better when C0s /C0 p increases. This is because required inductance value decreases with C0s /C0 p . Next, rejection characteristics near the resonance frequencies are discussed. For the purpose, each resonator is modeled by the simple LCR model also shown in Fig. 2, where

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Fig. 4.


Passband characteristic of L-matching network.

L m , Cm , and Rm are motional inductance, capacitance, and resistance, respectively, and C0 is the clamped capacitance. The resonance frequency fr is given by 1/2π(L m Cm )0.5 , while the antiresonance frequency f a is given by fr (1 + γ −1 )0.5 , where γ (=C0 /Cm ) is the capacitance ratio. The resonance and antiresonance quality factors, Q r and Q a , respectively, are given by ωr L m /Rm and ωa L m /Rm for this case. Two dips occur near the resonance f rp of the parallel resonator and the anti-resonance f as of the series resonator. When f rp and f as are not close to each other, the term Z s Y p in (3) is negligible. |S21 | at these frequencies and −3 dB rejection bandwidths contributed by each resonator are approximately given by 4π f as C0s Ri γ Qa γ |S21 | (at f = frp ) ≈ π f rp C0 p Ro Q r 1 BWs ≈ 4πC0s Ri γ π f rp2 C0 p Ro BW p ≈ γ

|S21 | (at f = fas ) ≈

(5) Fig. 5.

Bandwidth and attenuation of reject band variation with d.

(6) (7) (8)

where C0 p and C0s are C0 of the parallel and series resonators, respectively. BWs and BW p are the bandwidth contributed by the serial and parallel resonators, respectively. This analysis shows rough characteristics of a single resonator which will be used for further design. When these two nulls are placed in proximity, the transition bands of the two resonators will overlap, and a bump occurs. Provided that the resonator Q is somewhat large, the attenuation level Ae is determined by this bump height, which becomes large with an increase in the frequency separation d between the two nulls. On the other hand, the total transition bandwidth BWe is also determined by d. Fig. 5 shows a designed example with the structure in Fig. 2(b), where r = 0.69, γ = 15, and Q r = Q a = 500

while d is swept from 5 to 20 MHz. It is seen that smaller d makes the dip levels deeper and the bandwidth smaller. Anyway, Ae and BWe are in tradeoff. Fig. 6 compares the result shown in Fig. 5 when d is 10 MHz with results when one of the resonators is replaced with a simple capacitor. The topologies [Fig. 6(a.1)–(a.3)] are equivalent with the legends in Fig. 5, i.e., (a)–(c). So is Fig. 6(b.1)–(b.3). These results reveal that the following two synergy effects appear when two nulls are placed in proximity. 1) The upper edge of the rejection band becomes steep for the configuration (a). This is caused by fulfillment of (1) and (2) at a frequency just above the upper edge where the series resonator is capacitive and the parallel one is inductive. As a tradeoff, the lower edge becomes gradual. This effect also occurs to the configuration (b). In this case, the lower edge becomes steep while the upper edge becomes gradual. 2) For the configuration (a), the dip at f as is much deeper than the value given by (5) while that at f rp is mostly


Fig. 6.

Rejection band characteristics of L-networks when two resonators are combined.

Fig. 7.

Performance of L-network with various resonator quality factor.

Fig. 8.

equal to that given by (6). Furthermore, the bump height in the rejection band is lower than the value given by a product of values given by (5) and (6). This is due to the term Z s Y p in (3) is not negligible when the distance between the two nulls is small. These effects also occur to the configuration (b). It should be noted that setting f rp > f as for the case (a) and setting f as > f rp for the case (b) offer negative effects: the bump become higher, and the two edges become unbalanced. Fig. 7 shows the variation of |S21 | of structure in Fig. 2(a) with the resonator quality factor Q from 1000 to 250. With the Q decrease, the passband band edges become round and two notches do shallow. However, Ae and BWe do not change too much as expected, because d remains the same. Sharpness of the upper edge is mainly governed by the maximum value of Bode Q [12]. III. FABRICATION AND M EASUREMENT To demonstrate the basic cell design, two one-port SAW resonators were fabricated, and band reject filters shown

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Measured admittance of the serial and parallel SAW resonators.

in Fig. 6 were composed in a free circuit simulation tool called Quite Universal Circuit Simulator (Qucs) [13] using their measured admittance resonance characteristics. Two resonators A and B were fabricated on 42° YX-LiTaO3 (42-LT) substrate [14]. Copper was chosen as the electrode material, and the thickness was set at 300 nm. The SAW resonators employ the standard short-circuited (SC) reflector— interdigital transducer (IDT)—SC reflector structure shown in an inset of Fig. 8. The IDT has 65 finger pairs while the number of electrodes is 30 for each reflector, The IDT periodicity for the resonators 1 and 2 are 5.854 and 5.697 μm, respectively. Fig. 8 depicts the measured admittance of the two resonators. The solid and dashed curves represent that of the resonators 1 and 2, respectively. Strong resonance can be seen at: 1) 638 MHz and 2) 654 MHz, which are caused by the main SAW mode. Resonance Q of these resonators was estimated as circa 450 from the fitting of these responses to the modified Butterworth-van-Dyke (mBVD) model [15]. Spurious resonances can be seen at: 1) 881 MHz and 2) 912 MHz, which are caused by the bulk wave radiation intrinsic in 42-LT [16].

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Fig. 9. |S21 | of the basic cell circuit based on the mBVD model versus the measured resonators.

Fig. 10. Zoomed-in-view of Fig. 9 with a frequency range from 600 to 700 MHz.

Then the measured S-parameter files were loaded as a “black box” in Qucs, and the basic cell circuit was composed in combination with two built-in inductors. Their inductance was set at 137 nH from (1) and (2), and their Q factor was set at 50 on 600 MHz. In the design, r = 0.69. Figs. 9 and 10 show calculated transmission response |S21 | when the circuit shown in Fig. 6(b.3) was chosen. Here, a pair of resonator 1 is parallel connected to achieve double capacitance. This method is used to reduce the lithography area for each resonator. The rejection band with two dips can be seen at ∼654 and ∼662 MHz, which correspond to the resonance frequency of the resonator 2 and the antiresonance frequency of the resonator 1, respectively. For comparison, |S21 | calculated by using the mBVD model is also shown. In this calculation, the resonance Q was limited to 250 intentionally. Nevertheless, this calculation agrees quite well with the original simulation except the dip depth at ∼653 MHz, which is mainly determined by the anti-resonance Q of the series resonator. Although the resonator Q and assumed inductance Q was low (450 and 50, respectively), achieved insertion loss is relatively small. Another notch is seen at ∼907 MHz, which is caused by the bulk wave radiation. Use of other SAW substrates such as 128° YX-LiNbO3 [17] may relax this problem.

Fig. 11. |S21 | of the basic cell circuit based on the mBVD model versus the measured resonators.

Fig. 12. Zoomed-in-view of Fig. 11 with a frequency range from 600 to 700 MHz.

Figs. 11 and 12 show calculated transmission response |S21 | when the circuit shown in Fig. 6(a.3) was chosen. Here, a pair of resonator 2 is serial connected to reduce capacitance. The rejection band with two dips can be seen at ∼654 and ∼662 MHz, which correspond to the resonance frequency of the resonator 2 and the anti-resonance frequency of the resonator 1, respectively. Similar to the results shown in Figs. 6 and 7, the simulated result using the measured admittance agrees well with the one obtained by using the mBVD model. IV. M ULTISTAGE BAND R EJECT F ILTER D ESIGN Since cascade connection of N-stages generates 2N nulls, their proper allocation enables the rejection band to be wider, deeper, etc. Cascading with mirror inversion makes the input and output impedance identical, and the circuit can be used as an isolated band reject filter. Equations (7) and (8) indicate that the fractional bandwidth of SAW/BAW resonator is very narrow and is limited by γ or the electromechanical coupling coefficient K e2 . The bandwidth can be increased by cascading multiple stages and setting resonance frequencies appropriately. As an


TABLE I D ESIGN S PECIFICATION OF THE BAND R EJECT F ILTER

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First, basic properties of the unit cell are studied. It was shown how the passband insertion loss, rejection bandwidth, and its attenuation level change with the design for a unit cell. It was found that when two notches are placed in proximity, two synergy effects occur: 1) an extra matching point appears on one side of the transition band. This make the insertion loss at the point smaller and the transition band steeper and 2) the dip level becomes deeper, and the total rejection level becomes better. Then two SAW resonators were fabricated on 42-LT, and the filter operation was examined on the circuit simulator in combination with built-in inductors. The simulated result agrees well with the one based on mBVD model, and functionality of the basic cell design was demonstrated. Finally, the wide rejection band filter was designed for the given specification. The rejection bandwidth was expanded by cascading multiple unit cells with different design. The designed performance revealed effectiveness of the design.

R EFERENCES Fig. 13.

Performance of designed band reject filters.

example, here a notch filter for a specification given in Table I is designed. The structure in Fig. 2(b) is selected as the basic cell of the band reject filter. Use of SAW resonators on 42-LT are assumed, and γ is set at 15. The discussion in Section II indicated larger r results in better insertion loss. However, if r is too close to unity, the capacitance given by (1) or (2) will be extremely large and impractical. As a compromise, here r is set at 0.69 which corresponds to C0s = 8.2 pF. Location of resonance frequencies are adjusted so that the bump level is −27 dB, while the number of stages are adjusted so that the required rejection bandwidth is obtained. The designed result is shown in Fig. 13. Two cascading methods are designed. For the design (a), the input and output impedance are both 50 . The mirror inversion is applied, and the adjacent two inductors are combined to one. On the other hand, the input impedance is 50 and the output impedance is 72.4 for the design (b). For both cases, all the requirements given in Table I are satisfied. The maximum insertion losses in 470–603 MHz and 603–653 MHz are 0.93 and 1.90 dB, respectively, for the case (a), and are 0.82 and 1.54 dB, respectively, for the case (b). Note that required Q factor of the resonators is 250 for this specification. The value is quite easy to realize. As indicated in Fig. 13, there are two dips in |S11 | at frequencies close to the rejection band edges. They are the extra matching points mentioned in the last section. These points enhance the steepness of the transition band. V. C ONCLUSION This paper discusses design of a band reject filter composed of the impedance converters.

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Yulin Huang (S’14) was born in Chongqing, China, in 1990. He received the B.S. and master’s degrees in electrical engineering from the University of Electronic Science and Technology of China, Chengdu, China, in 2011 and 2013, respectively, where he is currently pursuing the Dr. Eng. degree with the Department of Electronic Engineering, under the supervision of Prof. J. Bao, and the Ph.D. degree with Chiba University, Chiba, Japan, under the supervision of Prof. K. Hashimoto. His current research interests include the application of MEMS switch in circuit application, fabricating a Terahertz device with MEMS technology, filter design based on SAW/BAW resonator, and acoustic wave coupling theory.

Jingfu Bao (M’06) was born in 1964. He received the B.S. degree from the University of Electronic Science and Technology of China (UESTC), Chengdu, China, in 1986, the M.S. degree in electrical engineering, and the Ph.D. degree in science of circuit and system from UESTC in 1989 and 1996. In 1998, he joined Sony Corporation, Tokyo, Japan, as a Hardware Engineer. He researched the Bluetooth chip and module, Wireless LAN chip and module, 1-Segment system wireless DTV chip and module for handset, etc. After 2005, he joined the School of Electronic Engineering, UESTC, where he is currently a Full Professor. His current research interests include RF MEMS, frequency synthesizer, and linear high amplifier. Dr. Bao has received many prizes including the First-Class Prize of Science and Technology Progress in Sichuan Province, Second-Class Prize and ThirdClass Prize of Science and Technology Progress in ministry of electronic in 1992 and 1995, respectively, the First-Class Prize of Sichuan Province Science and Technology Progress in electronic industry, the Outstanding Youth Prize of Ministry of Electronic in 1995, and the Outstanding Science and Technology Youth Prize in Sichuan Province in 1997.

Gongbin Tang (S’15) was born in Shandong, China, in 1987. He received the master’s degree in mechatronic engineering from Beijing Institute of Technology, Beijing, China, in 2012. He is currently pursuing the Dr.Eng. degree in instrument engineering with Shanghai Jiao Tong University, Shanghai, China, under the supervision of Prof. T. Han, and the Ph.D. degree with Chiba University, Chiba, Japan, under the supervision of Prof. K. Hashimoto. His current research interests include synthesize, design, fabrication, and characterization of novel piezoelectric film-based layered surface acoustic wave devices, and developing common simulation tools for transverse mode analysis.

Qiaozhen Zhang (S’15) was born in Anhui, China, in 1988. She received the master’s degree from the Huazhong University of Science and Technology, Wuhan, China, in 2013. She is currently pursuing the Ph.D. degree in instrument science and technology with Shanghai Jiao Tong University, Shanghai, China, and Chiba University, Chiba, Japan. Her current research interests include the design and fabrication of surface acoustic wave (SAW) devices and wireless SAW sensors.

Tatsuya Omori (M’98) was born in 1967. He received the B.Eng. degree in electronics engineering from Kogakuin University, Tokyo, Japan, in 1989, and the M.Eng. and Ph.D. (Eng.) degrees from Chiba University, Chiba, Japan, in 1991 and 1994, respectively. He joined Fujikura Co. Ltd., Sakura, Japan, in 1994, where he was involved in research and development of optical fiber cable and leaky coaxial cable. He has been a Research Associate with Chiba University since 1998. His current research interests include optical fiber sensors, SAW device design, and preparation of piezoelectric thin films and their applications. Dr. Omori is a member of the Institute of Electronics, Information and Communication Engineers of Japan, and the Institute of Electrical Engineers of Japan.

Ken-ya Hashimoto (M’83–SM’01–F’05) was born in Fukushima, Japan, in 1956. He received the B.S. and M.S. degrees in electrical engineering from Chiba University, Chiba, Japan, in 1978 and 1980, respectively, and the D.Eng. degree from the Tokyo Institute of Technology, Tokyo, Japan, in 1989. In 1980, he joined Chiba University as a Research Associate. In 1998, he was a Visiting Professor with the Helsinki University of Technology, Espoo, Finland. From 1998 to 1999, he was a Visiting Scientist with the Laboratoire de Physique et Metrologie des Oscillateurs, CNRS, Paris, France. From 1999 to 2001, he was a Visiting Professor with Johannes Kepler University, Linz, Austria. From 2005 to 2006, he was a Visiting Scientist with the Institute of Acoustics, Chinese Academy of Science, Beijing, China. From 2009 to 2012, he was a Visiting Professor with the University of Electronic Science and Technology of China, Chengdu, China. From 2013 to 2015, he was the Director of the Center for the Frontier Science with Chiba University. Since 2015, he has been a Visiting Professor with Shanghai Jiao Tong University, Shanghai, China. He is currently a Professor with Chiba University. His current research interests include simulation and design of various high-performance surface and bulk acoustic wave devices, acoustic wave sensors and actuators, piezoelectric materials, and RF circuit design. Dr. Hashimoto is a member of the Institute of Electronics, Information and Communication Engineers of Japan, the Institute of Electrical Engineers of Japan, and the Acoustical Society of Japan. He was appointed as a member of the speaker’s bureau of the IEEE Microwave Theory and Techniques Society. He served as a Guest Coeditor of the IEEE T RANSACTIONS ON M ICROWAVE T HEORY AND T ECHNIQUES S PECIAL I SSUE ON M ICROWAVE A COUSTIC WAVE D EVICES F OR W IRELESS C OMMUNICATIONS in 2001, and a Publicity Co-Chair of the 2002 and 2015 IEEE International Ultrasonics Symposia. He also served as an International Distinguished Lecturer of the IEEE Ultrasonics, Ferroelectrics, and Frequency Control (UFFC) Society from 2005 to 2006, an Administrative Committee Member of the IEEE UFFC Society from 2007 to 2009 and from 2014 to 2016, a Distinguished Lecturer of the IEEE Electron Devices Society from 2007 to 2009, and a General Co-Chair of the 2011 and 2018 IEEE International Ultrasonics Symposia. He received the Ichimura Industrial Award from the New Technology Development Foundation for Development of Optimal Substrate 42-LT for Radio Frequency Surface Acoustic Wave Devices in 2015.