Design, Realization, and Test of a UWB Radar Sensor ...

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REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < structure that does not allow to perform a specific design of the various radar subsystems and to easily adapt the radar to different application scenarios (body movements, subsurface sensing, and so on). In this paper, an analytical model of a UWB range gating radar sensor, including source, antennas, human body scattering, and receiver is presented. The model is used to perform a system design of a radar for breath activity monitoring with sub-millimeter movement resolution and fulfilling the FCC mask. Then, the various radar modules are designed and realized in hybrid technology by using a milling machine, surface mounted devices (SMD) and plastic dual inline package (PDIP) components. Some performance factors of the radar sensor are computed and experimentally verified. In particular, the radar range and ability to resolve target movements are evaluated by using a metallic panel. Finally, a comparison between the radar output signal, when it is monitoring a breathing subject, and the signal produced by a piezoelectric belt is performed.

2

The UWB signal (1) is sent to the transmitting antenna (TX) and, once reflected by the thorax, is captured by the receiving antenna (RX). The delayed output of the RR generator is used to drive a pulse generator (strobe source) whose output acts as the strobe signal for the receiver. If this pulse Vs(t), delayed for a time equal to the antenna-target round trip travel time, is present at the strobe input of the receiver together with the reflected signal Vr(t), the receiver output signal Vo(t) is locked to the thorax wall and, consequently, changes following the thorax movements. This signal, once amplified and filtered (VC in Fig. 1), is analog-to-digital converted for real time visualization or off-line processing. In this section the models related to the radar system (block A in Fig. 1) and the receiver (block B in Fig. 1) are developed and discussed in detail. A. Scheme of the Radar System In order to estimate the received signal when the radar antennas are pointing towards a target, the model reported in Fig. 2 has been considered [19].

II. UWB RADAR MODEL Fig. 1 shows a block scheme of the range gating UWB radar analyzed in this paper.

Fig. 2. UWB radar model.

The model takes into account the UWB transmitter source, the radiation impedances (ZT and ZR) and effective lengths

Fig. 1. Block scheme of the range gating UWB radar.

The first block is a square wave generator with a repetition rate (RR) in the 1-10 MHz range. The generator output is split for obtaining a square wave and its delayed replica. The RR generator drives the transmitter source that produces the UWB pulse (VT) that is typically a nth-derivative of the Gaussian pulse, among which the monocycle (first derivative) is the most used in practical applications [14], [18]. The time behavior of a monocycle with amplitude V0 and standard deviation  is given by: Vg t   V0

e te σ



t2 2 σ2

(1)

In the spectral domain, the monocycle waveform is described by the following equation:

VG f   j 2   2 f V0 2  e e  2   f 2 2

2

(2)

( l TE and l R E ) of the transmitting and receiving antennas, the distance between the antennas and the target (L), and the body scattering. In particular, the presence of the body is taken into account by means of the complex backscattering parameter (c) [19]. The c value depends on the antenna-body distance and its magnitude tends to the square root of the radar cross section (RCS) when the far field condition is satisfied. For near field exposure conditions this parameter has to be measured at the operating distance or obtained from numerical computations. The frequency domain relationship between the open circuit voltage of the transmitter (VG) and the voltage at the output of the receiving antenna (VR), closed on a matched load ZC (equal to the characteristic impedance of the input and output interconnecting lines), is given by: VR (f ) 







VG (f )   e  j 20 L SC  j   0  c 0 l TE l R E 1  T 1  R 2  Zc 4  L2

  

(3) where SC, not included in the model depicted in Fig. 2, is the scattering parameter taking into account the environmental clutter and the coupling between the transmitting and

> REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < receiving antennas, T and R are the reflection coefficients of the transmitting and receiving antennas, respectively, 0 and 0 are the characteristic impedance and the propagation constant of vacuum. By using (3) it is possible to evaluate the input-output frequency response of the radar and to compute the time domain behavior of the received signal Vr(t). It is worth noting that in case a radar system employing a single TX/RX antenna (monostatic radar) is considered, (3) can still be applied with some simple modifications. In particular, SC will have to take into account the environmental clutter and the reflections due to antenna mismatch, l TE  l R E , and T  R . Under the assumptions that transmitting and receiving antennas are well matched (T = R = 0), the respective antenna gains are constant and equal to GT0 and GR0 above a cut-off frequency f0 and null below it, and exploiting the relationship between gain and effective length [20], (3) can be rewritten in the simplified form:  c c  G T 0 G R 0 e j 20 L   jVG f  VR f    f 4  L2 0 

f  f0

In practical situations, the second asymptotic condition is generally verified and consequently (5) becomes: Vr t   V0 G T 0 G R 0 c c 

Vr t   V0 G T 0 G R 0 c σ c σ

e 8 L2

   j ( t  τ)  2 π f σ 2  0   1  Reerf      2 σ      

π

e

2σ

8  L2

e



2  f 0  2 2



(7)

          sin 2 f t 0  2 t     2    2  f 0        t 

Fig. 3 shows the time behaviour of the received voltage computed by using (5) and (7) and assuming V0 = 0.5 V,  = 100 ps, GT0 = 2.5, GR0 = 2.5, c = 0.3 m-1, f0 = 2.5 GHz, L = 25 cm, and  = 0. The figure outlines the good accuracy of the adopted asymptotic approximation. 4

(4) Received voltage (mV)

tc

Equation (5) Equation (7) Strobe

3 2

T

1 0 -1 -2 -3

t  τ 2 

2e

  2  f0   cos 2  f 0 t      2   t     2  f  2 0    

f  f0

where c is the speed of light in vacuum. It is worth noting that the above assumptions are usually satisfied by well-designed UWB antennas. Assuming that the complex backscattering parameter c is characterized by a constant magnitude and linear frequency dependence of the phase, which is a good approximation of the c parameter for a human being (see Section III.B and [19]), the time domain received voltage can be expressed in the following closed form:

3

-1

-0.5

2

0

0.5

1

Time (ns)

(5)

Fig. 3. Time behavior of the received voltage achieved by using (5) and (7). The delayed strobe signal is also reported.

t  2  f 0  ) (7) can  be well approximated by the following simple expression: At all-time instants close to  (i.e

where  is the time delay due to the combined effect of the frequency dependence of the phase of c and of the antennatarget round trip travel time, and erf() denotes the error function [21]. To better understand the time behaviour of the received voltage described by (5), small and large argument asymptotic representations of the error function can be used [21]:

while at time instants far from the peak, the received voltage becomes:

 2z    erf z    z 2 1  e  z 

It is worth noting that, according to (8) and (9), the received voltage shows a damped sinusoidal behavior characterized by a frequency equal to the antennas cut-off frequency f0. A similar result is also reached if a generic nth-derivative of the Gaussian pulse is taken into consideration.

z  1 (6) z  1

Vr t   Vr max cos 2 π f 0 t  τ 

Vr t   Vr max 2 π f 0 σ 2

sin 2 π f 0 t  τ  2 π f 0 t  τ 

(8)

(9)

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> R3. Under these hypotheses, always satisfied with the typical values of the circuit elements [9], solving (10) with initial null conditions ( V10  V3 0  0 ), the voltage across the capacitance C3 evolves according to a transient behavior described by means of the two time constants 1 and 2, reaching a steadystate value V3f. The equations for the evaluation of the value V3f and the time constants can be approximated as follows: V3f ( t c )  

γc T

t c  T / 2



t c  T / 2

Vr t  dt

(15)

where γc 

1 Zc  R on T  2 1 T R off

τ1  R 3

C1 C3 Z T ; τ 2   c  R on  C1  C 3  C1  C3 2  T  

Vo t c  

 π T   2π tc   2 γ c Vr max T0   sin  cos  T π  T0   T0 

(19)

where Vrmax is the received peak voltage and T0 = 1/f0. A radar sensor for breath activity monitoring can be characterised by means of some performance factors such as its delay sensitivity, movement resolution, receiver conversion and noise reduction factors. Concerning the time delay sensitivity of the UWB radar this can be defined as:

St c   ddt Vo t c  c

(20)

and hence: St c  

 2 π tc   π T  4 γc  sin   Vr max sin    T  T  T0   0 

(21)

The maximum value Smax of the delay sensitivity is given by:  π T   T  4γ  Smax  S t c  0   c Vr max sin   4 T     T0 

(22)

If the delay tc is located in time between 0 and T0/2, a good estimation of the actual sensitivity is its mean value given by:

Smean 

and tc is the time delay between the strobe pulse and the input voltage (see Fig. 3) By appropriately choosing the diodes and the signal repetition frequency, the γ c term can be made close to unity. Finally, taking into account that the circuit operates in a differential condition, the output voltage of the receiver is given by:

(18)

Equations (15) and (18) highlight that the receiver output voltage is the average value of the open-circuit voltage at the input evaluated over the time interval in which the strobe is on, showing that the receiver in Fig. 4 is able to correctly reconstruct the amplitude of the received pulse wave. By substituting (8) inside (15), (18) becomes:

(16)

(17)

5

T0 2

 π T  8 γc 2  St c  dt  Vr max sin    T0 0 π T  T0 

(23)

Using (22) and (23), it is possible to derive a design condition to optimize the time delay sensitivity of the UWB radar. In particular, choosing the gate aperture time interval equal to half of the time period T0:

T 

T0 2

(24)

the optimal time delay sensitivities assume the following expressions:

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2, the following result is obtained:

2 Vno V2 T  γ c2 ni  B eq B T

6

(31)

where Beq is the equivalent output noise bandwidth. The equivalent noise bandwidth of the receiver output is related to the previously defined 2 time constant by: B eq 

1 4 τ2

(32)

Substituting (32) inside (31) the following expression is obtained for the noise reduction factor: 2 Vno

Vni2



γ c2 Z  4   c  R on (C1 C 3 )B  2 

III.

.

(33)

SYSTEM DESIGN

In this section, the system design of a radar suitable for breath activity monitoring is developed. The main system requirements are a sub-millimeter movement resolution and the fulfillment of the FCC emission mask. The design flow starts taking into account two UWB antennas matched in the FCC band, and the complex backscattering parameter of a breathing subject. Then, by using the results of Section II, the strobe and monocycle time widths are evaluated together with the monocycle amplitude that fulfills FCC requirements. Finally, the time domain received signal at the antenna feed and at the radar output, produced by illuminating a breathing subject, are estimated. A. Antenna Parameters The two different antennas proposed in [22], [23] have been considered. The antenna in [22] belongs to the class of small element antennas with a drop shape, while the one in [23] is a planar monopole antenna with a half-heart shape. Thanks to their specific radiation properties the first antenna has been used as transmitting antenna and the second one as receiving antenna. In particular, the drop shaped antenna presents a very good fidelity factor allowing the radiation of a field tightly resembling the time shape of the source signal, while the half heart antenna has a high effective length suitable for the receiving section of the radar. Since the receiver requires a balanced input, a second half-heart antenna has been realized with a specular shape with respect to the one reported in [23]. The frequency domain radar model described by (3) needs as input the complex reflection coefficient of the antennas; these have been evaluated by measurements performed with a vector network analyzer (PNA E8363B by Agilent). The corresponding results, reported in Fig. 5(a) and (b), show a good antenna matching in the 3.1 - 10.6 GHz frequency band. The effective lengths of the two antennas have been computed by means of numerical simulations performed with a commercial CAD (Microwave Studio – MWS – by CST).

> REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < magnitude

0.06

phase

simulations

180

0

0.05

-40

-60

-50

-120

(a)

magnitude (m)

0

T

-30

S

0

3

6

9

12

(a)

0.03 0.02

E

60

0.04

0.01

l

magnitude (dB)

-20

S11 phase (°)

11

measurements

120

-10

-60

0

-180 15

-0.01

0

2

magnitude

4

6

8

10

Frequency (GHz)

Frequency (GHz)

phase

0.1

60

-30

0

-40

-60

-50

-120

(b)

phase (°)

magnitude (m)

-20

measurements

0.08

R

120

0.06

(b)

0.04

E

-10

simulations

l

180

11

0

S

S11 magnitude (dB)

7

0.02

-60

0

2

4

6

8

10

0

-180 12

0

2

Fig. 5. Reflection coefficient of the drop-shaped (a) and of the half-heart shaped (b) antennas versus frequency.

The antennas have been excited by means of a uniform plane wave, with a vertical polarized electric field EV, and the magnitude and phase of the voltage vF at the antenna feed point have been evaluated, from which: lE = vF/EV. The frequency behavior of the magnitude of lE is shown in Fig. 6(a) and (b) for the drop and half-heart shaped antennas, respectively. For validation purposes, the antenna effective length has been also measured in an anechoic chamber by using the three antenna method described in [24]. The obtained results, reported in Fig. 6(a) and (b), are in good agreement with the corresponding numerical simulations. B. Complex Backscattering Parameter for a Breathing Subject In order to evaluate the time domain radar responses when the antenna is pointing towards a breathing subject, two breathing phases corresponding to end expiration (EE) and end inspiration (EI) have been considered. Fig. 7 shows the c magnitude and unwrapped phase obtained by measuring, with a vector network analyzer, the S21 coupling coefficient of the two radar antennas with a subject placed 25 cm far from the radar (see Fig. 1). In particular, the reported results have been obtained in the two previously cited conditions (EE and EI) by using (3), where the distance L is computed from the point of the body surface closest to the antenna.

4

6

8

10

Frequency (GHz)

Frequency (GHz)

Fig. 6. Effective length of the drop-shaped (a) and of the half-heart shaped (b) antennas versus frequency.

0.0

2 EI phase EE phase

1.5

-5.0 10

3

1

-1.0 10

4

-1.5 10

4

-2.0 10 10

4

EI magnitude EE magnitude

0.5

0 1

2

3

4

5

6

7

8

9

Frequency (GHz)

Fig. 7. Measured c for a standing subject 20 cm far from the antenna at end inspiration (EI) and end expiration (EE).

C. Pulse Width and Amplitude The results of Section III.A show that the transmitting antenna has a cut-off frequency of about 2.5 GHz, that is higher than the cut-off frequency of the receiving one (about 1 GHz). On the basis of (24), a strobe source with a time width of about 200 ps, corresponding to a Gaussian pulse with  = 100 ps, is requested for the best delay sensitivity and movement resolution. The same standard deviation has been chosen for the monocycle source. With these choices, the

> REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < compliance with the FCC emission mask has been verified adopting the previously described radar equation (3). Numerical simulations have been performed by considering the transmitting antenna parameters, and a monocycle source with a repetition rate of 1 MHz. The computed EIRP, over a 1 MHz bandwidth, are reported in Fig. 8 for different values of the monocycle amplitude (V0). 0

E. Radar Responses In order to estimate the time domain radar output signal when the radar is monitoring the breath activity of a subject placed 25 cm far from the antenna, a FCC compliant monocycle with  = 100 ps, and V0 = 0.5 V has been used as source signal in the radar model in Fig. 2. Fig. 9 shows the received signals, in correspondence of EE and EI respiratory phases, computed by using the inverse Fourier transform of (3). In the simulations the SC term has been neglected, in order to evidence only the contribution due to the breath activity.

-50

3 2

-100

EI EE

1

-150

-200

V (mV)

FCC Mask 0.1 V 1.0 V 10 V 0

2

4

0

R

EIRP (dBm)

8

-1

6

8

10

Frequency (GHz)

Fig. 8. Computed EIRP for a monocycle source with  = 100 ps, RR = 1 MHz and various amplitudes.

Comparison with the FCC emission mask evidences that, in the considered case, pulse amplitudes lower than 1 V can be radiated in compliance with the mask. It is worth noting that this result is scalable in terms of signal repetition rate. In fact, if the RR is increased ten times up to 10 MHz, the pulse amplitude to fulfill the FCC mask must be reduced by a factor equal to the square root of 10. With reference to the FCC limit on peak EIRP over a 50 MHz bandwidth, the EIRP spectrum, for a monocycle pulse of amplitude 1 V, presents a maximum of about -69 dBm over a bandwidth of 1 MHz at about 3 GHz, as shown in Fig. 8. To evaluate the compliance with the peak EIRP limit, the EIRP spectrum must be pass-band filtered around 3 GHz. The signal at the filter output has been estimated multiplying the computed EIRP, shown in Fig. 8, by the frequency response of an ideal 50 MHz filter, centered at 3 GHz, and antitrasforming the resulting spectrum. The computed peak power of the signal is about -32 dBm, hence compliant with the FCC limit of 0 dBm [16], [17]. D. Receiver Parameters The breathing activity rate of a human being is of about 12-24 breaths per minute, corresponding to a fundamental frequency of 0.2 - 0.4 Hz. For this reason a receiver time constant of about 1 ms seems to be a reasonable choice. By using C1= 10 nF, C3 = 1 nF, R3 = 10 k, R4= 4.7 Mand considering that the strobe signal has a repetition rate in the 1 – 10 MHz range and time width of 200 ps, the hypotheses under which (15) is obtained are satisfied and a value of about 1 ms is obtained for the receiver time constant 2 (1 REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT)
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REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < C. Breath Activity Measurements The assembled radar has been used to measure the breath activity of a subject placed about 25 cm far from the radar antennas. For the sake of comparison, the same activity has been measured by using a piezoelectric belt (MLT1132 by ADINSTRUMENTS). The results of a first experiment are reported in Fig. 15(a). The figure shows that both the breathing and the apnea phases are well recorded by the developed radar. Moreover, the observed signal dynamic (about 40 mV peak to peak) is in a good agreement with that predicted using the results reported in Section III.E. 40 Radar Belt

30 20 10

(a)

0 -10 -20 -30 -40 0

20

40

60

80

100

Time (s) 40 Radar Belt

30 20 10

(b)

0 -10 -20 -30 -40 0

5

10

15

20

Time (s)

Fig. 15. Output of the UWB radar as compared with piezoelectric belt signal: (a) breathing activity followed by an apnea phase, (b) regular breathing activity with signal in phase opposition.

Fig. 15(b) shows the results of a second experiment performed with a subject always located about 25 cm far from the radar. It is interesting to note that in this experiment the radar output is 180° out of phase with respect to the piezoelectric belt output. This can be explained by looking at (19) that evidences how a delay in the received output can give rise to an increase or a decrease in the radar output signal, depending on the sign of the delay. The data shown in Fig. 15(b) have been FFT transformed and an average breath frequency of 0.24 Hz has been obtained both for the radar and the belt signals. By increasing the antenna-body distance the radar output signal reduces. If the transmitted monocycle amplitude is

12

chosen in order to satisfy FCC requirements, the maximum distance (radar range) giving a readable signal is of about 60 cm. This range is slightly inferior to that evaluated by using the metallic panel due to the lower c of the human body. Thanks to its specific characteristics the radar can have interesting applications, such as the monitoring of breath activity of astronauts during their rest phase in the crew quarters of the International Space Station [33].

VI. CONCLUSIONS In this paper a model of a ultra wideband radar sensor for breath activity monitoring which takes into account the radar source, the antennas, and a breathing subject at a given distance from the radar has been presented. The model equations have been arranged in a simplified form that evidences for the voltage at the receiver input a dumped sinusoidal behavior with a frequency equal to the highest cutoff frequency of the radar TX-RX antennas. Other important parameters as the values of the lumped elements of the range gating receiver and the strobe and transmitter source time widths and amplitudes can be designed on the basis of the obtained model equations. The design of the various radar subsystems, performed by using commercial CAD tools, has been discussed in detail. The subsystems have been fabricated by using a low cost planar technology and the complete radar system has been assembled. Measurements performed on the receiver allowed to estimate the conversion and the noise reduction factors. Moreover, by using a metallic panel, the radar range and movement resolution have been measured yielding results in good agreement with numerical simulations. The estimated radar range is about 1 m while the smallest detectable movement is lower than 0.1 mm. Finally the radar has been used for remote breath activity monitoring, by placing the radar in front of the thorax of a test subject. The recorded respiratory signals are in very good agreement with those obtained using a contact piezoelectric belt. With respect to previous designs available in scientific literature, the proposed radar adopts a very fast pulse generator based on a step recovery diode for the generation of the pulse used to excite the UWB antenna, and a receiver employing a couple of very fast zero bias Schottky diodes to improve the attitude of the radar to detect the small body movement related to the breath activity. Another advantage of the proposed radar system is its modular structure allowing an easy replacement and improvement of the various subsystems without rebuilding or redesigning the whole radar assembly. For example, the radar range can be improved by adding an amplifier block before the radar antenna and the output signal can be translated to higher frequencies, where the FCC requirements are relaxed, by adding a passive differentiator.

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[2]

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[6] [7]

[8]

[9] [10]

[11] [12] [13]

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Paolo Bernardi (M’66–SM’73–F’93–LF’01) received the Electrical Engineering Laurea and Libera Docenza degrees from the University of Rome “La Sapienza,” Italy, in 1960 and 1968, respectively. Since 1961, he has been with the Department of Electronics, University of Rome “La Sapienza,” where he became a Full Professor in 1976, continuing in office until 2010. From 2011 he is an external scientific consultant with the same department. Dr. Bernardi research activity has dealt with the propagation of electromagnetic waves in ferrites, modeling and design of MW components, interaction of EM waves with biological systems, and electromagnetic compatibility. He has authored or coauthored over 250 scientific papers and numerous invited presentations at international workshops and conferences. He was for many years an Editorial Board member for the IEEE Transactions on Microwave Theory and Techniques, for the Microwave and Optical Technology Letters, and for the AEI Alta Frequenza. During the years 1979 - 1980, he was the chairman of the IEEE Middle and South Italy Section and chapter chair of the IEEE MTT society, with the same section, from 1996 to 2001. He was the chairman of the International Union of Radio Science (URSI) Commission K on Electromagnetics in Biology and Medicine through the triennium 1993–1996. Dr. Bernardi was awarded with the IEEE Centennial Medal during the IEEE Centennial year 1984.

> REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) < Renato Cicchetti (S’83–M’83–SM’01) was born in Rieti, Italy, in May 1957. He received the Laurea degree in electronics engineering (summa cum laudee) from the University of Rome “La Sapienza,” Rome, Italy, in 1983. From 1983 to 1986, he was an Antenna Designer at Selenia Spazio S.p.A. (now Thales Alenia Space S.p.A.), Rome, Italy, where he was involved in studies on theoretical and practical aspects of antennas for space application and scattering problems. From 1986 to 1994, he was a Researcher, and from 1994 to 1998, he was an Assistant Professor at the Department of Electronics Engineering, University of Rome “La Sapienza,” where he is currently a Full Professor. In 1998, 2002, and in 2006 he was Visiting Professor at the Motorola Florida Corporate Electromagnetics Research Laboratory, Fort Lauderdale, FL, where he was involved with antennas for cellular and wireless communications. In 2012-2013 he was the Lead Editor of the Special Issue on “Wideband, Multiband, Tunable, and Smart Antenna Systems For Mobile UWB Wireless Applications” for the International Journal of Antennas and Propagation. His current research interests include electromagnetic field theory, asymptotic techniques, electromagnetic compatibility, wireless communications, microwave and millimeter-wave integrated circuits, and antennas. Dr. Renato Cicchetti is a Senior Member of the Institute of Electrical and Electronic Engineers, of the Italian Electromagnetic Society (SIEm) and he results listed in Marquis Who’s Who in the World and Who’s Who in Science and Engineering. Stefano Pisa (M’91) was born in Rome, Italy, in 1957. He received the Electronic Engineering and Ph.D. degrees from the University of Rome “La Sapienza,” Rome, Italy, in 1985 and 1988, respectively. In 1989, he joined the Department of Electronic Engineering, University of Rome “La Sapienza,” as a Researcher. Since 2001, he has been an Associate Professor with the same university. His research interests are the interaction between electromagnetic fields and biological systems, therapeutic and diagnostic applications of electromagnetic fields, and the modeling and design of MW circuits. He has authored over 150 scientific papers and numerous invited presentations at international workshops and conferences. He serves as a reviewer for different international journals. From 1995 to 2002, he was secretary of the IEEE Microwave Theory and Techniques Society (MTT-S)/Antennas and Propagation Society (AP-S) Central and South Italy Section Joint Chapter. He is currently "Consulting Member” of the “Scientific Committee on Physics and Engineering" of the "International Commission on Non-Ionizing Radiation Protection" and a member of the Advisory Group of the Dutch project “Electromagnetic Fields and Health”. Erika Pittella received the MS (cum laude) and PhD degrees in Electronic Engineering from Sapienza University of Rome, Italy, in 2006 and 2011, respectively. She is currently a research associate with the Department of Information Engineering, Electronics and Telecommunications (DIET), Sapienza University of Rome. Her main research activities are related to the modeling of UWB radars for the remote monitoring of cardiorespiratory activity and to the design of sources, antennas, and receivers of such systems. Her research interests also include dosimetric aspects of the interaction between electromagnetic fields radiated by UWB radar systems and exposed subjects.

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Emanuele Piuzzi (M’09) received the M.S. (cum laude) and Ph.D. degrees in electronic engineering from the Sapienza University of Rome, Rome, Italy, in 1997 and 2001, respectively. He is currently an Assistant Professor with the Department of Information Engineering, Electronics and Telecommunications, Sapienza University of Rome, where he is engaged in teaching electrical measurements. He is the coauthor of over 100 publications. His current research activities include the measurement of dielectric characteristics of fluids and granular materials through time- and frequency-domain reflectometry approaches, impedance pneumography, ultrawideband radar techniques for the monitoring of cardiopulmonary activity in patients, and electrical impedance tomography. He is also interested in the study of the interaction between electromagnetic fields and exposed subjects. Dr. Piuzzi is a member of the IEEE Instrumentation and Measurement Society, of the Italian Group of Electrical and Electronic Measurements (GMEE), of the Italian Electrotechnical Committee (CEI), and of the Italian Society of Electromagnetics (SIEm). He serves as a reviewer for different international journals. Orlandino Testa was born in August 1972, in Minturno, Italy. He received the Laurea degree (cum laude) in electronic engineering degree and the Ph.D. degree from the University of Rome “La Sapienza”, Rome, Italy, in 1997 and 2003, respectively. Since 2001, he has been a high school Teacher at the I.T.I.S. “G. Armellini” Institute of Rome, where he is involved in teaching electronics and telecommunications. He is also currently collaborating with the Department of Electronic Engineering, University of Rome “La Sapienza.” At present, he is studying high-frequency models for the analysis of radio coverage in indoor environments and tunnels with particular attention to EMC/EMI problems. His main research interests are propagation and radiation of electromagnetic fields, electromagnetic compatibility, microwave and millimeter-wave integrated circuits, and antennas.