Direct Torque Control for Sensorless Induction Motor ...

Direct Torque Control for Sensorless Induction Motor Drives Using an Improved H-Bridge Multilevel Inverter Javier Pereda, Student Member IEEE Juan Dixon, Senior Member IEEE

Mauricio Rotella ABB Chile Local Division Manager, Automation Product [email protected]

Pontificia Universidad Católica de Chile, Santiago, Chile [email protected], [email protected]

Abstract — This paper presents the application of a Sensorless Direct Torque Control (DTC) for an induction motor, using an improved H-Bridge multilevel inverter with 27-Levels. The inverter topology reduces the power sources from nine to only four active sources and three ultracapacitors. The power sources are unidirectional and non-redundant; scaled in power of three to optimize the number of voltage levels with a minimum of semiconductors and power sources. For the improved inverter topology, two additional control strategies are introduced; 1) an Inhibit Negative Currents (INC) controller, which solves the regeneration problem when unidirectional sources are used; and a 2) Proportional-Integral (PI) controller, which keeps the ultracapacitor voltages at the reference value. Both controls work on a Pulse Width Modulation (PWM) signal, where the INC control decides the levels among the PWM operates, and the PI controller changes the duty-cycle. A closed-loop estimator called Model Reference Adaptive System (MRAS) was used for the speed estimation, with the advantage of using the stator voltages and currents already obtained for the DTC. The application of the system was simulated and implemented in Matlab®/Simulink® software using the industrial controller AC800PEC from ABB, obtaining satisfactory results. The multilevel inverter was specially designed and built for this application. Index Terms—Direct Torque Control (DTC), Multilevel Inverter, Pulse Width Modulation (PWM), AC Machine Drives, Sensorless.

I.

INTRODUCTION

T

HE squirrel cage induction motor has been preferred by the Industry for years due to its low cost, high performance, reliability, ruggedness, and applicability. Nevertheless, the use of an induction motor in high performance applications requires complex electronic converters and controllers to obtain an adequate control of the motor speed, torque, current, and magnetic flux. Moreover, problems of conventional inverters include noise, harmonic contamination, torque jerk, motor deterioration, and losses in the inverter. The H-Bridge multilevel inverter is a considerable improvement in the voltage waveform of the motor in comparison with conventional inverters [1]. Even more, the number of voltage levels can be maximized if the H-Bridges use asymmetrical power sources scaled in power of three [2].

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The increase in the number of voltage levels reduces: i) the Total Harmonic Distortion (THD); ii) abrupt changes in voltage (dv/dt); iii) voltage and currents of common mode; iv) output filters; and v) switching losses. Switching losses are reduced because the main bridges, which carry 80% of total power, works at very low frequency when power sources are asymmetrical [2-4]. The drawbacks of asymmetric H-Bridge inverters are the large number of bidirectional floating power sources and the lack of modularity. To reduce these drawbacks, a special topology and control strategy was used [5]. This solution simplifies and reduces the number of power sources from nine bidirectional to only four; one source for the three phases of the main H-Bridges, which can be unidirectional or bidirectional, and three unidirectional sources for the intermediate H-Bridges (Aux-1). The smallest H-Bridges (Aux-2) use floating ultracapacitors. The Direct Torque Control (DTC) [6, 7] is considered as one of the strategies of control with the highest performance in AC Drives and is commercially available for conventional inverters. The DTC already has been used in multilevel inverters [8, 9] showing advantages over its application in inverters with fewer levels regarding the switching frequency and torque jerk [10]. The sensorless drives have the advantages to being economic and reliable. They have had an important development and commercial acceptation, and can estimate the rotor speed using the stator voltages and currents, so a tachometer is not necessary. These systems already have been tested in DTC obtaining satisfactory results [11-13]. The purpose of this work is to develop a sensorless DTC drive for induction motors, using an improved multilevel inverter with a reduced number of unidirectional power sources though new topologies and control strategies [5]. II.

MULTILEVEL INVERTER TOPOLOGY

The improved 27-Level inverter used is shown in Fig. 1. Each phase has three sources scaled in power of three (V, 3·V and 9·V) to maximize the number of levels to 27 (33), where V is the voltage of each level. Furthermore, this topology reduced from nine bidirectional power sources to only four unidirectional power sources plus three ultracapacitors.

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The reduction was possible with the addition of a new topology and control strategies, which are explained below. A. Power Source of the Main H-Bridges To reduce the number of power sources of the main converters from three to only one, the three H-Bridges are fed in parallel from only one DC Supply. With this topology, each windings of the motor must be isolated [14]. The main source can be bidirectional or unidirectional. For applications that require regeneration, a main bidirectional source is used and the smaller bridges are inhibited, and the inverter generates only with three levels. If the main converter use unidirectional source, the machine will not be able to operate in regenerative mode by long periods. The problem can be solved adding a reostatic brake at the dc-link. Three secondary transformers shifted in +20º, 0º, and -20º are used to generate minimum harmonics to the network.

controls the inverter variables such as regeneration and ultracapacitor voltages. However, the INC and PI controllers need the voltage reference from DTC to work on the PWM signal. A. Inhibit Negative Current (INC) Control The manipulated variables of the INC control are the levels among the PWM moves (levels jumped) to avoid negative currents at the DC-Link of any Aux-1 H-Bridges. The INC control is an on/off controller and is enabled only when the average power in any source is negative, which is detected by the rise of the capacitor voltage of the DC-Link (VAux-1). The flux diagram of the control is illustrated in Fig. 2, and it works taking binary decisions such as yes/no and using lookup tables with one input and two outputs. When the capacitor voltage (VAux-1) is greater than the setpoint (VAux-1*) the average power in the source is negative, so the INC control will be enabled. The INC avoids those levels where the current is negative (flow from H-Bridge to the power source), which depend on the current motor (Iout). The outputs of the lookup tables are the base level (VINCout) of the PWM signal and the number of levels (Delta) that must be avoided by the PWM signal. If the INC control is disabled, no level is avoided (Delta =1) and the base level is the discretized output voltage from DTC (VDTCout).

Fig. 1. Topology of the multilevel inverter used (27-Level) using an unidirectional main source.

B. Power Source of the Aux-1 H-Bridges Each one of the three sources that feed the intermediate H-Bridges (Aux-1) are unidirectional, independent and supplied from three transformers as the main source. To avoid the destruction of the source by regeneration, a special Pulse Width Modulation (PWM) control was used to Inhibit Negative Currents (INC) from the HBridges to the sources during permanent regeneration. C. Power Source of the Aux-2 H-Bridges The sources of the smallest H-Bridges were replaced by ultracapacitors, which maintain a constant voltage through a Proportional-Integral (PI) controller, regulating the duty-cycle of the PWM signal according to the effect of each level has on the charge or discharge of the ultracapacitors. III.

INVERTER CONTROL TECHNIQUES

The complete system has three controls in series: i) DTC; ii) INC control; and iii) PI controller. The DTC controls the motor variables such as motor speed, torque, current and flux, and is independent of the INC and PI controllers, which

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Fig. 2. Flux diagram: Inhibit Negative Current (INC) control.

B. Proportional-Integral (PI) Controller The manipulated variable is the duty-cycle of the PWM and the controlled variables are the ultracapacitor voltage and the current of the motor at the same time. The PI controller and PWM signal generation are adapted to obtain a multilevel signal with the INC control incorporated, as shown in Fig. 3. The error of the ultracapacitor voltage is the input of the PI controller, and its output changes its sign according to the effect that the PWM has in the ultracapacitor charge. Therefore, the average value of the carrier is controlled and the duty-cycle of PWM changed indirectly. The current control is done through a comparator between the carrier and the voltage reference from DTC. The output of the comparator is a PWM signal, which is adapted to be a multilevel PWM signal. The control uses the base level from the INC control (VINCout) to calculate the effect of the output voltage in the ultracapacitor charge, and to have a base level for the PWM signal, obtaining a multilevel PWM signal.

1111


Fig. 4 shows an example of multilevel PWM when the error of the ultracapacitors voltage is zero and the INC control is disabled (Delta =1). Fig. 5 illustrates the voltage and current of the motor with the control INC disabled and then enabled. When the INC is enabled, some levels are avoided in function of the motor voltage and current sign.

Fig. 3. PI controller of ultracapacitor voltages, control of the motor current, PWM signal generation, INC application and multilevel signal adaptation.

Fig. 4. Multilevel PWM signal with PI controller working and INC disabled.

IV.

DIRECT TORQUE CONTROL

A DTC based on load angle control was used to calculate the reference vector flux [15], which is compared with the estimated vector flux to obtain the voltage vector (Vs), which is transformed by a simple near voltage selector to get the voltage level of each phase. These levels are recalculated by the INC control and then by the ultracapacitor voltages controller before being sent it to the inverter [Fig. 6]. The controlled variables of the DTC are the magnitude of the stator flux and the torque or rotor speed of the motor. The manipulated variable is the vector of the stator voltage, which produces a change in the vector of flux, changing its magnitude and angle. Fig. 7 illustrates the manipulation of the flux vector and all the voltage vectors in the 27-level inverter. A. Flux Estimators The voltage and current vectors (Vs and Is) are calculated through a transformation from three phases to a stationary reference frame of two coordinates (α and β) as shown in (1) and (2), where vsA, vsB, and vsC are the stator motor voltages and a is an unitary vector (a=ej2·π/3).

(

)

(

)

Vs =

3 ⋅ v sA + a ⋅ v sB + a 2 ⋅ v sC 2

Is =

3 ⋅ i sA + a ⋅ i sB + a 2 ⋅ i sC 2

(2)

The equations that describe the model of the motor in the stationary reference frame are (3)-(7), where Ls, Lr and Lm are the stator, rotor, and mutual inductances; Rs and Rr are the stator and rotor resistances; Ir is the rotor current vector; Ψr and Ψs are the rotor and stator flux; and wm is the rotor speed. V S = Rs ⋅ I s +

dψ s dt

dψ r − j ⋅ wm ⋅ψ r dt ψ s = Ls I s + Lm I r ψ r = Lm I s + Lr I r

0 = Rr ⋅ I r +

Fig. 5. Motor voltage and current with PI controller working and INC control disabled and then enabled.

(1)

(3) (4) (5) (6)

The estimation of stator resistance Rs can be difficult, because it changes with the temperature and skin effect, producing low DTC performance. However, many solutions

Fig. 6. Block diagram of the complete system.

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to this problem have been extensively developed [16-20]. A simpler solution is not considering the voltage drop in the stator resistance. Nevertheless, this option gives a poor DTC performance at low speed because the voltage drop increases. The use of pure integrators in (3) is avoided because offset and drift are generated in the estimated flux. A solution is to replace the integrators by a low-pass filters like T/(1+sT), where T is the time constant, which must be large and the stator resistance variation has to be considered. Many others solutions for flux estimation have been development [21-26].

V.

EQUIPMENT

The programming was carried out in an AC800PEC industrial controller from ABB Company, equipped with a Field Programmable Gate Array (FPGA) for very fast firmware application and a Central Processing Unit (CPU) for fast control application. The acquisition of signals was done by isolated current and voltage transducers, and the digital communication was done through optic fibers. The 27-Level inverter implemented was a small size prototype, similar to the one shown in Fig. 1. Insulated Gate Bipolar Transistors (IGBTs) and a 3 kW, squirrel cage induction motor were used in the experimental prototype. The complete system implemented is shown in Fig 9.

(a) (b) Fig. 7. (a) Effect of voltage vector above the Flux vector. (b) Voltage vectors in a 27-Level Inverter.

B. Torque Estimator The torque τe is directly related to the angle δ between rotor and stator flux, and can be obtained using the vectors of flux, and current as in the following equations: p 3 L τe = ⋅ m ⋅ ⋅ψ s ×ψ r 2 Lr σ ⋅ Ls 3 p τ e = ⋅ ⋅ψ s × I s 2 2 3 p Lm τ e = ⋅ ⋅ ⋅ψ r × I s 2 2 Lr

(8) (9)

C. Rotor Speed Estimator A classical close-loop estimator for high performance applications was used, using only monitored stator voltages and currents to obtain the rotor speed. This observer is a Model Reference Adaptive System (MRAS), which estimate some state variables of the motor in a reference model and are then compared with the same state variables estimated by using an adaptive model. The difference between these state variables is then used in a PI controller, which output is the rotor speed. The speed is one of the inputs of the adaptive model [Fig. 8]. The use of pure integrators was avoided because they generate offset and drift in the estimations. Other signals can be used instead of rotor flux to avoid integration and use of stator resistance, as back e.m.f.s and reactive power. However, these were not used on this work. VS IS

REFERENCE MODEL Ψr Flux estimator Equations (3) and (5-6)

ADAPTIVE MODEL Flux estimator Equations (4) and (6)

SPEED TUNING Ψ`r x Ψr

εw

Ψ`r wm

Fig. 9. The 27-Level inverter and AC800PEC controller used.

(7)

PI

Fig. 8. Rotor speed observer using rotor flux.

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VI.

RESULTS AND DISCUSSION

Fig. 10 shows the simulation results of a DTC application, where the rotor speed and the stator flux were controlled directly, and the torque was obtained from the speed error through a Proportional-Integral-Derivative (PID) controller. Fig. 11 shows other simulation results of a DTC application, but in this case, the torque and stator flux were controlled directly without control of the rotor speed. The estimated rotor speed was just as the real speed and the reference speed value is reached in less than 200 ms. The reference torque is obtained with an imperceptible retard and low noise. The motor currents have sinusoidal waveforms, so the current control though PWM was not affected by the ultracapacitor voltages control. The fluxes are circumferences with very low noise (less than 0.02 Wb), and the stator flux magnitude is near to the reference with an error of 0.01 Wb produced in part by the priority of torque control over flux control. The ultracapacitor voltages are in the reference value (29 V) with a maximum error of 0.01V. Fig. 12 illustrates the experimental result of the motor voltages with the INC control disabled and enabled. The voltage THD is 4.7% and 3.6% for current where INC control is enabled. The PWM voltage can be seen in each level and the duty-cycle change to maintain the ultracapacitor voltages in the set-point and to control the motor current. When the INC control is enabled some levels are jumped to avoid regeneration and the voltages are slightly distorted on these levels. However, the voltage waveforms are satisfactory and the ultracapacitor voltages control and DTC are not affected.

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(a)

(a)

(b)

(b)

(c)

(c)

(d)

(d)

(e)

(e)

(f)

(f)

Fig. 10. Simulation results of rotor speed and flux control; (a) rotor speed; (b) torque; (c) motor currents; (d) stator and rotor flux; (e) stator flux; (f) ultracapacitors voltages.

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Fig. 11. Simulation results of torque and flux control; (a) rotor speed; (b) torque; (c) motor currents; (d) stator and rotor flux; (e) stator flux; (f) ultracapacitors voltages.

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160

va

vb

vc

Volts

80 0

-80 -160 0

5

160

10 Time(ms) 15

va

vb

20

25

vc

80 0 -80 -160 0

5

10 Time (ms) 15

20

25

Fig. 12. Experimental voltages with PI control and INC enabled and disabled

VII.

CONCLUSIONS

A Sensorless DTC using a simplified multilevel inverter by means of special topologies and control strategies has been development by simulations and experimental prototype. Experimental results showed a satisfactory control of the ultracapacitor voltages and regeneration avoid in the Aux-2. However, when INC is enabled, the THD has some increment in comparison with a conventional 27-Level inverter, but the proposed inverter was simpler and its voltages remain almost sinusoidal, improving the high-performance of the DTC. The simulations showed a high-accuracy control of the speed and torque, with very low torque ripple. The estimation of speed by MRAS was simple and precise at normal speeds. ACKNOWLEDGMENTS The authors acknowledge the financial support provided in part by Fondecyt under Project Nº 1070751, and in part by Iniciativa Científica Milenio (ICM), under NEIM project Nº P-04-048-F. J.P thanks Conicyt for a Ph.D scholarship. REFERENCES [1] L. Jih-Sheng and P. Fang Zheng, "Multilevel converters-a new breed of power converters," Industry Applications, IEEE Transactions on, vol. 32, pp. 509-517, 1996. [2] J. Dixon and L. Moran, "Multilevel inverter, based on multi-stage connection of three-level converters scaled in power of three," in IECON 02 [Industrial Electronics Society, IEEE 2002 28th Annual Conference of the], 2002, pp. 886-891 vol.2. [3] M. D. Manjrekar, P. K. Steimer, and T. A. Lipo, "Hybrid multilevel power conversion system: a competitive solution for high-power applications," Industry Applications, IEEE Transactions on, vol. 36, pp. 834-841, 2000. [4] S. Kouro, R. Bernal, H. Miranda, J. Rodriguez, and J. Pontt, "Direct Torque Control With Reduced Switching Losses for Asymmetric Multilevel Inverter Fed Induction Motor Drives," in Industry Applications Conference, 2006. 41st IAS Annual Meeting. Conference Record of the 2006 IEEE, 2006, pp. 2441-2446. [5] M. Rotella, G. Penailillo, J. Pereda, and J. Dixon, "PWM Method to Eliminate Power Sources in a Nonredundant 27-Level Inverter for Machine Drive Applications," Industrial Electronics, IEEE Transactions on, vol. 56, pp. 194-201, 2009. [6] M. Depenbrock, "Direct self-control (DSC) of inverter-fed induction machine," Power Electronics, IEEE Transactions on, vol. 3, pp. 420429, 1988. [7] I. Takahashi and T. Noguchi, "A New Quick-Response and HighEfficiency Control Strategy of an Induction Motor," Industry Applications, IEEE Transactions on, vol. IA-22, pp. 820-827, 1986.

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