Dual Frequency Band Integrated Antenna Array

Dual Frequency Band Integrated Antenna Array Pavel Bezousek, Milan Chyba, Vladimir Schejbal, Jan Pidanic University of Pardubice, Studentska 95, 532 10, Pardubice, Czech Republic [email protected], [email protected], [email protected], [email protected] Abstract—Frequently, the antennas of collocated Primary Surveillance Radar (PSR) and Secondary Surveillance Radar (SSR) are situated separately with a common rotational axis. That is not too advantageous in modern individual secondary radar systems, where large vertical apertures (LVA) antennas, quite comparable with the vertical aperture of the PSR antenna are more popular due to their higher gain and lower earth reflection induced problems. A design of new integrated antenna arrays of 3-D PSR and MSSR with no compromise regarding the antenna parameters is presented. Index Terms—primary surveillance radar, secondary surveillance radar, integrated antenna, large vertical apertures antennas

I.

INTRODUCTION

Frequently the antennas of collocated Primary Surveillance Radar (PSR) and Secondary Surveillance Radar (SSR) are situated separately with a common rotational axis [1], [4]. However this configuration is satisfactory only if a small vertical aperture SSR antenna is used. In modern individual secondary radar systems large vertical apertures (LVA) antennas, quite comparable with the vertical aperture of the PSR antenna are more popular due to their higher gain and lower earth reflection induced problems. However location of two high vertical size antennas one over the other on the common mast should lead to a very high antenna system, suffering from mechanical instability and electrical troubles in adverse weather and from transportation difficulty in the case of mobile systems. To solve this problem, some radar systems used a common reflector for the both antennas with separate illuminators. It needs complicated feeds and results in compromise in technical parameters of the both antennas. Another way to solve this problem is to integrate PSR and SSR phased antenna arrays at a common surface. In this paper a design of new integrated antenna arrays of 3-D PSR and MSSR with no compromise regarding the antennas parameters is presented. II.

the band of 2.7 – 2.9 GHz with a central frequency wavelength of λ0 = 107.14 mm. The receiving beams are synthetized in the receiver after dual frequency conversion. Naturally the SSR antenna array will consist of identical columns of radiating elements, fed by a three beam horizontal distribution network. On the contrary the PSR array is formed by horizontal rows of antenna elements, each row being equipped with its own transmitter – receiver block. They are fed by vertical feeders of the transmitted signal, of the control signal and of the 1st and 2nd local oscillators signals. The horizontal and vertical spacing of SSR antenna grid is constrained to be dSSR ≤ 0.9 λ0SSR = 254.7 mm. The vertical spacing of the horizontal PSR antenna rows should be dPSRv ≤ 0.6 λ0PSR = 64.3 mm. For the horizontal rows of the PSR antenna the non-resonant slotted waveguide was chosen due to its very low insertion loss. The optimum horizontal spacing of the PSR antenna elements was determined by an electromagnetic field simulation as dPSRh = 66 mm, being near the waveguide half-wavelength. The horizontal spacing between the nearest SSR antenna elements is then 198 mm. The vertical spacing of the horizontal waveguides was chosen to be dPSRv = 60 mm ≤ 0.6 λ0PSR, then the vertical spacing of the SSR antenna grid is 180 mm. The SSR signal should be vertically polarized in space and therefore the maximum separation of the PSR and SSR antennas is achieved for the horizontally polarized PSR signal. The horizontal narrow wall slotted waveguide with alternating slot slopes generates predominantly a horizontally polarized signal in space. The residual vertical polarization should be suppressed by inserting vertical conducting diaphragms in front of the slotted waveguides forming below-cutoff waveguides for the vertical polarization (Fig. 1). In the gaps between two adjacent diaphragms the vertical feeders of the SSR array columns could be hidden and the SSR antenna radiating elements is situated in front of these vertical feeders.

INTEGRATION OF PSR AND SSR ANTENNAS

In order to get equally spaced antenna elements for the both antenna arrays we need to match their grid distances both in horizontal and vertical directions. The secondary radar antenna [5], [6] creates three beams in the horizontal plane i.e. the sum (Σ), the differential (Δ) and the control (Ω) beams, all with the same vertical patterns in the frequency band 1.03 – 1.09 GHz at the central frequency wavelength λ0 = 283 mm. The 3-D primary radar antenna [7] generates one transmitting beam and eight receiving beams all with identical horizontal patterns in The described research was supported by the project No. FR-TI2/480 of the Czech Ministry of Industry and Trade.

Figure 1. Integration of PSR and SSR antennas.

The inter-diaphragm separations and their length seriously affect not only the vertical polarization component of the PSR signal but also the horizontal one. The same is true about the SSR antenna elements. The SSR antenna element grid is principally greater than the PSR one. The utilization of suspended stripline techniques shown in Fig. 2 brings some advantages [5], [6], [13] - [18]. Using multilayer techniques, one can have layers with patch radiating elements and layers with beam-forming networks. The use of plated-through holes and pins to make RF connections can eliminate cables and connectors. That allows creating very compact design [5], [6]. The vertical feeders are mounted at the front face of the primary radar antenna so there are severe constrains on their dimensions. To save the scare space the branch line couplers should have been dramatically distorted [17] obtaining finally the form, shown in the Fig. 3. It was experimentally proven that such a structure creates periodically displaced perturbances in the PSR antenna array leading to higher PSR antenna sidelobes. Hence it was necessary to add the fictitious identical SSR radiators at each PSR row and column, from which only 1/9 are really active. Usually simple dipoles are used as SSR antenna elements, but such dipoles should be situated about a quarter wavelength (in our case about 70 mm) ahead a reflector plane. Despite the dipoles represent principally a vertically oriented conducting structure they slightly interfere also with the horizontally polarized transmitted PSR wave showing an adverse effect on the PSR antenna sidelobes. That is why the first step in the integrated antenna design was the slotted waveguide design with the polarization diaphragms and SSR antenna radiating elements. III.

SSR ANTENNA

Figure 3. Branch line couplers.

The impedance match of this structure is extremely narrowband, therefore dual frequency match at two separated frequencies has been used. The element was optimized by numerical simulations [6]. In the final construction, the SSR radiating element is formed by a sandwich of the backplane substrate, and the patch substrate isolated by a PUR layer. The parallel impedance matching at the two frequencies appeared to be critically dependent on the element dimensions. To maintain the electrical characteristics requirements in production a specific assembling technology has been developed. The number of elements should be N = 8 and they are 180 mm spaced apart. The assembled SSR array column is shown in Fig. 5. According to system requirements, an elevation pattern shape is a modified cosecant pattern as is shown in Fig. 6. Various synthesis methods have been developed such as [1], [8] - [12]. The well-known Woodward– Lawson method can be easily realized.

After recognition that the standard dipole is not a suitable SSR antenna radiating element due to its large penetration into the field of the PSR antenna other candidates were assumed. The radiating slot as a dual element to the dipole with zero penetration depth is not convenient in our case, because it is oriented horizontally for a vertical polarization and hence it spends a too large space in the horizontal direction. The solution was found in a special patch shown in Fig. 4. The vertically oriented and unsymmetrically driven patch is situated only about 10 mm over the reflector, formed by a frequency sensitive surface, etched in the backplane substrate metallization.

Figure 2. Suspended stripline.

Figure 4. Preliminary design of SSR patch element.

Figure 7. Array factors for the Σ, Δ and SLS beams.

Figure 5. The assembled SSR array column.

The synthesized pattern exactly equals the desired pattern at the sample points due to the orthogonality of the component beams, resulting pattern can be easily modified and the tolerance requirements could be lower. The radiation characteristics of an array are given by the pattern multiplication principle i.e. they are equal to the antenna element factor multiplied by the array factor. The resulting pattern is specified by system requirements considering the properties of patch antenna (see [6] for S11 and element radiation patterns), i.e. the values of array factor should be higher than demanded values. The numerical simulation for required modified cosecant pattern (in fact that is the array factor) shown by solid line and the measured radiation patterns at frequencies of 1.03, 1.06 and 1.09 GHz are depicted in Fig. 6. The design of azimuth pattern is rather more complicated as sum, difference and sidelobe suppression (SLS) beams should be created as is shown in Fig. 7. The Taylor n-bar distribution [8] - [12] for 40 dB maximum sidelobes has been considered for Σ beam, which is formed by 27 elements with spacing of 198 mm.

Figure 6. SSR elevation patterns.

The considered modification has acceptable low sidelobe levels and relatively high efficiency for Σ, Δ and SLS beams. In fact, for the given design sidelobe level, the Taylor distribution provides the narrowest beamwidth, i.e. the maximum directivity. The given modification increases slightly the sidelobe levels. Similarly, the amplitudes for Δ and SLS beam should be modified. That allows creating the Σ, Δ and SLS beams, which are not optimal but could be accepted from the system point of view. An auxiliary “backfill” antenna is added to cover backlobe radiation. The array factors for the Σ, Δ and SLS beams are shown in Fig. 7 considering the directivities of individual beams and the backfill antenna for f =1.06 GHz. The SLS beam level is approximately by 20 dB higher than Σ beam sidelobe levels. That is enough for a satisfactory SLS function. Similarly, the SLS beam gain is by 28.5 dB lower than the Σ beam one for ω = 0o and that fulfills requirements. It is clear that the proposed solution is a reasonable compromise from the system viewpoint. That allows the utilization of a simple feeder network for Σ, Δ and SSL beams avoiding rather complicated feeder network producing an independently optimized Δ illumination. IV.

PSR ANTENNA

The PSR antenna array consists of 32 horizontal rows of 77 radiating elements. Each row is made of a waveguide slot array fed by its own transmitter – receiver block. All the identical slotted waveguides are assembled so that they alternate the direction of tilt to further reduce cross-polarization as is shown in Fig. 1. The PSR antenna structure is reinforced by a system of vertical fins perpendicular to the waveguides, ascending ahead and forming short sections of sub-cutoff waveguides for the vertically polarized wave as is shown in Fig. 1 and 8. The PSR antenna requires four vertical signal distribution networks, i.e. the transmitted signal feeder, the control signal feeder and feeders of the 1st and the 2nd local oscillator (LO) signals. Frequency bands and distributed powers of the signals are in the Tab. 1. The control signal is used for a rolling amplitude and phase control of the individual row receiver channels. The control signal is injected by a special hybrid waveguide-microstrip directional coupler into the opposite end of the waveguide to the input of the transmitted signal.

Figure 8. Top view of the integrated SSR and PSR antenna arays.

The horizontal antenna pattern is equal to a common horizontal pattern of individual slotted waveguides. The synthesis was done using well-known Taylor distribution [1], [8] - [12]. Utilization of numerical simulations and testing allow the assembly of suitable waveguide slot arrays. The S11 of waveguide slot array is depicted in Fig. 9. The details including numerical simulations of azimuth patterns (array factors) with random errors, and S11 measurements of 12 waveguide slot arrays and measurements of azimuth patterns for various frequencies could be found in [7]. An architecture of transmit and receive modules is shown in Fig. 10. A power of 10 W is amplified by high power amplifier (HPA) to 1 kW for any waveguide slot array. A radar receiver operates on the super-heterodyne principle [12]. The receiver filters the signal by a band-pass filter (BPF) to separate desired target signals from unwanted interference. After modest RF amplification by a low noise amplifier (LNA), the signal is filtered by the BPF and shifted to the first intermediate frequency (1st IF) by mixing with the first localoscillator (1st LO) frequency. The second conversion stage is necessary to reach the final (second) IF without encountering serious image- or spurious-frequency problems in the mixing process. The super-heterodyne receiver varies the LO frequency to follow any desired tuning variation of the transmitter without disturbing the filtering at IF.

Figure 9. Measurement of S11 for waveguide slot array.

The 3-D PSR radiates all the transmitted power in only one beam, horizontally shaped by the common slotted waveguides horizontal pattern. The ideal vertical shape of the transmitter beam is the favorite cosec/cosine pattern. Each horizontal row is fed by the individual power amplifier. Adjustment of the output power of each amplifier to a specific magnitude is quite complicated and hence the natural requirements of production and maintenance engineers are to feed all rows with an equal power. It leads to the condition of a uniform signal amplitude vertical distribution at the aperture. For a practical realization of the vertical transmitter signal feeder it is beneficial to keep the phase differences between the same output divider branches as small as possible. The pattern should then be shaped using the signal phase distribution only. The phase synthesis of the desired pattern is described in [18], where the optimized phase distribution and the resulting vertical pattern are presented. The receiving beams are synthetized in the receiver after dual frequency conversion. To reduce dimensions, the 2nd local oscillator feeder is realized using symmetrical stripline feeders shown in Fig. 11. The suspended stripline techniques shown in Fig. 2 are applied for the other feeders.

All the vertical feeders having a uniform amplitude distribution at their 32 outputs use 1P2T symmetrical Wilkinson dividers. The phase distributions at the feeder outputs are also uniform except that of the transmitted signal feeder. The phase differences at its outputs are performed by different signal paths lengths. TABLE I.

FREQUENCY BANDS AND INPUT POWERS OF FEEDER SIGNALS FOR THE VERTICAL PSR ANTENNA

Signal

Frequency band

Transmitted Control 1st LO 2nd LO

2 700 – 2 900 MHz 2 700 - 2 900 MHz 2 050 - 2 250 MHz 610 MHz

Input power 100 mWimp 100 mWimp 10 mWCW 10 mWCW

Figure 10. Architecture of transmit and receive modules.

Figure 11. Stripline cross section of the PSR antenna 2nd LO feeder.

Eight partial receiving beams use a monopulse technique. Basically, two various methods could be used – optimal sidelobe levels and optimal gain. The modification with optimal sidelobe levels uses the Taylor distribution for sum pattern. The sidelobe level is chosen accordingly to system requirements. Theoretically, nearly arbitrary low level can be reached. A half-power beamwidth of 4 deg. has been used for the central frequency. Instead of Taylor distribution, the requirement of feeding all rows with an equal power has been used for synthesis. The sum and differential patterns for optimal sidelobe level are shown in [7]. The modification with improved gain of pattern has been proposed. The half-power beamwidth of 3.4 deg. has been obtained for the central frequency. That is gain increasing of 0.7 dB in comparison to the previous modification. The sum and differential patterns for maximum gain are shown in [7]. Both basic modifications were used for partial receiving beams. Of course, the most suitable variant were considered for any partial receiving pattern. Utilization of an appropriate relative phasing between the elements allows the beam scanning. Moreover, any beam could be slightly modified according to system requirements. V.

CONCLUSION

Frequently the antennas of Primary Surveillance Radar (PSR) and Secondary Surveillance Radar (SSR) are situated separately with a common rotational axis. However this configuration is satisfactory only if a small vertical aperture SSR antenna is used. In modern individual secondary Large vertical apertures (LVA) antennas, quite comparable with the vertical aperture of the PSR antennas, are very popular thanks to their higher gain and lower earth reflection induced problems. However, location of two high vertical size antennas for the PSR and SSR, one over the other on the common mast should lead to a very high antenna system, suffering from mechanical instability, higher wind loading, electrical troubles in adverse weather and from transportation difficulty in the case of mobile systems. To solve this problem, some radar systems used a common reflector for the both antennas with separate illuminators. It needs complicated feeds and results in compromise in technical parameters of the both antennas. The paper presents the other way, i.e. the integration of PSR and SSR phased antenna arrays at a common surface.

Design details of the new integrated array antenna of 3-D PSR and MSSR with no compromise regarding the antennas parameters are presented. The utilization of suspended stripline techniques shown in Fig. 2 brings some advantages. Using multilayer techniques, one can have layers with patch radiating elements and layers with beam-forming networks. The use of plated-through holes and pins to make RF connections can eliminate cables and connectors. That allows creating very compact design. Both numerical simulations including random errors of amplitudes and phases and several measurements are presented. REFERENCES [1] [2] [3]

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