Experimental Demonstration and Field-Trial of an Improved Optical ...

JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 33, NO. 20, OCTOBER 15, 2015

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Experimental Demonstration and Field-Trial of an Improved Optical Fast OFDM Scheme Using Intensity-Modulation and Full-Field Detection Xing Ouyang, Wei Jia, Paul Gunning, Paul D. Townsend, and Jian Zhao

Abstract—We experimentally investigate an improved fast orthogonal frequency division multiplexing (OFDM) scheme using intensity modulation and full-field detection. This new fast OFDM algorithm exhibits better back-to-back and transmission performance than the conventional one, and is shown to support a 40-Gb/s signal over a 480-km re-circulating loop-based single-mode fiber (SMF) with 1-dB penalty. We compare this scheme with the direct-detection (DD) system using the same algorithm, and show that the DD system cannot support 60-km SMF at 40 Gb/s. Finally, we demonstrate the proposed scheme over 124 km of BT Ireland’s field-installed fiber without inline optical amplification. The results show that this scheme can be a promising solution to address the gap between direct detection and coherent detection for applications in short metro networks and long-reach Ethernet and access networks. Index Terms—Detection, modulation, orthogonal frequency division multiplexing.

I. INTRODUCTION RTHOGONAL frequency division multiplexing (OFDM) has become a promising transmission technology due to its high spectral efficiency and robustness to dispersion [1]– [8]. Optical intensity modulation and direct detection (IM/DD) OFDM schemes, such as discrete multi-tone, have been extensively investigated for access networks and Ethernet due to their simplicity and low cost [9]–[14]. Fast OFDM is one of the most attractive OFDM technologies in these applications because the discrete cosine transform (DCT) in this scheme can directly generate a real-valued signal for IM/DD without additional Hermitian symmetric extension. In earlier works, IM/DD fast OFDM was transmitted over both multi- and single-mode fiber (SMF) at 1.55 μm [11], [12] as well as photonic band-gap fiber at 2 μm [13], [14]. However, IM/DD schemes have limited performance due to the dispersion-induced fading effect, and the transmission reaches are commonly less than 50 km at 40 Gb/s. Although coherent detection, which has been widely

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Manuscript received June 17, 2015; revised July 30, 2015; accepted August 12, 2015. Date of publication August 16, 2015; date of current version September 12, 2015. This work was supported in part by the Science Foundation Ireland under Grants 13/TIDA/I2718, 11/SIRG/I2124, and 12/IA/1270, and EU 7th Framework Program under Grant 318415 (FOX-C). X. Ouyang, W. Jia, P. D. Townsend, and J. Zhao are with the Photonic Systems Group, Tyndall National Institute, University College Cork, Cork, Ireland (e-mail: [email protected]; [email protected]; paul.townsend@tyndall. ie; [email protected]). P. Gunning is with the BT Research & Innovation, Suffolk IP5 3RE, U.K. (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/JLT.2015.2469143

used in long-distance transmission systems, can overcome the performance bottleneck [15], this technology exhibits high cost and high power consumption, and so may not be suitable for applications in short metro networks and long-reach Ethernet. Full-field detection (FFD), by extracting both the amplitude and the phase using an asymmetric Mach–Zehnder interferometer (AMZI), can provide a good balance between performance and cost. This scheme was demonstrated with both on-off keying [16] and offset differential quadrature phase shift keying [17] formats. In a more complicated form of FFD using two AMZIs and two pairs of photodiodes [18]–[20], higher-level formats such as M-ary phase shift keying and 16 amplitude quadrature modulation (QAM) were also reported. However, due to the mechanism of full-field reconstruction, the dispersion tolerance of FFD systems is not as good as that in coherent detection, especially for high-level formats. Consequently, applying fast OFDM, which has high resilience against dispersion and easily upgraded format levels, to IM/FFD systems can be a promising solution to enable higher data rate and/or longer transmission reach while maintaining low cost. However, the conventional fast OFDM scheme cannot realize optimal performance when the channel impulse response is not symmetrical. This issue becomes prominent in FFD where imperfect full-field recovery may augment this asymmetry, resulting in degraded performance. Recently, we propose an efficient algorithm which exhibits improved performance in compensating asymmetric channel impulse responses [21]. The feasibility and advantages of the algorithm are validated analytically and numerically in both wireless fading channel and multimode fiber systems. In this study, we experimentally investigate this algorithm in the IM/FFD systems. It is found that the bit error rate (BER) performance can be improved by around one order of magnitude compared to the system using the conventional algorithm even in the back-to-back case. A 40-Gb/s four amplitude shift keying (4ASK) fast OFDM signal, with the spectral efficiency equivalent to 16 QAM conventional OFDM, can therefore transmit over 480-km SMF via IM/FFD without optical dispersion compensation. This is in contrast to the IM/DD system that cannot support 60-km SMF transmission at the same data rate and using the same algorithm. We have systematically investigated the system, including signal launch power, the differential time delay -(DTD) of the AMZI, and the length of training symbols (TSs) for channel estimation. Finally, we demonstrate the scheme over 124 km of BT Ireland’s field-installed SMF without in-line optical amplification. These results show that the

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Fig. 1.

the frequency response of the arbitrary waveform generator (AWG), including the roll-off effect in digital-to-analogue conversion. The peak-to-average power ratio (PAPR) is controlled by clipping to balance the average output power and clipping-induced distortion. The transmitted fast OFDM sequence consists of a start-of-frame (SOF) symbol for synchronization, a training sequence for channel estimation, and payload symbols. At the receiver, the optical signal is processed by an AMZI with π/2 differential phase shift and Δt DTD. The value of Δt is chosen to balance the precision of phase reconstruction and the noise impact. The two outputs of the AMZI are detected by two photodiodes, sampled by analogue-to-digital converters (ADCs). The sampling rate of the ADCs does not have to be the same as that of the AWG but should be higher than the Nyquist rate. Assuming that the time interval between samples is T and the baseband optical field is |E(t)|·exp(jϕ(t)), where E(t) and ϕ(t) are the optical field and its phase, respectively, V+ (pT) and V− (pT) in Fig. 1 at the sampling time pT are:

Principle of the improved IM/FFD fast OFDM scheme.

proposed scheme can be a promising low-cost solution for applications in short metro networks and long-reach Ethernet and access networks.

Fig. 1 shows the principle of the proposed system. Similar to the conventional algorithm, inverse DCT (IDCT) is used for data encoding. Assuming that ai,k and si,n are the kth subcarrier data in the frequency domain and the nth sample in the time domain in the ith OFDM symbol, si,n can be derived as: N −1

= ε0 · ai,0 +

εk · ai,k /2 · exp(jπk/(2N ))

ε2N −k · ai,2N −k /2

k =N +1

· exp(jπk/(2N )) · exp(j2πkn/(2N ))

(3)

In the case that ϕ(pT)–ϕ(pT–Δt) is small, sin(ϕ(pT)–ϕ(pT– Δt)) ࣈ ϕ(pT)–ϕ(pT–Δt). By recovering the phase difference in the time interval Δt, the full optical field can be reconstructed by

· exp(j2πkn/(2N )) 2N −1

where ϕA M Z I (= π/2) is the differential phase of the AMZI. If the Δt value is not large, |E(pT–Δt)| ࣈ |E(pT)|. The validity of this approximation also depends on parameters such as signal formats, dc bias, and accumulated dispersion, but commonly, Δt should be less than the inverse of the signal baud rate [16], [17]. With this approximation, we can derive:

≈ sin(ϕ(pT ) − ϕ(pT − Δt)).

k =1

+

(2)

Vf (pT ) ∝ (V+ (pT ) − V− (pT ))/(V+ (pT ) + V− (pT ))

εk · ai,k · cos(πk(2n + 1)/(2N ))

k =0 N −1

± 2 |E(pT )| · |E(pT − Δt)| · cos(ϕ(pT ) − ϕ(pT − Δt) + ϕA M Z I )

II. PRINCIPLE

si,n =

V± (pT ) ∝ |E(pT )|2 + |E(pT − Δt)|2

(1)

where N is √ the number of subcarriers and j is the imaginary unit. εk is 0.5 for k = 0, and 1 for k = 1, . . . , N − 1. The derivation on the right-hand side of Eq. (1) sets up the relation between the IDCT and the inverse discrete Fourier transform: si,n can be viewed as a multicarrier signal with 2N subcarriers and the data on subcarriers are [ε0 · ai,0 , ε1 · ai,1 /2 · exp(jπ/(2N )), . . . , εN −1 · ai,N −1 /2 · exp(jπ(N − 1)/(2N )), 0, εN −1 · ai,N −1 /2 · exp(jπ(N + 1) /(2N )), . . . ε1 · ai,1 /2 · exp(jπ(N − 1)/(2N ))]. In the proposed scheme, a zero-padded guard interval (GI), instead of a symmetric extended GI [15], is added for dispersion compensation. The length of the GI should be larger than that of the channel impulse response. Although it is not implicitly shown in Fig. 1, the signal is pre-equalized to compensate for

Vf u ll (pT ) ≈ (V+ (pT ) + V− (pT ))1/2 p · exp j × asin(Vf (qT )) · T /Δt . (4) q =−∞

Ideally, Vf u ll (pT ) is obtained given a sufficiently small Δt. In practice, low-frequency components of the noise accumulate in the summation module, so have to be suppressed by the use of a high-pass electrical filter [16]. Once the full optical field consisting of the amplitude and phase is recovered, the signal can be decoded similarly to in coherent detection. When the sampling rate of the recovered optical field is not the same as that of the AWG, re-sampling is firstly performed to synchronize fast OFDM symbols. In the proposed algorithm, the received fast OFDM symbols, including 2N ) the GI, are firstly zero-padded to a length of 2N (defined as ri,n and a 2N-point DFT is then used to transform the signal to the frequency domain. The mth output of the DFT in the ith symbol

OUYANG et al.: EXPERIMENTAL DEMONSTRATION AND FIELD-TRIAL OF AN IMPROVED OPTICAL FAST OFDM SCHEME

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is derived as: 2N 2N bi,m = F {ri,n } = F {s2N i,n ⊗ hn } = Hm · Si,m

(5)

where F{·} is the DFT and ࣹ represents the convolution. s2N i,n and h2N n are the time-domain signal in the ith OFDM symbol and the channel impulse response, both of which are zero-padded to and s2N the length of 2N. Hm and Si,m are the DFT of h2N n i,n , respectively. From Eq. (1), Si,m can be further derived as:

N −1

Si , m =

si , n · exp(−j2πmn/(2N ))

n=0

= exp(jπm/(2N )) ·

⎧ (c0 a i , 0 + j · ck ai , k ) ⎪ ⎪ ⎪ k = 1 , −1 , . . . ⎪ ⎪ ⎪ ck ai , k ) ⎨ (cm ai , m + j ·

m=0 m = 1...N − 1

k −m = 1 , −1 , . . .

⎪ 0 ⎪ ⎪ ⎪ ⎪ ⎪ ⎩ (cm ai , 2 N −m + j ·

k −m = 1 , −1 , . . .

m=N ck a i , 2 N −k )

m = N + 1...2N − 1.

(6) The fixed term for the mth output in Eq. (6), exp(jπm/(2N), can be readily compensated in practice. cm are real constants, including the coefficients εm . Observation of the right-hand side of Eq. (6) shows that the crosstalk (that multiplied by j) is orthogonal to the signal with π/2 phase difference. In addition, for any m at the output of the 2N-point DFT, the crosstalk is only from subcarriers whose index k satisfies that k-m is an odd value. Therefore, at the channel estimation stage, if the training data are inserted every two subcarriers in the TSs, the crosstalk terms can be eliminated. For example, we can use M TSs: in the first M/2 TSs, even subcarriers are set to be zero and the channel response, Hm · cm , of odd subcarriers is estimated. Conversely, in the second M/2 TSs, odd subcarriers are set to be zero so that the response of even subcarriers can be obtained. We employ the estimated Hm · cm and combine Eqs. (5) and (6) to get: ∗ real{bi,m · exp(−jπm/(2N )) · Hm · cm } ⎧ 2 m=0 ⎪ ⎪ |H0 c0 | ai,0 ⎪ ⎪ ⎪ 2 ⎨ |Hm cm | ai,m m = 1...N − 1 = ⎪ 0 m=N ⎪ ⎪ ⎪ ⎪ ⎩ 2 |Hm cm | ai,2N −m m = N + 1...2N − 1

Fig. 2.

Experimental setup for the improved IM/FFD fast OFDM scheme.

|H2N −k c2N −k |2 in Eq. (8) introduces diversity and so results in better performance when the channel response is asymmetric. This benefit has much significance in practical implementations because the impulse responses of electrical/optical devices are not ideally symmetric. In addition, in FFD, the signal pulse or spectral profile is degraded by the impairments or noise in the full-field recovery process, which augments the asymmetric effect. The proposed scheme can thus exhibit a higher performance improvement. III. EXPERIMENTAL SETUP

(7)

where the superscript ∗ represents the conjugate and Real{·} is the real part. An interesting implication of Eq. (7) is that more freedom, 2N outputs of the DFT, bi,m , 0 ≤ m ≤ 2N − 1, can be used to recover the N transmitted data, ai,k , 0 ≤ k ≤ N − 1: (|Hk ck |2 + |H2N −k c2N −k |2 )ai,k = real{bi,k · exp(−jπk/(2N )) · Hk∗ ck } ∗ + real{bi,2N −k · exp(jπk/(2N )) · H2N −k c2N −k }. (8)

Here the frequency responses, Hk and H2N −k , do not have to be the same, removing the symmetric constraint in the conventional algorithm. In fact, the averaging of |Hk ck |2 and

Experiments were carried out to illustrate the performance of the proposed scheme, and its advantage over the IM/FFD system using the conventional algorithm and the IM/DD system using the new algorithm. Fig. 2 shows the experimental setup for both re-circulating loop based transmission and field trial over 124 km of BT Ireland’s SMF between Cork City and Clonakilty, Ireland. Bi–polar 4–ASK data streams were encoded with Gray coding in MATLAB. The fast OFDM signal consisted of 256 subcarriers, of which 208 subcarriers were used for data modulation. Three subcarriers in the zero–frequency region were not modulated, allowing for ac–coupled amplifiers. The last 45 subcarriers were zero–padded to avoid aliasing. Both the conventional and proposed fast OFDM algorithms were implemented, with symmetric–extended and zero–padded GI added to each symbol, respectively. The GI length was 32, which was sufficient for the dispersion of up to 500–km SMF with 17 ps/nm/km and 20–GHz electrical bandwidth. Pre–equalization was integrated in the algorithm to compensate for the transmitter–side response, and the PAPR was limited to 11 dB by clipping.

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The generated fast OFDM signal was then uploaded into an AWG, which interleaved two 12–GS/s digital–to–analogue converters (DACs) with optimized timing alignment to achieve a sampling rate of 24 GS/s. The bandwidth of the AWG excluding the sinc–function roll–off was 7.5 GHz and these effects were considered in the pre–equalization algorithm. The signal rate was around 40 Gb/s. By considering the GI and 7% forward error correction overhead, the net data rate was around 33 Gb/s. In contrast to the coherent detection, the FFD scheme does not require a high–specification laser, and lasers with several MHz linewidth are sufficient to avoid phase–noise induced penalty [17]. In the experiment, a distributed feedback DFB laser with ∼1–MHz linewidth was used to generate the optical carrier and a Mach—Zehnder modulator (MZM) with Vπ of ∼3 V was used for intensity modulation. The MZM was biased at the quadrature point and the peak-to-peak voltage of the electrical fast OFDM signal was ∼1.3 V, which ensured the system operated in the linear region. The modulated optical signal was then amplified and transmitted over either a recirculating loop comprising 60 km of SMF with ∼14-dB fiber loss or the 124-km field-installed fiber link (∼27-dB fiber loss). In the first scenario, the signal polarization was randomized after each recirculation using a polarization scrambler (POL). The noise figure of the loop amplifier was ∼5.5 dB and a 0.8-nm optical band-pass filter (OBPF) was used in the loop to suppress the amplified spontaneous emission (ASE) noise. The launch power was varied for investigation. At the receiver, the optical signal was detected with an optically pre-amplified receiver and a variable optical attenuator (VOA) was used to vary the input power to an erbium-doped fiber amplifier (EDFA). In the second scenario, the fiber loss was high and so the VOA was not used to avoid additional degradation of optical signal-to-noise ratio (OSNR). In both cases, the pre-amplifier was followed by an OBPF with a 3-dB bandwidth of 0.64 nm, a second EDFA, and another optical filter with a 3-dB bandwidth of 0.8 nm. The second EDFA and OBPF at the receiver ensured a fixed optical power into the photodiodes and so mitigated the influence of thermal noise when the attenuation of the VOA was varied to obtain different OSNR values. This stage could be removed in a practical implementation. The optical signal then passed through a Kylia AMZI with a tunable DTD and π/2 differential phase shift. The two outputs of the AMZI were detected by two photodiodes. Both detected signals were sampled by a 50-GS/s real-time oscilloscope. In off-line processing (implemented in MATLAB), an algorithm based on the principle of [5] was used to automatically identify the SOF symbol and determine the DCT/DFT window. Due to the ac-coupled receiver amplifiers, manuallyoptimized bias was added to the two signals before they were added and subtracted for full-field reconstruction. The bandwidth of the high-pass filter (HPF) for suppressing the phase noise was varied for investigation. After full-field recovery, both the conventional and new fast OFDM algorithms were studied. In the conventional algorithm, a DCT was used for subcarrier demultiplexing. However, the performance of this algorithm was not optimal when the system impulse response was asymmetric. In the new fast OFDM algorithm case, zeros were padded


Fig. 3. BER versus the fiber length when the received power is –8 dBm using the new fast OFDM algorithm. The signal launch power is 2 dBm and the DTD of the AMZI is 38 ps.

to each OFDM symbol and a 2N-point DFT was then used to transform the signal to the frequency domain. Channel estimation was implemented using the principle as described above. Two sets of data were extracted independently with each consisting of ∼900 measured fast OFDM symbols, giving a total number of measured 4-ASK symbols of 208 × 900 × 2 = 374 400. The BER was obtained by direct error counting. For comparison, we also implemented the IM/DD fast OFDM system with the proposed algorithm. The experimental parameters were the same as those in the FFD system. At the receiver, instead of an AMZI and two photodiodes, a single photodiode was used for direct detection and full-field recovery was bypassed in the decoding algorithm. IV. EXPERIMENTAL RESULTS We firstly measured the electrical back-to-back performance by directly connecting the AWG and the oscilloscope, and the obtained minimum BER was ∼1 × 10−5 . This error floor reflects the limitation of electrical equipment used in the experiment, such as the bandwidth and quantization resolution of DAC/ADC. We then launched the signal to the optical re-circulating loop setup. Fig. 3 shows the obtained minimum BER versus the fiber length when the received power (after the receiver VOA) is around –8 dBm. This received power ensures minimized OSNR degradation at the receiver side and the OSNR is mainly limited by the inline ASE noise. It can be seen that the minimum BER in the optical back-to-back case is 10−4 , around one order magnitude of reduction compared to the electrical back-to-back case. This degradation represents the limitation of optical devices in the experiment such as the bandwidth and nonlinearity of devices, and thermal noise of photodiodes. Note that at 0 km, the signal phase is close to zero while the phase noise during the full-field recovery still exists, and so the performance is slightly poorer than that at 60 km, at which point the value of signal phase increases due to dispersion. In fact, additional results show that when the phase recovery path in Fig. 1 (i.e., the Vf path) is neglected for 0 km, the performance is similar to that at 60 km. We can also see in the figure that the system can support 480 km with a minimum BER of ∼10−3 . By using full-field recovery, dispersion can be compensated similarly to in coherent

OUYANG et al.: EXPERIMENTAL DEMONSTRATION AND FIELD-TRIAL OF AN IMPROVED OPTICAL FAST OFDM SCHEME

Fig. 4. BER versus OSNR using the new fast OFDM algorithm. The signal launch power is 2 dBm and the DTD of the AMZI is 38 ps.

Fig. 5. BER versus OSNR using the conventional (C) and the new algorithms. The signal launch power is 2 dBm and the DTD of the AMZI is 38 ps.

detection and the dispersion-induced fading effect does not occur. The reduction of the minimum BER with distance is due to the residual penalty arising from fiber nonlinearity and impairments in the full-field recovery process, as well as the limitation of the maximal OSNR. In order to see the influence of these effects clearly, Fig. 4 shows the BER versus OSNR at different fiber lengths. Different OSNR values are obtained by varying the VOA before the first EDFA at the receiver. It can be seen that the OSNR penalty after 480 km with respect to the optical back-to-back case is around 1 dB. In addition, the ASE noise accumulates in the fiber link and the maximal OSNR at 480 km is