High-power-factor electronic ballast based on a single power ...

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 47, NO. 4, AUGUST 2000

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High-Power-Factor Electronic Ballast Based on a Single Power Processing Stage Márcio Almeida Có, Student Member, IEEE, Domingos S. L. Simonetti, Member, IEEE, and José Luiz F. Vieira, Member, IEEE

Abstract—A new high-power-factor (HPF) electronic ballast is introduced in this paper. The proposed topology is based on a single power processing stage, and provides a high-frequency voltage to the fluorescent lamps in addition to presenting an HPF to the utility line. The power processing stage is formed by a half-bridge circuit operating above the resonant frequency, thus providing zero-voltage switching. The self-oscillating technique is employed, which increases the converter reliability with great simplicity. HPF is achieved by using a nonconventional boost stage operating in discontinuous conduction mode, which results in a lower dc-bus voltage than that produced by the conventional boost. Theoretical analysis and experimental results have been obtained for two 40-W fluorescent lamps operating at 40-kHz switching frequency and 220-V line voltage. Index Terms—Lighting, power-factor correction, soft switching.

I. INTRODUCTION

A

RTIFICIAL lighting is responsible for the consumption of a significant portion of the overall generated electricity. Much effort has been devoted to the search for more efficient lighting sources to replace incandescent lamps. The gas discharge lamp technology has been of relevance for this purpose, particularly, the most popular of all is the fluorescent lamp. It is usually preferred because it inherently has longer lifetime and higher efficiency, when compared to the incandescent lamp. All gas discharge lamps have negative impedance characteristics that require some form of current-limiting control to prevent their destruction by excessive current. This problem has been solved by means of a magnetic ballast, composed by a series inductor. In spite of their low cost, magnetic ballasts present the following problems, mainly associated with their operation at line frequency (50 or 60 Hz): flickering, high size and weight, and low power factor. These problems can be eliminated with the use of electronic ballasts, which have become cost competitive with conventional ballasts, particularly when overall costs are considered (the initial cost plus increase in lifetime and energy savings) [1]–[5]. Electronic ballasts based on self-oscillating technique possess inherent current-limiting control to the lamps, usually operate at high resonant frequency, and yield high striking voltage. Manuscript received August 17, 1998; revised April 9, 2000. Abstract published on the Internet April 21, 2000. An earlier version of this paper was presented at the 1995 IEEE Power Electronics Specialist Conference, Atlanta, GA, June 12–15, 1995. The authors are with the Departamento de Engenharia Elétrica, Universidade Federal do Espírito Santo, Vitória, ES 29060-970, Brazil (e-mail: [email protected]). Publisher Item Identifier S 0278-0046(00)06823-4.

They can also provide shutdown protection in the event of a lamp failure or lamp removal. parallel resonant converter A self-oscillating half-bridge has been an attractive choice for this application, due to its competitive cost and high reliability [1], [6]. The resonant converter is fed by a dc voltage source generated from the ac mains by a diode bridge rectifier, as shown in Fig. 1. It operates above the resonant frequency to provide zero-voltage switching (ZVS). However, from the ac line viewpoint, it absorbs a current with a low power factor and high harmonic distortion. High power factor (HPF) and low total harmonic distortion (THD) are features specified by international regulations and also required by utility companies. The advantages of HFP include a reduction in the line current harmonics and, consequently, in the rms line current. In this way, the utility line is more efficiently utilized [7]. The use of a continuous conduction mode (CCM) boost power factor corrector (PFC) is possible but not useful, because it requires two processing stages [2], [8] which reduces the reliability and increases the overall cost. Therefore, a single power processing stage has been the preferred option [7], [9]–[16]. In this way, this paper proposes an HPF electronic ballast operating in discontinuous conduction mode (DCM) that results in a high-performance, reliable solution. An additional feature of the proposed topology is that it presents input–output galvanic isolation.

II. CONVERTER TOPOLOGY The power stage diagram of the proposed HPF electronic ballast is shown in Fig. 2, and is described as follows: ; • input voltage – ; • input diode bridge rectifier – ; • MOSFET half-bridge with the • three-winding high-frequency transformer ; turns ratio: • boost inductor ; ; • high-frequency diode , and ; • parallel resonant circuit formed by • dc-link capacitor ; • blocking capacitor ; • self-oscillating gate-drive transformer ; and , which eliminates • high-frequency input filter switching frequency harmonic current. A nonconventional boost stage operating in DCM is formed and the transformer secondary , by the boost inductor

0278–0046/00$10.00 © 2000 IEEE

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Fig. 1. Self-oscillating half-bridge series resonant converter.

Fig. 2.

Power stage diagram of the proposed HPF electronic ballast.

as shown in Fig. 3. This secondary produces a high-frequency . By using a suitable turn ratio, discontinuous square voltage during the full line cycle is achieved. current conduction in Due to the high-frequency filter, a high-quality current flows prevents the negative current in the utility line. The diode across the boost inductor, besides blocking, achieving a peak and . Just one fast diode is value of oscillations between used in this topology, in contrast to that presented in [17], which employs a fast diode bridge. is equal to , As can be noted in Fig. 3, if the voltage current increases linearly the

(1) is the duty ratio (equal to 0.5), and where is the switching frequency.

The peak current follows a sinusoidal envelope defined by the input voltage. Since the transformer primary voltage is ( blocks the dc voltage level), the turns ratio should be 1:2. is equal to , the current decreases linearly When with the ratio (2) can be maintained slightly greater The dc-bus voltage , which ensures DCM than the maximum ac peak voltage during the entire cycle of utility line. A conventional boost greater than twice to achieve similar stage requires results [18]. This is an advantage of this topology mainly in can be maintained 220-V input voltage applications, where less than 400-V dc. At this voltage, the MOSFET’s rms current levels are lower, yielding higher efficiency than that of other solutions.

CÓ et al.: HPF ELECTRONIC BALLAST

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Fig. 3. AC/DC nonconventional boost converter.

The tertiary with a turn ratio of provides the necessary voltage level to drive the two series-connected fluorescent lamps. In addition, it ensures proper circuit isolation for the lamps. A further simplification of this nonconventional boost converter is shown in Fig. 4(a), and a simulation result for the inductor current is shown in Fig. 4(b). This result was obtained V, V, and . for: The details of the current conduction mode are emphasized by Hz. As this conusing a small switching frequency verter operates in DCM, the input current naturally follows the sinusoidal waveform of the input voltage, providing HPF to the utility line. III. PRINCIPLE OF OPERATION To establish the principle of operation the following assumptions are made. square wave has amplitude equal to , which is • The assured by the 1:2 turn ratio transformer . • The output voltage profile of the input rectifier is simply the rectified ac voltage, since current conduction occurs for the entire 60-Hz cycle. ) during • The input voltage remains constant (at value a switching cycle since the switching frequency is much greater than the supply frequency. • The input stage of this ballast always operates as a boost is always larger than the converter, since the voltage . This is ensured by an maximum AC peak voltage value. appropriate is large enough to be considered a • The capacitance voltage source. • At high frequency, the fluorescent lamps can be consid[3]. ered as a resistive load of value Based on the above assumptions, this electronic ballast can be viewed as two simplified independent converters. The first parallel resonant conone is obtained when the half-bridge

Fig. 4. (a) Simplified circuit of nonconventional PFC boost converter. (b) Input current simulation result.

verter is considered as a resistive load of value . The resulting converter is that shown in Fig. 3, which represents an ac/dc nonconventional boost converter, that operates as a DCM PFC stage. It is a simple but effective method of achieving HPF.

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Fig. 5.


Half-bridge LC parallel resonant converter.

The second one is obtained when the nonconventional boost converter is replaced by the voltage source . The resulting converter is shown in Fig. 5, which is an isolated half-bridge parallel resonant converter. It has been an attractive converter used in electronic lighting systems. Consequently, the proposed electronic ballast operation can be understood as the cascade operation of the two independent converters referred above.

IV. STAGES OF OPERATION AND WAVEFORMS In order to describe the operating stages of this electronic ballast, the following assumptions are made. • The MOSFETs’ model is formed by an ideal switch , an , and a constant capacitance . ideal diode • The transformer magnetizing current is negligible. • The commutation time is negligible compared to the switching period. is large enough to prevent the dc level • The capacitance of current to circulate through the transformer primary. At steady state, a switching cycle operation of this electronic ballast is represented by seven stages. The equivalent circuits of these stages are shown in Fig. 6. Fig. 7 shows the most relevant waveforms of the proposed circuit, along with the gate signals, obtained from the simulation study. The output stage behaves like a half-bridge parallel resonant converter, operating above the resonant frequency due to the self-oscillating gate-drive technique, ensuring ZVS. At startup, circuit operates with a high quality factor (when the the lamps are off, their equivalent resistance is very high), which ensures high ignition voltage to strike the lamps. At steady state, circuit is designed to establish the lamps’ rated curthe rent [2], [19]. A square voltage is applied to the resonant cell, yielding a quasi-sinusoidal current. Fig. 7 also shows both the

transformer secondary voltage and the resonant inductor . current The transformer primary current is obtained by adding the reflected boost stage current, without its dc level, and the reflected resonant current, as described by

(3)

is the average boost inductor current in a where switching period. The converter operation can be described as follows. was zero Just before instant , the inductor current and the transformer primary current was freewheeling through switch . : At instant , the 1st. Stage—First Commutation is turned off. The current can be considered conswitch and . The and stant and flows through capacitances voltages change linearly until becomes zero. As this commutation time is much lower than the switching period, it can be neglected. Linear Increasing Through 2nd. Stage— : At instant , the diode starts to conduct, and just is turned on at zero voltage. Hence, after this, the switch and follow during this stage, the currents

(4) (5)

. This stage ends at instant when : At 3rd. Stage— Linear Increasing Through instant , starts to conduct the current . During this stage,


813

(a)

(b)

(c)

(d)

(e)

(f)

(g) Fig. 6.

Equivalent circuits of operation stages. (a) First stage. (b) Second stage. (c) Third stage. (d) Fourth stage. (e) Fifth stage. (f) Sixth stage. (g) Seventh stage.

the current keeps on increasing linearly according to (4). At and become the end of this stage, the currents

and

where

is the maximum value of the boost inductor current

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Fig. 7. Simulated main waveforms along with the gate-drive signal of the converter.

in a switching period, and is the maximum value of the resonant current. : At instant , 4th. Stage—Second Commutation flows through capacthe switch is turned off. The current and , and their voltages change in a linear fashion itances

until becomes zero. Similarly to the first stage, this commutation time can also be neglected. Linear Decreasing Through 5th. Stage— : At instant , the diode starts to conduct, and is turned on at zero voltage. right after this, the switch


During this stage, the currents

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and

follow (6)

(7) . This stage ends at instant when Linear Decreasing Through 6th. Stage— : At instant , starts to conduct the current . During this stage, the current still decreases linearly and according to (6). At the end of this stage, the currents become Fig. 8. of .

and

Electronic ballast PF (solid line) and THD (dotted line) as a function

7th. Stage—Freewheeling : During this stage, the remains zero and the current freewheels through current according to (8)

V. RELEVANT ANALYSIS The significant characteristics of the proposed electronic ballast are defined by the input current, the power factor, and the THD, whereas the main design parameters to be determined are the boost inductance, the resonant parameters, and the switches currents. A. Input Current and the secAs the ratio between the transformer primary is 2, current conduction occurs in the ac power ondary supply during the entire 60-Hz cycle. Consequently, the input follows an envelope solely defined by the current peak input voltage waveform, according to (1). current, , and , The increasing and decreasing time of respectively, are given by

Fig. 9. Normalized boost inductance as a function of .

where (13)

B. Power Factor and THD (9)

The power factor is defined by

(10) Due to the high-frequency input filter, the ac line current is given by the instantaneous mean value of the boost inductor current, for each half cycle of the utility line, according to the following [20]: (11)

(14) where (15) Assuming that the input voltage has no harmonic components, the power factor can be given by

Substituting (1), (9), and (10) into (11), using a duty ratio of 0.5 into (1) results in (16) (12)

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Fig. 10. Complete diagram of the proposed EB.

Considering unit displacement factor, the THD can be defined by THD

(17)

The proposed electronic ballast power factor and THD as a function of given by (16) and (17), respectively, are shown in Fig. 8. The power factor and the THD of this electronic ballast are better than those of the conventional boost operating in DCM. These improvements are obtained because for the same output voltage , the boost inductor current in the proposed topology decreases with a rate twice of that for the conventional one, as indicated by (2). C. Boost Inductance The boost inductance is obtained from (15), considering that , where is the estithe output power is given by: mated electronic ballast efficiency. Hence, the normalized boost inductance is defined by

quality factor at the natural frequency for

resonant frequency

(22) (23)

parallel circuit is fed by the high-frequency The . square-wave voltage source of magnitude Its fundamental component, obtained from the Fourier analysis, is given by (24) parDue to the self-oscillating gate-drive technique, the allel resonant circuit can operate at the undamped natural fre, assuming that the gate-drive transformer does quency, not operate in the saturation region, and the MOSFET turn-on time is neglected [2], [19]. At this frequency, the amplitudes of and currents the fundamental component of voltage across and are, respectively, [2] through (25)

(18)

(26)

where

(27) (19)

The normalized boost inductance, as a function of , can be obtained from Fig. 9. D. Resonant Parameters parallel resonant circuit, shown in Fig. 5, is a secondThe order low-pass filter, which can be described by the following [2], [19]: undamped natural frequency

When the fluorescent lamps are off, they can be considered as an open circuit. Therefore, the quality factor at startup is very high. As shown by (25), the voltage across the lamps will be parallel high enough for striking them. At steady state, the procircuit operates above the resonant frequency viding ZVS. The power delivered to the lamp is obtained from (26) (28)

(20) VI. DESIGN PROCEDURE AND EXAMPLE

characteristic impedance

(21)

A design procedure for this electronic ballast, along with a practical example, is presented next.


817

(a)

(b)

(c)

(d)

Fig. 11. (a) Input voltage v (100 V/div) and input current i (0.5 A/div), time scale: 2.5 ms/div. (b) Rectified input voltage v (100 V/div) and boost inductor current i (0.5 A/div), time scale: 2.0 ms/div. (c) M commutation, v (100 V/div) and i (1 A/div), time scale: 2.5 s/div. (d) M commutation, v (100 V/div) and i (1 A/div), time scale: 2.5 s/div.

C.

A. Input Data The input data are as follows: rms AC input voltage output power fluorescent lamp rated current switching frequency

V/60 Hz; W; A; kHz.

and

Parameters

From (13) and (19), and assuming and .

, the result is:

D. Power Factor, THD, and Boost Inductance From Figs. 8 and 9, the following values are obtained: % and mH. E. Resonant Parameters

B. Selection of DC-Link Voltage As the electronic ballast input stage always operates as a boost must be larger than the maximum ac converter, the voltage V for a 220-V ac line. In this case, peak voltage V has been selected.

The lamps’ equivalent resistance is . From (22) and assuming is ob. For , tained. Equation (28) results in nF, and using (20), (21), and (23) are obtained mH, and kHz. The switching frequency

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(e)

(f)

(g) Fig. 11. (Continued.) (e) Transformer primary voltage and current: v (50 V/div) i (1.0 A/div), time scale: 5 s/div. (f) High-frequency fluorescent lamp voltage v (50 V/div) and current i (0.25 A/div), time scale 2.5 s/div. (g) Fluorescent lamps current i (0.25 A/div), time scale 10 ms/div.

can be adjusted slightly around by modifying the gate-drive rise time (time to transformer turns ratio, which affects the achieve the MOSFET threshold voltage). F. Transformer The transformer secondary must have a 1:2 turn ratio to ensure DCM operation, where the input current naturally follows the sinusoidal input voltage waveform. As can be seen in Fig. 4(b), there is a dc current level in this secondary. The core saturation can be avoided by limiting the flux density T swing. For the core selected in this case, has been adopted. The tertiary turn ratio is 1:0.9, which adapts the voltage to drive two series-connected 40-W fluorescent

lamps. Due to , the voltage across the transformer primary is , which is used to design the a square wave of amplitude transformer.

VII. EXPERIMENTAL RESULTS An electronic ballast prototype was built to meet the input data specifications. The complete diagram is shown in Fig. 10, whose parameters and components are the following: mH, 130 turns on core EE 42/15, IP6-Thornton; • mH, 66/2 turns on core EE 30/14, IP6-Thornton; • mH, 37 turns on core EE 20/10, IP6-Thornton; • nF/ V, (polypropylene); •


• •

F/ F/

V; V and

819

F/

V, (polypropy-

lene); • transformer : 50/100/45 turns on core EE42/20, IP6Thornton; : 5/14/14 turns on core EE 20/10, • transformer IP6-Thornton; – : 1N 4004; • input diode rectifier bridge, : IRFP 360 (International Rectifier); • : SK3GF04 (SEMIKRON), -1N • 759; k , F/ V, • startup circuit: -DB3, -1N 4004; , pF/250 V; • snubber circuit: -BC547, • overvoltage and overcurrent circuit: k , k , k , F/ V, -1N914, -1N758. , and the diac . The startup circuit is formed by voltage reaches the breakover of the diac ,a When the . The drain voltage, turn-on pulse is applied to switch previously held high by , is switched to ground, thus starting prevents generation of further startup the oscillation. Diode pulses. and avoids high The snubber circuit formed by at startup. The overvoltage and overcurrent circuit provide shutdown protection in case of lamp failure or lamp removal. V, Experimental waveforms were obtained for: A, and are shown in Fig. 11. The input ac current and voltage, which demonstrate the HPF of this electronic ballast, are shown in Fig. 11(a). The rectified input voltage and the boost inductor current are shown in Fig. 11(b). The MOSFET and commutations can be seen in Fig. 11(c) and (d) respectively, which shows the turn-on at ZVS. The transformer primary voltage and current are shown in Fig. 11(e). The highfrequency fluorescent lamp voltage and current are shown in Fig. 11(f), and the fluorescent lamp current, showing its unmodulated envelope, can be seen in Fig. 11(g), from which one can obtain 1.53 crest factor. The experimentally obtained character% , and %. istics were: VIII. CONCLUSION This paper has introduced an HPF electronic ballast based on a single power processing stage, that operates at a lower dc-bus voltage, when compared to conventional boost topologies. A nonconventional boost stage is based on a high-frequency square voltage, obtained from a transformer secondary, which is connected in series with the rectified input voltage. Therefore, the dc-bus voltage can be maintained slightly greater than the maximum ac peak voltage. This is especially of interest for 220-V mains, where the dc voltage can be kept below can 400 V. Therefore, MOSFET’s with lower voltage and be used compared to the conventional boost stage. Other single power processing stage topologies generally need a high dc-bus voltage to achieve HPF with low THD. This topology uses a reduced number of semiconductor components compared with other single or two power processing

stage topologies. This is an important feature for the electronic ballast reliability. The requirement of a transformer can be considered a drawback. However, this transformer ensures proper galvanic isolation for the lamps. The proposed electronic ballast operation can be understood as the cascade operation of two independent converters. The first one is a nonconventional boost converter operating in DCM. It is a simple and effective method of achieving HPF. The second one is an isolated self-oscillating half-bridge parallel resonant converter, operating above the resonant frequency to provide ZVS, which is an attractive feature, due to its competitive cost and high reliability. If a constant switching frequency is required in spite of resonant component variations, a commercial gate drive such as IR2155 or IR2151 can be employed in the control circuit. Since there is a dc current level in the transformer secondary, the design must be done to avoid core saturation, which can be performed by limiting the flux density. Theoretical analysis, a design procedure, and an example have been presented. Experimental results were obtained for two 40-W series-connected fluorescent lamps operating at 40-kHz switching frequency and 220-V line voltage, which demonstrate the HPF and efficiency characteristics of this electronic ballast.

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Márcio Almeida Có (S’99) was born in Vitória, Brazil, in 1968. He recieved the B.E. degree from the Federal University of Espírito Santo, Vitória, Brazil, and the M.S. degree from the Federal University of Santa Catarina, Florianópolis, Brazil, in 1990 and 1993, respectively, both in electrical engineering. He is currently working toward the Ph.D. degree in electrical engineering at the Federal University of Espírito Santo. Since 1997, he has been a Professor in the Federal Technologic Education Center of Espírito Santo, Vitória, Brazil. His research interests include switch-mode power supplies, highpower-factor converters, and lighting systems.

Domingos S. L. Simonetti (S’92–M’95) was born in Vitória, Brazil, in 1961. He received the degree in electrical engineering from the Federal University of Espírito Santo, Vitória, Brazil, the M.Sc. degree from the Federal University of Santa Catarina, Florianópolis, Brazil, and the Ph.D. degree from the Polytechnical University of Madrid, Madrid, Spain, in 1984, 1987, and 1995, respectively. Since 1984, he has been a Professor in the Electrical Engineering Department, Federal University of Espírito Santo. His research interests include highpower-factor rectifiers, active power filters, low-loss converters, and machine drives.

José Luiz F. Vieira (S’90–M’95) was born in Muqui, Brazil, in 1958. He received the B.E. degree from the Federal University of Espírito Santo, Vitória, Brazil, the M.S. degree from the Federal University of Rio de Janeiro, Rio de Janeiro, Brazil, and the Ph.D. degree from the Federal University of Santa Catarina, Florianópolis, Brazil, in 1981, 1986, and 1993, respectively, all in electrical engineering. He is presently a Titular Professor in the Electrical Engineering Department, Federal University of Espírito Santo, which he joined in 1982. He is a Member of the Power Electronics and Electric Drives Laboratory, where he conducts research on power electronics. His main research interests include high-efficiency power converters, switching-frequency power supplies, and lighting systems.