energies Article
Hybrid Modulation of Bidirectional Three-Phase Dual-Active-Bridge DC Converters for Electric Vehicles Yen-Ching Wang, Fu-Ming Ni and Tzung-Lin Lee * Department of Electric Engineering, National Sun Yat-sen University, Kaohsiung 80424, Taiwan;
[email protected] (Y.-C.W.);
[email protected] (F.-M.N.) * Correspondence:
[email protected]; Tel.: +886-7-525-2000 (ext. 4197) Academic Editors: Frede Blaabjerg and Dirk Söffker Received: 9 March 2016; Accepted: 23 June 2016; Published: 27 June 2016
Abstract: Bidirectional power converters for electric vehicles (EVs) have received much attention recently, due to either grid-supporting requirements or emergent power supplies. This paper proposes a hybrid modulation of the three-phase dual-active bridge (3ΦDAB) converter for EV charging systems. The designed hybrid modulation allows the converter to switch its modulation between phase-shifted and trapezoidal modes to increase the conversion efficiency, even under light-load conditions. The mode transition is realized in a real-time manner according to the charging or discharging current. The operation principle of the converter is analyzed in different modes and thus design considerations of the modulation are derived. A lab-scaled prototype circuit with a 48V/20Ah LiFePO4 battery is established to validate the feasibility and effectiveness. Keywords: DAB; LiFePO4 battery; EV charger
1. Introduction Fossil-fuel shortages and environmental concerns have strongly impacted the energy industries during the last decades. As one of the promising solutions to reduce the usage of the fossil fuel and carbon emissions, electric vehicles (EVs) are receiving much attention. Various infrastructures and charging standards for EVs have been presented considering different power levels as well as charging fields [1]. Among them, the direct-current charging standard, which is defined to deliver high power and high current charging capability, normally requires an off-board charger. Traditionally, a unidirectional dc converter is able to satisfy the requirement of refreshing battery energy. Recently, various economical and technical issues about using EVs to support the power grid have been discussed [2,3]. In this sense, power has to be delivered between the utility and EVs to meet demand response and/or take advantage of time-of-use electricity rates. On the other hand, EVs should be able to act as a standalone backup power in emergency situations [4–6]. Regarded as distributed energy resources, in this scenario EVs require an interface dc converter with both charging and discharging capabilities. The dual active bridge (DAB) dc converter is a widely used circuit topology intended for bidirectional power flow, due to its soft-switching characteristic, high efficiency and low volume [7]. The DAB converter is typically controlled by using so-called phase-shifted modulating method, in which two-level square waves with a designated phase shift are applied to the isolated transformer. In this sense, power flow and its direction can be controlled by the phase difference. However, this control method suffers from low conversion efficiency under light-load condition due to high conduction and switching losses. In order to improve the efficiency of the DAB converter, various modulating schemes have been presented. In [8,9], one of voltage pulses across on the transformer was replaced by a three-level pulse. In this sense, the DAB converter can operate in a wide-range soft-switching mode and its conduction loss can also be reduced by considering the reactive power Energies 2016, 9, 492; doi:10.3390/en9070492
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of the converter [10]. However, the required algorithm needs heavy computation to determine the of the converter However, the required algorithm needs heavy computation to determinewere the operating point [10]. of the converter. The so‐called triangular and trapezoidal modulations operating of the converter. The so-called triangular andthe trapezoidal modulations were proposed proposed point by imposing two three‐level voltage pulses on transformer [11,12]. By shaping the by imposing two three-level voltage pulsescan on the transformer [11,12]. switching By shapingand the transformer transformer current, both modulations provide zero‐current reduce the current, bothcurrent modulations can provide zero-current andof reduce the gain circulating currentrating. under circulating under light‐load conditions switching at the cost voltage and power light-load conditions at the cost of voltage gain and power rating. Moreover, various optimization Moreover, various optimization modulations have been proposed to reduce the peak current [10] or modulations have been proposed to reduce the peak current [10] or reactive power [13,14], but they reactive power [13,14], but they are not suitable for real‐time applications due to their complexity. are not suitable for real-time applications due to their complexity. In order for high‐power applications, the bidirectional three‐phase DAB (3ΦDAB) converter has order for [15]. high-power applications, thefeatures bidirectional three-phase DAB compared (3ΦDAB) converter been Inpresented The 3ΦDAB converter higher power density with the has been presented [15]. The 3ΦDAB converter features higher power density compared with the single‐phase DAB (1ΦDAB) converter. It is worth noting that the current frequency of the 3ΦDAB single-phase DAB times (1ΦDAB) converter. It is worth noting that the current frequency of the 3ΦDAB converter is three of that of the 1ΦDAB converter. Thus the volume of the switching‐ripple converter three times of that of the 1ΦDAB converter. Thus theHowever, volume of similarly the switching-ripple filters can is be significantly reduced in the 3ΦDAB converter. to 1ΦDAB filters can be significantly reduced with in thephase‐shifted 3ΦDAB converter. However, similarly to 1ΦDAB converter, converter, the 3ΦDAB converter modulation still performs poor efficiency at the 3ΦDAB converter with phase-shifted modulation still performs poor efficiency at low-power low‐power operation [16–21]. In [20], a hybrid modulation strategy was proposed for the 3ΦDAB operation [16–21]. In [20], a hybrid modulation strategy was proposed for the 3ΦDAB converter, converter, including phase‐shifted, triangular and trapezoidal modulations. The converter is able to including phase-shifted, triangular and trapezoidal modulations. The converter is able to operate operate at the highest efficiency with a changeable modulating scheme according to the operating at the highest efficiency with a changeable modulating scheme according to the operating point. point. This study was focused on the estimations and analysis of the various modulations, but no This study was focused on the estimations and analysis of the various modulations, but no information information was provided on how to determine optimizing modulation, which may be very difficult was provided on how to determine optimizing modulation, which may be very difficult in real-time in real‐time implementation. Furthermore, the considered triangular modulation may substantially implementation. Furthermore, the considered triangular modulation may substantially increase the increase the complexity of the modulation scheme. complexity of the modulation scheme. The authors have presented the 3ΦDAB converter with hybrid modulation in previous The authors have presented the 3ΦDAB converter with hybrid modulation in previous research [21]. The converter is able to switch its modulation between phase‐shifted and trapezoidal research [21]. The converter is able to switch itseven modulation between phase-shifted modes autonomously to maintain efficiency in the light‐load condition. In and this trapezoidal paper, we modes autonomously to maintain efficiency even in the light-load condition. In this paper, we further further focused on theoretical analysis of the mode transition based on loss analysis of the circuit. The condition of when the modulation modes switch is pre‐determined based on efficiency analysis. focused on theoretical analysis of the mode transition based on loss analysis of the circuit. The condition Design of modes the modulation also proposed and onthe corresponding of whenconsiderations the modulation switch is are pre-determined based efficiency analysis.suggested Design control progress with flowchart are for hybrid modulation is also provided. A lab‐scale considerations of the modulation also proposed and the corresponding suggested control platform progress consisting of a 1.2 kW 3ΦDAB converter with a 48V/20Ah LiFePO 4 battery was established to verify with flowchart for hybrid modulation is also provided. A lab-scale platform consisting of a 1.2 kW the proposed method. 3ΦDAB converter with a 48V/20Ah LiFePO4 battery was established to verify the proposed method. 2. Review of DAB Operation 2. Review of DAB Operation Figure 1 shows the 3ΦDAB converter. Two sets of three-phase half bridges are deployed with a Figure 1 shows the 3ΦDAB converter. Two sets of three‐phase half bridges are deployed with a three-phase three‐phase isolated isolated transformer. transformer. Various Various modulating modulating strategies strategies have have been been presented. presented. The The phase-shift phase‐shift modulation is able to provide zero-voltage switching (ZVS) for all switches, but this feature may modulation is able to provide zero‐voltage switching (ZVS) for all switches, but this feature may lose lose in case of wide voltage variation. In order to assure in light load operation, in case of wide voltage gain gain and and load load variation. In order to assure ZVS ZVS in light load operation, the the trapezoidal modulation was presented [12,13,20,22]. Here, we briefly review both modulations. trapezoidal modulation was presented [12,13,20,22]. Here, we briefly review both modulations.
Figure 1. The bidirectional three‐phase DAB DC/DC converter. Figure 1. The bidirectional three-phase DAB DC/DC converter.
2.1. 3ΦDAB Phase‐Shift Modulation 2.1. 3ΦDAB Phase-Shift Modulation In phase‐shift modulation, the switches in the primary‐side legs are designed to operate at 50% In phase-shift modulation, the switches in the primary-side legs are designed to operate at duty ratio, while all three legs are with a 120° phase shift among them. On the other hand, the 50% duty ratio, while all three legs are with a 120˝ phase shift among them. On the other hand, secondary‐side legs are controlled with designated phase shift with respect to the primary‐side legs for proper power transfer between two dc sides. Figure 2a shows the theoretical waveforms of both
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the secondary-side legs are controlled with designated phase shift with respect to the primary-side legs for proper and power transfer between two dc sides. Figure 2a The shows theoretical waveforms phase voltages phase current with phase‐shift modulation. Vanthe is a‐phase voltage in the of both phase voltages and phase current with phase-shift modulation. The V is a-phase voltage an primary side and Vrm’ is the secondary‐side r‐phase voltage Vrm referred into primary‐side. As can be in the primary side and Vrm ’ is the secondary-side r-phase voltage Varm referred into primary-side. seen, both phase voltages show a six‐step square wave. The variable i is the a‐phase current that is As can be seen, both phase voltages show a six-step square wave. The variable is the the a‐phase a-phase determined according to voltage difference applied on the Leqa, which is defined iaas current that is determined according to voltage difference applied on the Leqa ,a + N which is defined 2Lr). Thus the equivalent inductance referred to the primary side. In this case, L eqa is equal to (L as the a-phase equivalent inductance referred to the primary side. In this case, L is equal to eqa power transfer PD can be determined based on integrating both voltage and current. As shown in 2 (La + N Lr ).(1) Thus power PD can determined based onδ integrating bothsides voltage Equations and the (2), PD is transfer dependent on be the phase‐shift angle between two of and the current. As shown in Equations (1) and (2), P is dependent on the phase-shift angle δ between two D transformer [18,21]: sides of the transformer [18,21]:
dVi 2 2 2 )2 , for 0 2 ( δ π 2δ dV i ¨ p3 ´2 q, for 0 ď δ ď 3 PD “ Leqa PD
3
ωLeqa
2π
3
dV PD dVi i 2 ( δ2 )π , for π PD “ L ¨ pδ ´ 18 ´ q, for 3ď δ ď ωLeqaeqa π 18 3 2
(1) (1)
2
2 2π 33
(2) (2)
(a)
(b)
Figure 2. Theoretical waveforms of 3ΦDAB converter. (a) Phase‐shifted modulation; (b) Trapezoidal modulation. Figure 2. Theoretical waveforms of 3ΦDAB converter. (a) Phase-shifted modulation; (b) Trapezoidal modulation.
The parameter d is defined as the voltage gain, equal to (NVbat/Vi), and ω is the switching frequency in radians per second. The theoretical phase‐shifted angle δ for a corresponding P D also The parameter d is defined as the voltage gain, equal to (NVbat /Vi ), and ω is the switching can be expressed as the following Equations (3) and (4): frequency in radians per second. The theoretical phase-shifted angle δ for a corresponding PD also can be expressed as the following Equations (3) and (4): 4 d Vi 2 16 d 2 2Vi 4 72 d LeqaVi 2 PD , for 0 b 2 3 4dπVi2 ´ 16d2 π62dV Vi4i ´ 72dπωLeqa Vi2 PD π δ“ , for 0 ď δ ď 3 6dVi2
18d Vi 324d Vi 72d Vi 1296d LeqaVi PD 2
δ“
18dπVi2
2
2
2
2
2
4
(3) (3)
2
2 b ,2 for 2 4 2 2 2 2 2 3 2π ´ 324d π 36 Vi dV ´i 72d π Vi ´ 1296dπωLeqa Vi PD 3 π , for ď δ ď 3 3 36dVi2
(4) (4)
2.2. 3ΦDAB Trapezoidal Modulation 2.2. 3ΦDAB Trapezoidal Modulation For 3ΦDAB trapezoidal modulation, the two phases of the converter operate in parallel, which For 3ΦDAB trapezoidal modulation, the two phases of the converter operate in parallel, which can can be viewed as 1ΦDAB trapezoidal modulation. Figure 3 shows the equivalent circuit. Since the be viewed as 1ΦDAB trapezoidal modulation. Figure 3 shows the equivalent circuit. Since the b-phase b‐phase and c‐phase are in parallel, Leqb and Leqc are in parallel connection. Therefore, the total equivalent inductance LeqT can be expressed as Equation (5):
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and c-phase are in parallel, Leqb and Leqc are in parallel connection. Therefore, the total equivalent inductance LeqT can be expressed as Equation (5): LeqT “ Leqa ` Leqb,c “ 1.5 ¨ Leqa
(5)
where Leqb,c is equal to 0.5Leqa assuming equal three-phase total equivalent inductances, Leqa , Leqb and Leqc . Figure 2b shows the typical voltage and current waveforms of trapezoidal modulation. Energies 2016, 9, 492 4 of 13 The power conversion PT can be expressed as Equations (6) and (7) [18,21]. Controllable variables include phase-shifted angle δT , the primary- and secondary-side duty cycles D1 , D2 . LeqT Leqa Leqb , c 1.5 Leqa (5) Vi 2 PT “ ¨ rD1 2 ´ dpD1 ´ δT q2 s (6) where Leqb,c is equal to 0.5Leqa assuming equal three‐phase total equivalent inductances, L eqa, Leqb and 2πωLeqT Leqc. Figure 2b shows the typical voltage and current waveforms of trapezoidal modulation. The D power conversion PT can be expressed as Equations (6) and (7) [18,21]. Controllable variables include D2 “ 1 (7) d phase‐shifted angle δT, the primary‐ and secondary‐side duty cycles D 1, D2.
Figure 3. The equivalent circuit with trapezoidal modulation. Figure 3. The equivalent circuit with trapezoidal modulation.
Due to parallel operation, voltage and side of the transformer have the Vi 2current2in the primary P [ D1 d ( D1 T )2 ] (6) T following relationship: 2L eqT
Vbn
ib “ ic “ ´0.5i a D1 2 “ Vcn “D´0.5V d an “ ´Vi {3
(8)
(7) (9)
Thus PT could be simple controlled by two variables δT and D1 . Once the variable δT is decided, Due to parallel operation, voltage and current in the primary side of the transformer have the the duty cycle D1 can be obtained as shown in Equation (10): following relationship: b D1 “
´dδT `
i b i c 02 . 5 i a 2 2 d2 δ T 2 ` p1 ´ dqVi ¨ pdδT Vi ` 2πωLeqT PT q 1´d
Vbn V cn 0 .5V an Vi / 3
(8) (10) (9)
3. Hybrid Modulation Design Thus PT could be simple controlled by two variables δT and D1. Once the variable δT is decided, In this paper, a hybrid modulating strategy was proposed to allow the converter to be able the duty cycle D 1 can be obtained as shown in Equation (10): to autonomously switch between phase-shift and trapezoidal modes according to output current. 2 2 d T dis2pre-determined T 2Vi 2 on 2efficiency LeqT PT ) analysis, which includes The turning point for mode transition T (1 d )Vi ( dbased (10) D1 device loss and magnetic loss. The former part consists of both conduction and switching losses. 1 d The latter one is estimated based on the datasheet of core material. Table 1 shows the circuit parameters used in this paper. The primary-side and secondary-side switches are IGBTs (IXGK60N60BsD1, 600 V, 3. Hybrid Modulation Design 75 A, IXYS Co., Ltd., Milpitas, CA, USA) and MOSFETs (IXFK120N20P, 200 V, 120 A, IXYS Co., Ltd., In this paper, a hybrid modulating strategy was proposed to allow the converter to be able to Milpitas, CA, USA), respectively. autonomously switch between phase‐shift and trapezoidal modes according to output current. The turning point for mode transition is pre‐determined based on efficiency analysis, which includes device loss and magnetic loss. The former part consists of both conduction and switching losses. The latter one is estimated based on the datasheet of core material. Table 1 shows the circuit parameters used in this paper. The primary‐side and secondary‐side switches are IGBTs (IXGK60N60BsD1, 600 V, 75 A, IXYS Co., Ltd., Milpitas, CA, USA) and MOSFETs (IXFK120N20P, 200 V, 120 A, IXYS Co., Ltd., Milpitas, CA, USA), respectively.
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Table 1. Circuit parameters.
Table 1. Circuit parameters. Parameter Name Value Parameter Name Value Prated Rated power 1.2 Prated Rated power 1.2 fs Switching frequency 20 Switching frequency 20 350 Vi fs Input voltage Input voltage 350 Vbat Vi Output voltage 42–58.4 bat Output voltage 42–58.4 V Csnubber Snubber capacitors 1.0 snubber Snubber capacitors 1.0 215 La LC HVS auxiliary inductors b Lc HVS auxiliary inductors 215 6 Lr LLsa L Ltb Lc LVS auxiliary inductors LVS auxiliary inductors 6 65 Lr Ls Lt Lleakage Leakage inductance Leakage inductance 65 6 NLleakage Turn ratio Turn ratio 6 265 Rp N Primary-side winding resistance of the transformor Primary‐side winding resistance of the transformor 265 102 RS Rp Secondary-side winding resistance of the transformor Rds RS Conductor Resistance of secondary-side switches Secondary‐side winding resistance of the transformor 102 22 Conductor Resistance of secondary‐side switches 22 Rds
Unit Unit kW kW kHz kHz Volt Volt Volt Volt nF nF µH μH µH μH µH μH ‐ mΩ mΩ mΩ mΩ mΩ mΩ
3.1. Loss and Efficiency 3.1. Loss and Efficiency For EV application, the charging operation with adjustable charging current according to the For EV application, the charging operation with adjustable charging current according to the operator is the main focus. That is why we design the converter for high efficiency from light load operator is the main focus. That is why we design the converter for high efficiency from light load to to heavy load. The power flow of the proposed DAB circuit can be controlled between the primary heavy load. The power flow of the proposed DAB circuit can be controlled between the primary and secondary sides, so the theoretical analysis has addressed dual operation for both charging and and secondary sides, so the theoretical analysis has addressed dual operation for both charging and discharging behavior. Figure 4 shows phase-shifted angle andand dutyduty ratioratio withwith respect to output current discharging behavior. Figure 4 shows phase‐shifted angle respect to output forcurrent for both modulating modes, which were calculated by adjusting phase‐shifted angle. Based both modulating modes, which were calculated by adjusting phase-shifted angle. Based on the modulation parameters in Figure 4, converter loss and its efficiency can be derived. on the modulation parameters in Figure 4, converter loss and its efficiency can be derived.
(a)
(b)
Figure 4. 4. Modulation Modulation parameters Figure parameters with with respect respect to to current. current. (a) (a) Phase‐shifted Phase-shiftedmodulation; modulation; (b) Trapezoidal modulation. (b) Trapezoidal modulation.
The 50% duty ratio makes phase‐shifted modulation less loss in high current operation in The 50% duty ratio makes phase-shifted modulation less loss in high current operation in Figure 5a. Figure 5a. However, the phase‐shifted modulation suffers from high switching loss in low current However, the phase-shifted modulation suffers from high switching loss in low current range due range due to loss of ZVS. On the other hand, the trapezoidal modulation benefits in low current to range due to reduced circulating current in Figure 5b, but the conduction loss significantly increases loss of ZVS. On the other hand, the trapezoidal modulation benefits in low current range due to reduced circulating current in Figure 5b, but the conduction loss significantly increases with increasing with increasing output current. Therefore, the crossover point of the efficiency curves in Figure 5d is output current. Therefore, the crossover point of the efficiency curves in Figure 5d is located around 2 A, located around 2 A, which is a preferred selection of the mode‐switching current I mode. Despite the which is aloss preferred selection of the mode-switching current Imode . Despite themethod power to loss equations power equations and simulation results, we still use the trial‐and‐error locate the and simulation results, we still use the trial-and-error method to locate the crossover point of the crossover point of the efficiency curves and select 2 A as the mode‐switching current Imode in the efficiency curves and select 2 A as the mode-switching current Imode in the experiment. The efficiency experiment. The efficiency curves and the preferred mode‐switching current I mode is measured and curves and the preferred mode-switching current Imode is measured and verified in the experimental verified in the experimental results in a later section. results in a later section.
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(a)
(b)
(c)
(d)
Figure 5.5. Simulation Simulation result result of of the the power power loss loss and and efficiency. efficiency. (a) (a) Detailed Detailed loss loss with with phase-shift; phase‐shift; Figure (b) Detailed loss with trapezoidal; (c) The curves of power loss; (d) Efficiency curve. (b) Detailed loss with trapezoidal; (c) The curves of power loss; (d) Efficiency curve.
Because the experimental platform is over‐designed for protection, the operation range where Because the experimental platform is over-designed for protection, the operation range where the the trapezoidal modulation provides better efficiency is limited. If the 600V/40A IGBTs and trapezoidal modulation provides better efficiency is limited. If the 600V/40A IGBTs and 200V/50A 200V/50A MOSFETs are chosen, the crossover point will up to 3 A but the platforms will have MOSFETs are chosen, the crossover point will up to 3 A but the platforms will have higher risk. higher risk. 3.2. Control Scheme 3.2. Control Scheme Figure 6a shows the proposed control block diagram, including battery charging scheme and Figure 6a shows the proposed control block diagram, including battery charging scheme and hybrid modulator. The logical switches shown are to explain control flow only. Both constant-current hybrid The (CV) logical switches shown inare explain control flow only. (CC) and modulator. constant-voltage modes are designed the to battery charging algorithm. In theBoth CC constant‐current constant‐voltage (CV) modes are designed the battery mode, logic switch(CC) SCV isand set at off-state so that the modulator input is simplyin determined bycharging current algorithm. In the CC mode, logic switch S CV is set at off‐state so that the modulator input is simply command Icomd only. On the other hand, battery voltage Vbat is required for CV operation. As shown, determined by current command I comd only. On the other hand, battery voltage Vbat is required for CV logic switch SCV is changed to on-state to produce the modified current ICV that is added to Icomd for operation. As shown, logic switch S CV is changed to on‐state to produce the modified current ICV that the hybrid modulator. is added to I comd for the hybrid modulator. Based on the pre-determined current turning-point Imode , a suitable modulation, phase-shifted or trapezoidal mode, is selected to generate a phase-shifted angle and/or duty ratio. As shown, in phase-shifted modulation, the logic switch SPS is turned on to allow a PI controller generating phase angle command δ according to the current command Icomd . On the other hand, duty ratio command D1 can be produced by turning on STA for the trapezoidal mode. The required phase angle δT is the phase-shifted angle at current turning-point Imode . Figure 6b shows the mode transition of the hybrid
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modulator, where Imode is set as 2 A. As shown, δ and D1 are controllable variables for the current command larger and less than 2 A, respectively. Energies 2016, 9, 492 7 of 13
(a)
(b)
Figure 6. The control scheme and hybrid modulation. (a) The control scheme of the converter; (b) Phase angle and duty cycle of hybrid modulation.
Based on the pre‐determined current turning‐point Imode, a suitable modulation, phase‐shifted or trapezoidal mode, is selected to generate a phase‐shifted angle and/or duty ratio. As shown, in phase‐shifted modulation, the logic switch S PS is turned on to allow a PI controller generating phase (b) (a) angle command δ according to the current command Icomd. On the other hand, duty ratio command Figure 6. The control scheme and hybrid modulation. (a) The control scheme of the converter; Figure 6. The control scheme and hybrid modulation. (a) The control scheme of the converter; (b) Phase D1 can be produced by turning on S TA for the trapezoidal mode. The required phase angle δ T is the (b) Phase angle and duty cycle of hybrid modulation. angle and duty cycle of hybrid modulation. phase‐shifted angle at current turning‐point Imode. Figure 6b shows the mode transition of the hybrid modulator, where Imode is set as 2 A. As shown, δ and D1 are controllable variables for the current Based on the pre‐determined current turning‐point Imode, a suitable modulation, phase‐shifted or command larger and less than 2 A, respectively. There are two variables, duty cycle D1 and phase angle δT in trapezoidal modulation. trapezoidal mode, is selected to generate a phase‐shifted angle and/or duty ratio. As shown, in There are from two phase-shift variables, duty cycle D1 and phase angle T in trapezoidal modulation. When When switching to trapezoidal modulation, the δphase angle δ generated by phase-shift phase‐shifted modulation, the logic switch S PS is turned on to allow a PI controller generating phase switching from phase‐shift to trapezoidal modulation, the phase angle δ generated by phase‐shift modulation is sent to trapezoidal modulation as a fixed variable. During the trapezoidal modulation, angle command δ according to the current command I comd. On the other hand, duty ratio command is sent beto changed trapezoidal modulation as S a PSfixed variable. During trapezoidal themodulation phase angle won’t by the PI because the in Figure 6a is off. This the hybrid modulator D1 can be produced by turning on STA for the trapezoidal mode. The required phase angle δ T is the modulation, the phase angle won’t be changed by the PI because the S PS in Figure 6a is off. This design reduces the bouncing problem betweenmode modulations. phase‐shifted angle at current turning‐point I . Figure 6b shows the mode transition of the hybrid hybrid modulator design reduces the bouncing problem between modulations. Figure 7 shows the ofA. As the proposed control After obtaining a battery charging modulator, where I modeflowchart is set as 2 shown, δ and D1scheme. are controllable variables for the current Figure 7 shows the the proposed control scheme. obtaining or discharging command, theflowchart converterof determines its operation based onAfter battery voltage. a Inbattery charging command larger and less than 2 A, respectively. charging or discharging command, the converter determines its operation based on battery voltage. case, CC mode is two normally chosen and theDhybrid modulator the modulating There are variables, duty cycle 1 and phase angle δdetermines T in trapezoidal modulation. strategy When In charging case, CC mode is normally chosen and the hybrid modulator determines the modulating switching to trapezoidal modulation, the phase angle by phase‐shift according to from Imode .phase‐shift At the end of charging, CV is started instead to δ generated decrease charging current. strategy according to Imode. At the end of charging, CV is started instead to decrease charging current. is sent to trapezoidal modulation fixed the trapezoidal Themodulation charging process is ended as the charging currentas is a less than variable. 0.4 A. ForDuring discharging requirement, The charging process is ended as the charging current is less than 0.4 A. For discharging requirement, modulation, the phase angle won’t be changed by the PI the SPS in Figure is preventing off. This only current command is needed. A low-voltage limitation 42because V is considered in this case6a for only current command is needed. A low‐voltage limitation 42 V is considered in this case for hybrid modulator design reduces the bouncing problem between modulations. battery from over-discharging. preventing battery from over‐discharging. Figure 7 shows the flowchart of the proposed control scheme. After obtaining a battery charging or discharging command, the converter determines its operation based on battery voltage. In charging case, CC mode is normally chosen and the hybrid modulator determines the modulating strategy according to Imode. At the end of charging, CV is started instead to decrease charging current. The charging process is ended as the charging current is less than 0.4 A. For discharging requirement, only current command is needed. A low‐voltage limitation 42 V is considered in this case for preventing battery from over‐discharging.
Figure 7. The flowchart of the proposed control scheme. Figure 7. The flowchart of the proposed control scheme.
Figure 7. The flowchart of the proposed control scheme.
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4. 4. Experimental Results Experimental Results AnAn experimental prototype is established to measure efficiency of theof three-phase DAB converter experimental prototype is established to measure efficiency the three‐phase DAB with the hybrid modulation, experimental areparameters shown in Table 1. Considering power converter with the hybrid and modulation, and parameters experimental are shown in Table 1. Considering power rating, voltage rating and over‐design for protection, the converter circuit use rating, voltage rating and over-design for protection, the converter circuit use IGBTs (IXGK60N60BsD1, IGBTs V, 75 and A) as primary‐side switches and (IXFK120N20P, 600 V, 75 (IXGK60N60BsD1, A) as primary-side600 switches MOSFETs (IXFK120N20P, 200MOSFETs V, 120 A) as secondary-side 200 V, 120 A) as secondary‐side switches. A DC power supply is applied to theprimary side while switches. A DC power supply is applied to theprimary side while the LiFePO4 battery (TY-D 20-48, the LiFePO 4 battery (TY‐D 20‐48, 20AH, Ty Dynamic Co., Ltd., New Taipei City, Taiwan) is applied 20AH, Ty Dynamic Co., Ltd., New Taipei City, Taiwan) is applied to the secondary side. A three-phase to the secondary side. A transformer is composed by three single‐phase ferrite‐core transformer is composed bythree‐phase three single-phase ferrite-core transformers (ferrite core, EE55-28-21-MB3, transformers (ferrite core, EE55‐28‐21‐MB3, New Industry Co., of Ltd., Taiwan). The New Favor Industry Co., Ltd., Taipei, Taiwan). TheFavor leakage inductance theTaipei, transformer is referred leakage inductance of the transformer is referred to the high‐voltage side. The control strategy of the to the high-voltage side. The control strategy of the converter is accomplished via usage of a digital converter is accomplished via usage of a digital signal processors (DSPs, TMS320C28335, Texas signal processors (DSPs, TMS320C28335, Texas Instruments, Dallas, TX, USA). Instruments, Dallas, TX, USA). 4.1. Steady State 4.1. Steady State Figure 8a shows Van , Vrm , ia waveforms when the converter is operated at IB * = 1 A. Obviously, Figure 8a shows Van, Vrm, ia waveforms when the converter is operated at IB* = 1 A. Obviously, trapezoidal modulation is enabled with parameters D1 = 23.5% (11.75 µs), D2 = 25.86% (11.75 µs) and trapezoidal modulation is enabled with parameters D1 = 23.5% (11.75 μs), D2 = 25.86% (11.75 μs) and δT = 5.2˝ (0.72 µs). Note that the transformer current ia is approximately equal to 0.1 A due to dead δT = 5.2° (0.72 μs). Note that the transformer current ia is approximately equal to 0.1 A due to dead time. Figure 8b shows key waveforms in phase-shifted modulation at I*B * = 10 A, in which the phase time. Figure 8b shows key waveforms in phase‐shifted modulation at I B = 10 A, in which the phase ˝ . Figure 8c shows the waveforms also in phase-shifted modulation but with I * = ´10 A angle δ is 30 angle δ is 30°. Figure 8c shows the waveforms also in phase‐shifted modulation but with IB*B = −10 A discharging current. discharging current.
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(b) Figure 8. Cont.
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Figure 8. Steady‐state waveforms. (a) Trapezoidal modulation with 1 A charging current; Figure 8. Steady-state waveforms. waveforms. (a) (a)Trapezoidal Trapezoidalmodulation modulationwith with 1 1 A A charging charging current; current; Figure 8. Steady‐state (b) Phase‐shifted modulation with 10 A charging current; (c) Phase‐shifted modulation with 10 A (b) Phase-shifted modulation with with 10 10 A A charging charging current; current; (c) (c) Phase‐shifted Phase-shifted modulation modulation with with 10 10 A A (b) Phase‐shifted modulation discharging current. discharging current. discharging current.
4.2. Transient Behaviour 4.2. Transient Behaviour 4.2. Transient Behaviour Figure 9a illustrates the transient waveform when the charging current command Ibat* increases * increases Figure 9a illustrates the transient waveform when the charging current command Ibat * increases Figure 9a illustrates the transient waveform when the charging current command I bat from 1 to 4 A. As expected, the modulation is successfully switched from trapezoidal mode to from 1 to 4 A. As expected, the modulation is successfully switched from trapezoidal mode to to from 1 to 4 A. As expected, the modulation is successfully switched from trapezoidal mode phase‐shifted mode. Figure 9b shows the transient behavior when the charging current command phase-shifted mode. Figure 9b shows the transient behavior when the charging current command phase‐shifted mode. Figure 9b shows the transient behavior when the charging current command Ibat* is increased from 0 to 4, 6 and 8 A, respectively. As shown, the charging current Ibat can is able to * is increased from 0 to 4, 6 and 8 A, respectively. As shown, the charging current I IIbat can is able to * is increased from 0 to 4, 6 and 8 A, respectively. As shown, the charging current I batfollow the current command I bat bat*. bat can is able to *. follow the current command I * bat In normal operation, the comd is tuned slowly in constant‐voltage mode, so follow the current command I batcurrent command I . InPI normal operation, the current command Icomd isis tuned slowly in in constant-voltage mode, so the the In controller in the hybrid modulator can designed for fast‐tracking ability with normal operation, the current command I comd be tuned slowly constant‐voltage mode, so PI controller in the hybrid modulator can be designed for fast-tracking ability with hard-to-notice hard‐to‐notice overshoot transient. Since the transient experiments are extreme case of the PI controller in the hybrid modulator can be designed for fast‐tracking ability with overshoot transient. Since the transient experiments are caseactually of current-command-I current‐command‐I comd adjustment, the current in extreme Figure 9b the overshoot hard‐to‐notice overshoot transient. Since the spikes transient experiments are is extreme case comd of expressed in terms of 2 s/div, which is induced by the fast‐tracking PI controller. adjustment, the current spikes in Figure 9b actually is the overshoot expressed in terms of 2 s/div, current‐command‐Icomd adjustment, the current spikes in Figure 9b actually is the overshoot which is induced by the fast-tracking PI controller. expressed in terms of 2 s/div, which is induced by the fast‐tracking PI controller.
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(a) Figure 9. Cont.
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Figure 9. Transient waveforms. (a) Waveforms when the modulation switches; (b) Waveforms when Figure 9. Transient waveforms. (a) Waveforms when the modulation switches; (b) Waveforms when Figure 9. Transient waveforms. (a) Waveforms when the modulation switches; (b) Waveforms when charging current changes. charging current changes. charging current changes.
4.3. CV Operation 4.3. CV Operation 4.3. CV Operation Figure 10 shows shows battery battery voltage voltage V Vbat and the converter converter Figure 10 and battery battery current current when when the the operation operation of of the Figure 10 shows battery voltage Vbat bat and battery current when the operation of the 0–t converter changes from CC mode to CV mode. The converter runs in CC mode at I bat* = 10 A during t 1 period, * changes from CC mode to CV mode. The converter runs in CC mode at* Ibat = 10 A during t0 –t1 changes from CC mode to CV mode. The converter runs in CC mode at I bat = 10 A during t0–t1 period, which is modulated by the phase‐shifted scheme. When the battery voltage is equal to 58.4 V at t 1, period, which is modulated by the phase-shifted scheme. When the battery voltage is equal to 58.4 V which is modulated by the phase‐shifted scheme. When the battery voltage is equal to 58.4 V at t 1, the mode switches to CV mode. During the the CV mode, in at t1charging , the charging mode switches to CV mode. During CV mode,the theconverter converterstill still operates operates in the charging mode switches to CV mode. During the CV mode, the converter still operates in phase‐shifted modulation modulation during –t2 period since the charging current I mode = 2 A. phase-shifted during tt1–t period since the charging current I bat is larger than I is larger than Imode = 2 A. phase‐shifted modulation during t11–t22 period since the charging current Ibat bat is larger than Imode = 2 A. When I bat is less than 2 A, the converter changes its modulation to the trapezoidal mode during the When Ibat is less than 2 A, the converter changes its modulation to the trapezoidal mode during the bat is less than 2 A, the converter changes its modulation to the trapezoidal mode during the ttWhen I 2–t3 period. After Ibat is smaller than 0.4 A at t3, the battery charging is terminated. –t period. After Ibat is smaller than 0.4 A at t3 , the battery charging is terminated. t22–t33 period. After Ibat is smaller than 0.4 A at t3, the battery charging is terminated.
Figure 10. The CC/CV waveform. Figure 10. The CC/CV waveform. Figure 10. The CC/CV waveform.
4.4. Efficiency 4.4. Efficiency 4.4. Efficiency The measured efficiency curve is shown in Figure 11a. The efficiency is improved as the The measured measured efficiency is shown in efficiency Figure 11a. The efficiency is improved as the The efficiency isexample, shown in the Figure 11a. The isfrom improved as to the88.66% charging charging current less than 2 curve A. curve For is efficiency increased 83.04% at charging current less than 2 A. For example, the efficiency is increased from 83.04% to 88.66% at current less than 2 A. For example, the efficiency is increased from 83.04% to 88.66% at I = 1 A. mode Imode = 1 A. Figure 11b gives detailed measurements of the power loss in phase‐shifted mode and the Iresult in the trapezoidal mode for light load condition. Label on1 and on2 stand for the turn‐on loss mode = 1 A. Figure 11b gives detailed measurements of the power loss in phase‐shifted mode and the result in the trapezoidal mode for light load condition. Label on1 and on2 stand for the turn‐on loss of primary‐ and secondary‐side switches, respectively; label off1 and off2 stand for the turn‐off loss of primary‐ and secondary‐side switches, respectively; label off1 and off2 stand for the turn‐off loss
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Figure 11b gives detailed measurements of the power loss in phase-shifted mode and the result Energies 2016, 9, 492 11 of 13 in the trapezoidal mode for light load condition. Label on1 and on2 stand for the turn-on loss of Energies 2016, 9, 492 11 of 13 primary- and secondary-side switches, respectively; label off1 and off2 stand for the turn-off loss of of primary‐ secondary‐side switches, respectively; the label cond1 and cond2 stand for the primaryand and secondary-side switches, respectively; the label cond1 and cond2 stand for the conduction of primary‐ and secondary‐side switches, respectively; the label cond1 and cond2 stand for the conduction loss and of primary‐ and secondary‐side switches, label respectively; label Tr stands for loss. the loss of primarysecondary-side switches, respectively; Tr stands for the transformer conduction loss of primary‐ and secondary‐side switches, respectively; label Tr stands for the transformer loss. Transformer loss occupies over 60% loss, which might be improved by using low Transformer loss occupies over 60% loss, which might be improved by using low core loss material. transformer loss. Transformer loss occupies over 60% loss, which might be improved by using low core loss material. This results verifies the availability of hybrid method. This results verifies the availability of hybrid method. core loss material. This results verifies the availability of hybrid method.
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Figure 11. Analysis of efficiency and loss. (a) Measured efficiency of the converter with hybrid Figure 11. Analysis of efficiency and loss. (a) Measured efficiency of the converter with hybrid Figure 11. Analysis of efficiency and loss. (a) Measured efficiency of the converter with hybrid modulation; (b) Histogram of the power loss. modulation; (b) Histogram of the power loss. modulation; (b) Histogram of the power loss.
Figure 12 shows the charging and discharging efficiency curves with phase‐shift modulation, Figure 12 shows the charging and discharging efficiency curves with phase‐shift modulation, Figure 12 shows the charging and discharging efficiency curves phase-shift modulation, where the overall efficiency during discharging operation is 0.5% to with 1% lower than the charging where the overall efficiency during discharging operation is 0.5% to 1% lower than the charging where the overall efficiency during discharging operation is 0.5% to 1% lower than the charging efficiency. The voltage gain d becomes reciprocal and decreases when discharging from low voltage efficiency. The voltage gain d becomes reciprocal and decreases when discharging from low voltage efficiency. The voltage gain d becomes reciprocal and decreases when discharging from low voltage to to high voltage. According to power transfer Equations (1) and (2), the phase‐shift angle δ increases to high voltage. According to power transfer Equations (1) and (2), the phase‐shift angle δ increases high voltage. According to power transfer Equations (1) and (2), the phase-shift angle δ increases based based on the decreased voltage gain d while maintaining the same transfer power, so the peaks of based on the decreased voltage gain d while maintaining the same transfer power, so the peaks of on the decreased voltage gain dcausally while maintaining theaggravate same transfer soloss the peaks of transfer transfer three phase current raises which the power, transfer and reduce the transfer three phase current causally raises which aggravate the transfer loss and reduce the three phase current causally raises which aggravate the transfer loss and reduce the efficiency slightly. efficiency slightly. efficiency slightly.
Figure 12. Measured charging and discharging efficiency. Figure 12. Measured charging and discharging efficiency. Figure 12. Measured charging and discharging efficiency.
5. Conclusions 5. Conclusions This paper proposes and realizes a hybrid modulation strategy designed for the bidirectional This paper proposes and realizes a hybrid modulation strategy designed for the bidirectional three‐phase dual‐active bridge (3ΦDAB) DC converter designed for EV charger applications. The three‐phase dual‐active bridge (3ΦDAB) DC converter designed for EV charger applications. The hybrid modulation, which consists of the phase‐shifted and trapezoidal modulation, covers the hybrid modulation, which consists of the phase‐shifted and trapezoidal modulation, covers the
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5. Conclusions This paper proposes and realizes a hybrid modulation strategy designed for the bidirectional three-phase dual-active bridge (3ΦDAB) DC converter designed for EV charger applications. The hybrid modulation, which consists of the phase-shifted and trapezoidal modulation, covers the every power loading condition with the most efficient method and increases the over-all efficiency. The proposed control strategy applies two different modulations in on-time operation according to the magnitude of the charging/discharging current. A corresponding suggested control flowchart for hybrid modulation is also provided. A lab-scaled converter prototype with a 48V/20Ah LiFePO4 battery is established to validate the feasibility and the effectiveness of the proposed modulation method. The results show that efficiency can be increased under light load conditions. Acknowledgments: This work is funded by the Ministry of Science and Technology of Taiwan under grant 104-2221-E-110-037-. Author Contributions: Fu-Ming Ni, Yen-Ching Wang and Tzung-Lin Lee conceived and designed the experiments; Fu-Ming Ni performed the experiments; Fu-Ming Ni and Yen-Ching Wang analyzed the data; Tzung-Lin Lee contributed reagents/materials/analysis tools; Yen-Ching Wang wrote the paper. Conflicts of Interest: The authors declare no conflict of interest.
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