Microwave Liquid Crystal Technology - MDPI

crystals Review

Microwave Liquid Crystal Technology Holger Maune *,† , Matthias Jost † , Roland Reese † , Ersin Polat Rolf Jakoby

, Matthias Nickel

and

Institute for Microwave Engineering and Photonics, Technische Universität Darmstadt, 64283 Darmstadt, Germany; [email protected] (M.J.); [email protected] (R.R.); [email protected] (E.P.); [email protected] (M.N.); [email protected] (R.J.) * Correspondence: [email protected]; Tel.: +49-6151-16-28450 † These authors contributed equally to this work. Received: 2 August 2018; Accepted: 28 August 2018; Published: 5 September 2018

Abstract: Tunable Liquid Crystal (LC)-based microwave components are of increasing interest in academia and industry. Based on these components, numerous applications can be targeted such as tunable microwave filters and beam-steering antenna systems. With the commercialization of first LC-steered antennas for Ku -band e.g., by Kymeta and Alcan Systems, LC-based microwave components left early research stages behind. With the introduction of terrestrial 5G communications systems, moving to millimeter-wave communication, these systems can benefit from the unique properties of LC in terms of material quality. In this paper, we show recent developments in millimeter wave phase shifters for antenna arrays. The limits of classical high-performance metallic rectangular waveguides are clearly identified. A new implementation with dielectric waveguides is presented and compared to classic approaches. Keywords: liquid crystal; millimeter wave devices; steerable antennas; dielectric waveguide; rectangular waveguides; communication systems

1. Introduction Liquid Crystal (LC) technology paved the way for today’s communication and information industry. It found wide application in display technology and enabled modern devices for mobile computing such as smartphones, being present everywhere. Most of these devices (except for OLED) are based on the birefringence properties of LC. In the optical domain LC is packed between two polarization grids and is used to adaptively turn the polarization angle of light transmitted perpendicular to the LC’s surface. This twisted nematic (TN) cell is the foundation principle for all modern liquid crystal displays (LCD). This principle has been developed in Switzerland and the US beginning in the 1960s and patented early 1970s [1,2], respectively. Figure 1 shows the fundamentals of most of today’s LCDs. Birefringence is a material property which is known in microwave engineering as anisotropy represented by the permittivity tensor 

ε uu  ε˜ =  ε vu ε wu

ε uv ε vv ε wv

 ε uw  ε vw  · ε 0 ε ww

(1)

within the material’s coordinate system (uvw). The main differences compared to optical applications are founded in the much larger wavelength (up to several cm instead of less than µm). The rotation of the polarization angle is a niche application at microwaves. There, the anisotropy is usually adopted in order to present a variable effective permittivity ε eff to the electromagnetic wave. In addition, specific Crystals 2018, 8, 355; doi:10.3390/cryst8090355

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Light on

Light off

electromagnetic design rules must be fulfilled. Especially impedance matching is critical for the device’s dimensions. In the following section, the microwave liquid crystal technology is described starting from fundamental device concepts, to material properties and characterization. After the introduction of a use case, high-performance LC-based phase shifters for antenna systems are presented for 100 GHz communication systems. Millimeter wave and submillimeter wave communication systems are a very hot topic in nowadays research. As we will see, high-performance liquid crystal devices can play a Version August 24, 2018 submitted to Crystals 2 of 26 crucial role.

Bias on

Bias off

Polarization grid & bias electrodes

LC

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Polarization grid & bias electrodes

Light on

Light off

Light source

Bias on

Bias off

Figure 1. Working principle TN-cellused used in in LCDs. LCDs. (Right) unbiased LCLC rotates polarization Figure 1. Working principle ofofa aTN-cell (Right)the the unbiased rotates polarization ◦ . The light can pass two twisted polarization grids and the pixel is on. When the LC is biased by 90 ◦ by 90 . The light can pass two twisted polarization grids Polarization grid and the pixel is on. When the LC is biased (Left), the polarization of light is not rotated. Hence, the pixel is off as the light cannot pass the second bias electrodes (Left), the polarization of light is not rotated. &Hence, the pixel is off as the light cannot pass the second polarization grid. polarization grid. Solid Anisotropic * Isotropic **

Nematic Anisotropic

LC

2. How to Use LC at Microwave Frequencies?

Liquid Isotropic

The mixtures used for microwave applications are thermotropic calamitic nematic LCs. The phase Polarization grid bias electrodes transitions occur at certain temperatures, &where the liquid crystalline mesophase is defined thermodynamically stable between the melting point Light sourceof the crystalline phase and the clearing point ε to the isotropicε∥liquid phase. In this mesophase, liquid crystals show an orientational order but lack εL Tunable Range translational periodicity. In comparison to the smectic phase, the elongated molecules tend to align Figureε⊥1. Working principle of a TN-cell used in LCDs. (Right) the unbiased LC rotates polarization parallel tobyeach other but are not organized into layers. The fundamental properties of nematic LCs 90◦ . The light can pass two twisted polarization grids and the* pixel powdersis on. When the LC is biased ϑ ** single-crystals Melting Clearing versus temperature are shown in Figure 2. LC exhibits an anisotropy in the crystalline well as in the (Left), the polarization ofpoint light is not rotated.point Hence, the pixel is off as the light cannot pass theassecond nematic phase, whereas only the nematic phase can be used for electrically tunable RF circuits/devices. polarization grid. Figure 2. Overview on different phases of nematic liquid crystals. (Left) shows the material’s crystalline Solid Nematic for differentLiquid structure and the resulting permittivity orientations. In the (Right) picture the different Isotropic Anisotropic Anisotropic * Isotropic ** phases can be differentiated by the LC’s optical properties.

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from fundamental device concepts, to material properties and characterization. After the introduction of a use case, high-performance LC-based phase shifters for antenna systems are presented for 100 GHz ε communication systems. Millimeter wave and submillimeter wave communication systems are a very ε∥ εL hot topic in nowadays research. Tunable As weRange will see, high-performance liquid crystal devices can play a ε ⊥ crucial role.

33

Melting Clearing 2. How to use LC at Microwave point Frequencies? point

28 29 30 31

34 35 36 37 38 28 39 29 40 30 41 31 32

ϑ

* powders ** single-crystals

The mixtures used for microwave applications are thermotropic calamitic nematic LCs. The Figure 2. Overview on different phasesofofnematic nematicliquid liquid crystals. thethe material’s crystalline Figure 2. Overview on different phases crystals.(Left) (Left)shows shows material’s crystalline phase transitions occur at certain temperatures, where the liquid crystalline mesophase is defined structure and the resulting permittivity for different orientations. In the (Right) picture the different structure and the resulting permittivity for different orientations. In the (Right) the different thermodynamically stable between the melting point of the crystalline phase and picture the clearing point to phases candifferentiated be differentiated by the LC’soptical opticalproperties. properties. phases can be by the LC’s the isotropic liquid phase. In this mesophase, the nematic mixtures show an orientational order but lack translational periodicity. In comparison to smectic mixtures, the elongated molecules tend to align from fundamental device concepts, to material properties andfundamental characterization. After the introduction parallel to each other but are not organized into layers. The properties of nematic LCs of a use case, high-performance LC-based phase shifters for antenna systems are presented for as 100inGHz versus temperature are shown in Figure 2. LC exhibits an anisotropy in the crystalline as well the communication systems.only Millimeter wave and submillimeter communication systems are a very nematic phase, whereas the nematic phase can be used forwave electrically tunable RF circuits/devices. hot topic in nowadays research. As we will see, high-performance liquid crystal devices can play a crucial role.

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The molecules can be depicted with a rod-like shape where their uniaxial anisotropy is used to realize tunable devices. Due to the molecule’s rotational symmetry, the permittivity tensor Equation (1) reduces to     εk 0 0 ε uu 0 0     ε˜ =  0 ε vv (2) 0  · ε0 =  0 εk 0  · ε0 0 0 ε ww 0 0 ε⊥

with the (uvw)-coordinate system aligned with the director ~n of the LC volume, according to Figure 3. The director is a macroscopic unit vector giving the main direction of the molecules inside a unit volume element. The values of the permittivities ε ⊥ and ε k , and therefore, the LC’s anisotropy n n=

√


ε⊥ −

p

(3)

εk

3 of 26

are material and temperature dependent, refer to Section 3.

e u ε|| e v ε⟂

εm||

εm⟂ ex ey

LC unit volume

LC director n

LC molecule

Figure 3. Definition of the director ~n for a single LC unit volume element in the device’s coordinate system (xyz) and relation to the permittivities defined in the material’s coordinate system (uvw) with the material properties ε mk , ε m⊥ and the macroscopic unit volume properties ε k , ε ⊥ .

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The molecules electric displacement field D interacts withshape an electric field E according to The can be depicted with a rod-like where their uniaxial anisotropy is used to realize tunable devices. Due to the molecule’s rotational symmetry, the permittivity tensor Equation 1 D = ε˜ · E. (4) reduces to     εk 0 0 ε uu 0 0   side and  the permittivity  As displacement and electric fields on one tensor on the other side ε˜ =  0 ε vv (2) 0  · ε0 =  0 εk 0  · ε0 are defined in different coordinate systems, effective displacement must consider the rotation of the 0 0 ε ww 0 0 ε⊥ coordinate systems by introducing the rotation matrix R. with the (uvw)-coordinate system aligned with the director ~n of the LC volume, according to Figure 3. = R˜εthe R−1main · E. direction of the molecules inside a unit (5) The director is a macroscopic unit vector D giving volume element. The values of the permittivities ε ⊥ and ε k , and therefore, the LC’s anisotropy n The degree of freedom to realize tunable devices is now this rotation matrix R, leading to a tunable √ p effective permittivity (3) r n = ε⊥ − εk i2 h ε eff = [ε ⊥refer · sin to (6) (Θsection )]2 + ε3.k · cos (Θ) , are material and temperature dependent,

electric fieldcoordinate D interacts with anFor electric field when E according with The the tilt angledisplacement Θ between both systems. example, E = E0to·~ex , a continuously tunable effective permittivity ε eff can be achieved as D = ε˜ · E. (4)  ε k for ~eu k ~ex  As displacement and electric fields on  one side and the permittivity tensor on the other side are defined (7) ε ⊥ displacement for ~eu k ~must ey ∨~euconsider k ~ez , the rotation of the coordinate eff = in different coordinate systems, εthe effective    systems by introducing the rotation matrix ε . .R. .ε else ⊥

k

D = R˜εR−1 · E.

(5)

The degree of freedom to realize tunable devices is now this rotation matrix R, leading to a tunable effective permittivity r h i2 ε eff = [ε ⊥ · sin (Θ)]2 + ε k · cos (Θ) , (6)

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This example can be visualized Version August 24, 2018 submitted to Crystals

with a simple parallel-plate capacitor. The LC material is 4 of 26 sandwiched between two electrodes and an electric field is applied as shown in Figure 4.

eu

eu ev

ev

E=E0 ex

E=E0 ex

eu ev

E=E0 ex

ex ey

εeff = ε ≈ εm

εm⟂ < εeff < εm

εeff = ε⟂ ≈ εm⟂

Figure 4. Idealized parallel-plate capacitor with LC as dielectric filling for the three cases defined in Equation (7), (Left)~e~ , (Right) , and(Center) (Center)“else”. “else”.Note: Note:Interface Interfaceeffects effects between between LC and ~eu~euk k~ey~e,yand Equation 7, (Left) ueukk~e~ xe,x (Right) electrodes are neglected for simplification.

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The LC material in tunable devices is described by the continuum theory [3]. The system’s energy This example can be visualized with a simple parallel-plate capacitor. The LC material is sandwiched is defined based on the Gibbs Free Energy, which is affected from the elastic deformations, external between two electrodes and an electric field is applied as shown in Figure 4. electric and magnetic fields as well as anchoring effects. Thus, an increase in the Gibbs Free Energy is The LC material in tunable devices is described by the continuum theory [3]. The system’s energy minimized by changing the orientation of the director to reach equilibrium. is defined based on the Gibbs Free Energy, which is affected from the elastic deformations, external ZZZ ZZZ electric and magnetic fields as well as anchoring effects. Thus, an increase in the Gibbs Free Energy is Wf = df = d f elastic + d f electric + d f magnetic + d f surface (8) minimized by changing the orientation of the director to reach equilibrium. V

ZZZ

45 46

47 48 49

V

ZZZ

W different df = d f elastic + d f electric + dpractical f magnetic implementation + d f surface (8) From this, three methods can be derived for the of alignment f = of LCs. V V In the presence of an external electric field, the energy From this, three different methods can be derived for the practical implementation of alignment iD E of LCs. 1 h d f electric = − ε 0 ε r,⊥ + ∆ε r (cos Θ)2 E2 . (9) In the presence of an external electric2 field, the energy

i D E energy, if a certain threshold is causing a re-orientation of the director ~1n tohachieve minimum2system d f electric = − ε 0 ε r,⊥ + ∆ε r (cos Θ) E2 . (9) energy is overcome. It should be noted that 2 ∆ε is positive for millimeter wave optimized nematic LCs. Therefore, the director tends to align in a way that is parallel to the electric field lines. is causing a re-orientation of the director ~n to achieve minimum system energy, if a certain threshold In principle, the same is valid for magnetic fields, where energy is defined as energy is overcome. It should be noted that ∆ε is positive for millimeter wave optimized nematic LCs. h the electric field lines. iD E Therefore, the director tends to align parallel 1 to d f magnetic = − µ0 µr,⊥ + ∆µr (cos Θ)2 H2 (10) In principle, the same is valid for magnetic fields, where energy is defined as 2 i D4) can E also be used to align LC At last, surface forces (which have 1beenh neglected in Figure d f magnetic = − µ0 µr,⊥ + ∆µr (cos Θ)2 H2 (10) molecules. Any physical impurities on the 2 boundaries of the cavity result in interaction between the boundary and the LC molecules. According to the Rapini–Papoular model [4–6], this interaction At last, surface forces (which have been neglected in Figure 4) can also be used to align LC energy density is given as molecules. Any physical impurities on the boundaries of the cavity result in interaction between the boundary and the LC molecules. According to the Rapini-Papoular model [4–6],i this interaction energy 1h d f surface = Wp (sin (Θ − Θ0 ))2 + Wa (sin (Φ − Φ0 ))2 (11) density is given as 2

50 51 52 53 54 55 56 57 58

i with the director and the preferred and azimuthal angles specified by (Θ, Φ) 1 halignment for the polar d f surface = Wp (sin (Θ − Θ0 ))2 + Wa (sin (Φ − Φ0 ))2 (11) and (Θ0 , Φ0 ). The energy depends2 on the anchoring strength Wa and Wp , which are LC material and anchoring surface-dependent parameters. with the director and the preferred alignment for the polar and azimuthal angles specified by (Θ, Φ) In practical applications, different orientation mechanisms are combined according to Table 1. and (Θ0 , Φ0 ). The energy depends on the anchoring strength Wa and Wp , which are LC material and For example, the combination of surface anchoring and electrical biasing concepts as it is used in the anchoring surface dependent parameters. LCD technology, compare Figure 5a. Without the electric field, the directors are aligned in state I In practical applications, different orientation mechanisms are combined according to Table 1. For and with applied field they change their orientation towards state II, at which the numbering of the example, the combination of surface anchoring and electrical biasing concepts as it is used in the LCD states has been arbitrarily chosen. This mechanism is preferred in planar devices. One reason for the technology, compare Figure 5a. Without the electric field, the directors are aligned in state I and with limitation to planar devices is that the forces of the surface anchoring are weak compared to electric applied field they change their orientation towards state II, at which the numbering of the states has been arbitrarily chosen. This mechanism is preferred in planar devices. One reason for the limitation to planar devices is, that the forces of the surface anchoring are weak compared to electric forces. Hence,

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forces. Hence, the tuning speed from on- to off-state, which are dependent on the sheet thickness, are much lower than those from off- to on-state.

Forces used Forces used for for state I I state

Table 1. Summary of combination of different alignment mechanism and their application to tunable Table 1. Summary of Note combination different alignment mechanism and their application to tunable microwave devices. that the of numbering of the states has been arbitrarily chosen. “Not tunable” microwave devices. Note thatelectrical the numbering of the states has been arbitrarily chosen. “Not tunable” denotes combinations where tuning cannot be achieved. E.g., a surface-surface combination denotes combinations where electrical tuning be combinations, achieved. E.g.,which a surface-surface will result in a static configuration. “Not used”cannot denotes can be usedcombination in principle will result in a static configuration. “Not used” denotes combinations, which can be used in principle but are not used in practice. but are not used in practice. Magnetic Magnetic Not used Not used

Surface Surface Electric Electric Magnetic Magnetic

Not used Not used Characterization only Characterization only

Forces used for state II Forces used for state II Electric Electric Planar devices Planar devices Waveguide devices Waveguide devices Not used Not used

Surface Surface Not tunable Not tunable Planar devices Planar devices Not used Not used

For volumetric waveguide-based components, solely electrical biasing concepts (compare For5b) volumetric waveguide-based components, solely electrical biasing concepts (compare Figure are preferred, e.g., by applying two electrodes in perpendicular spacial directions. Figure 5b) areonpreferred, e.g., by applying two electrodes in perpendicular By switching one pair of electrodes, the directors are aligned for state I. By spacial togglingdirections. the other By switching on one pair of electrodes, the directors are aligned for state I. By toggling the other pair of electrodes, the directors will orient into state II. By proper superposition of both fields, any pair of electrodes, directors willcan orient into state II. ByItproper superposition ofaboth fields, any alignment betweenthe these two states be reached as well. is also possible to rotate pair of magnets alignment these two states can be reached as well. It is also possible to rotate athis pairapproach of magnets around thebetween structure to realize a variable alignment, compare Figure 5c. Obviously, is around theinstructure realize a variableand alignment, compare Figure 5c. Obviously, approach is not smart terms of to space requirements power consumption. Hence, this methodthis is only applied not smart in terms of space requirements and power consumption. method for material characterization and for a first proof-of-concept in theHence, lab, seethis Section 3. is only applied for material characterization and for a first proof-of-concept in the lab, see Section 3.

LC LC LC volume volume volume with with with director director director

⚓ ⚓ ⚓

⚓ ⚓ ⚓

Electrodes Electrodes Electrodes (off) (off) (off) Electrodes Electrodes Electrodes (biased) (biased) (biased)

ERF ERF E =E =E e0xe0xex RF 0 =E


⚓ ⚓ ⚓

⚓ ⚓ ⚓

⚓ ⚓ ⚓

⚓ ⚓ ⚓

⚓ ⚓ ⚓

⚓ ⚓ ⚓

εεeff ε eff → eff → ε

NNN S SMagnets SMagnets Magnets

NNN

HH =H =H eByeByey DCH DC DC B=H

EDC EDC E=E =E eBxeBxex DC B=E

HH =H =H eBxeBxex DCH DC DC B=H



(a)

SSS

NNN

EDC EDC E=E =E eBxeBxex DC B=E

⚓ ⚓ ⚓

eyeyey



⚓ ⚓ ⚓

exexex

⚓ ⚓ ⚓ EDC EDC E=E =E eByeByey DC B=E

State II State II State ε →IIε

State II State State ε → εI

εεeff ε⟂⟂ eff → eff → ε⟂

Layer Layer for for for surface surface surface anchoring anchoring anchoring ⚓ ⚓ ⚓Layer

⚓ ⚓ ⚓

SSS

(b)

(c)

Figure 5. 5. Possible Possible combinations combinations of of different different orientation orientation mechanisms mechanisms with with state state II and and II II at at top top and and Figure bottom, respectively: (Left) combination of surface anchoring and electric actuation as used for planar bottom, respectively: (Left) combination of surface anchoring and electric actuation as used for planar devices; (Center) (Center) both both electric electric actuations actuations as as used used in in waveguide waveguide components components and and (Right) (Right) both both magnetic magnetic devices; actuations as used for material characterization. actuations as used for material characterization.

At this point it is clear, how the alignment of the director can be technically realized. For application in tunable components, the RF signal and the control (bias) must be discriminated. For magnetic alignment, the bias field and the RF signal’s E-field are decorrelated by definition. This is different for both other cases, where the electric field is used to reach at least one state. The only

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At this point, it is clear how the alignment of the director can be technically realized. For application in tunable components, the RF signal and the control (bias) must be discriminated. For magnetic alignment, the bias field and the RF signal’s E-field are decorrelated by definition. This is different for both other cases, where the electric field is used to reach at least one state. The only difference in RF signal and bias E-field is the frequency. The minimization of overall system’s energy according to Equation (8) must hold also in this case. The temporal behavior of the LC volume is encapsulated in the elastic term as this includes the mechanical materials properties. Due to the viscosity, the director cannot follow an electric field with fast changing amplitude. Additionally, the dielectric contrast required for LC steering is much higher at DC than at RF frequencies. Hence, the influence of the RF field can be neglected. This is also relevant for the linearity of the components. The electric tuning of the material leads automatically to a non-linear behavior of the components, as the RF field will also change the state of LC. There are just few measurements of LC’s linearity published. In 2006, IP3 measurements of two different nematic liquid crystals have been published in [7]. Measurements of a phase shifter showed that the IP3 of this device is in the range of 60 dBm. Since then, no other measurements have been published. The excellent large signal characteristics obtained in this measurement already fulfill the requirements of many RF applications. Furthermore, the achieved performance makes linearity measurements of LC very difficult, as standard measurement equipment is at its boundaries. Hence principles for measuring passive intermodulation (PIM) must be adapted for LC non-linear characterizations. This specific topic will gain more interest in future as commercialization advances. Up to now, no devices showed significant non-linearity. After the fundamental device concepts have been discussed, the material must be characterized before we will show implementation advances around 100 GHz. For material characterization different realizations have been discussed recently. In the following we will show the most important ones. 3. Material Properties and Characterization The state-of-the-art setup for LC characterization at 19 GHz and 30 GHz is a precise resonator-based setup, making use of the cavity perturbation technique. In [8] the resonator based dual mode characterization technique was introduced, where a resonator is measured with and without LC inside a sample holder made of silica, see Figure 6a. Due to the dielectric properties of LC, the field distribution, and therefore, the resonance frequency of the excited mode will shift compared to the reference measurement. By this, the permittivity as well as the loss tangent of the material can be determined. Although this kind of measurement is limited to one single frequency, the dielectric properties of the LC can be determined much more precisely than with broadband measurement setups. A detailed mathematical description of this method can be found in [8]. By using a dual-mode cylindrical resonator, the permittivity and dielectric loss of LC can be measured for both parallel and perpendicular orientation in one single step, see Figure 6b. For this, two perpendicularly aligned modes are excited. The terms “parallel” and “perpendicular” define the two extreme cases of orientations of the LC molecules, where the long axes of the molecules are aligned parallel or perpendicular with respect to the applied RF field, respectively. The silica tube is placed in the center of the resonator to ensure maximum field interaction, and therefore, high sensitivity. The resonator is excited by rectangular waveguides being weakly coupled to the resonator by means of a small coupling iris, see Figure 6c. During the characterization, the LC’s long axes are aligned parallel to the silica tube with the help of permanent magnets.

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Insertion Insertion hole hole for for silica silica tube tube Silica Silica tube tube

LC LC Resonator Resonator

Coupling Coupling iris iris TM TM010 010 010

TE TE111 111 111

Resonator Resonator

(a) Basic setup

(b) Used RF modes

(c) Complete resonator

Figure6.6.Resonator Resonatorsetup setupfor fordual dualmode modeLC LCcharacterization characterizationbased basedon onthe thecavity cavityperturbation perturbationtechnique technique Figure at19 19GHz. GHz. at

Withthis thischaracterization characterizationmethod, method,the theLC LCmixtures mixturescould couldbe beimproved improvedin interms termsof ofloss losstangent tangentas as With wellas asrelative relativematerial materialtunability tunability well −εε⊥⊥ εε k− = k (9) ττLC ,, (12) LC = εεkk

ascan canbe beseen seenin inFigure Figure7. 7. The Thefirst firsttunable tunablemicrowave microwavedevices devicesin inthe theearly early2000’s 2000’swere wererealized realizedby by as usingdisplay displayLCs LCssuch suchasasK15 K15 BL111. later, new mixtures were specifically synthesized for using oror BL111. AA bitbit later, new mixtures were specifically synthesized for the Version August 24, 2018 submitted to Crystals 8 of 26 the microwave range. The LC mixture used the components presented in this is the microwave range. The LC mixture being being used for the for components presented in this paper is paper the mixture mixture GT5-26001 from Merck KGaA with a permittivity between 2.39 < εr < 3.27 GT5-26001 from Merck KGaA, Darmstadt, Germany with a permittivity between 2.39 < and ε r < dielectric 3.27 and losses of 0.007 δ >>0.0022. comparison of different LC mixtures is summarized in Tablein2. dielectric losses> oftan 0.007 tan δ >A0.0022. A comparison of different LC mixtures is summarized A review recent development can be found [9]. in [9]. Table 2. A on review onmaterial recent material development can beinfound Table 2. Material properties of different LC mixtures at room temperature.

0.03

LC Mixture εr,k

E7 K152.98 BL111 3.25 0.02 CLHS-1 3.2 GT3-23001 3.28 GT5-26001 3.27 GT5-28004 3.32

Material loss tan δmax

GT-series LCs

Display LCs tan δk

0.009 0.0084 0.0033 0.0038 0.0022 0.0014

19 GHz εr,⊥ tan δ⊥ 2.53 2.64 2.46 2.46 2.39 2.4

τε (%)

0.022 BL111 15 0.0218 18.77 0.0131 23.1 0.0143 25.2 0.007 27 0.0043 27.7

Experimental LCs 1 kHz εr,k εr,⊥

TC (◦C)

γ1 (mPa s)

19.6 22.0

5.2 5.5

58 85

252 283

8.0 3.7 3.5

4.0 2.7 2.7

173.5 146 151

727 1958 5953

0.01

GT3-23001 One advantage of LC compared to other technologies such as semiconductors is a decreasing GT5-26001 dielectric loss with increasing frequency in the range above 15 GHz. Measurements up to 1.5 THz have been conducted, 0.00 using a time-domain spectroscopy (TDS) THz measurement system as shown in 20 25 30 10 15 Figure 8. Tunability τ (%) The beam generated by a Ti-sapphire laser is split εup in the beginning. While one part of the beam is used as probe beam, the other one is passing a delay station before hitting the THz emitter, e.g., Figure 7. Development from 2000’s mirrors where first of LC’s microwave 19 GHz GHz from the a biased GaAs-based photoconductive emitter.properties The THzatbeam is guided viaearly parabolic through display LCs have been used towards recent LC mixtures specifically synthesized for microwave the sample holder and afterwards to the ZnTe crystal for THz detection. The THz as well as the probe applications. The mixtures were measured room temperature. beams are propagating collinearly through at the crystal. There, the THz beam induces a birefringence, which can be read out by the linearly polarized probe beam. By this, the linear polarization is rotated by the THz beam. Together with a λ/4-wave plate, a beam splitting polarizer and a set of balanced photodiodes, the THz beam’s amplitude can be mapped by monitoring the rotation of the probe beam’s polarization at a variety of delay times. Therefore, the LC properties can be determined by

Ti-sapphire laser

Beam splitter Probe beam

ZnTe crystal

Balanced photodiode detector

Material loss tan δma

BL111

K15

0.02

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0.01

GT3-23001

Table 2. Material properties of different LC mixtures at room temperature. GT5-26001

0.00 LC Mixture 10

εr,k

19 GHz tan δ15 εr,⊥ tan δ⊥ 20τε (%) k

1 kHz εr,k 25 εr,⊥

TC (◦ C)

γ1 30s) (mPa

τ15 ε (%) 19.6 E7 2.98 0.009 2.53 Tunability 0.022 5.2 58 252 BL111 3.25 0.0084 2.64 0.0218 18.77 22.0 5.5 85 283 FigureCLHS-1 7. Development3.2of LC’s microwave at 19 GHz from the early 2000’s where first 0.0033 2.46 properties 0.0131 23.1 3.28 25.2specifically 8.0 4.0 173.5 for 727 displayGT3-23001 LCs have been used0.0038 towards 2.46 recent 0.0143 LC mixtures synthesized microwave GT5-26001 3.27 0.0022 2.39 0.007 27 3.7 2.7 146 1958 applications. The mixtures were measured at room temperature. GT5-28004 3.32 0.0014 2.4 0.0043 27.7 3.5 2.7 151 5953 One advantage of LC compared to other technologies such as semiconductors is a decreasing dielectric loss with increasing frequency in the range above 15 GHz. Measurements up to 1.5 THz have been conducted, using a time-domain spectroscopy (TDS) THz measurement system as shown in Figure 8. Ti-sapphire laser

Beam splitter ZnTe crystal

Probe beam

Mirror

Balanced photodiode detector

Wollaston prism Sample holder

Delay stage

THz emitter

THz beam Parabolic mirror

Figure 8. Schematic of a time-domain spectroscopy THz measurement system as used in [10] for LC characterization. By comparing the measurement results of an empty and a LC-filled sample holder, the dielectric properties of LC can be determined.

The beam generated by a Ti-sapphire laser is split up in the beginning. While one part of the beam is used as probe beam, the other one is passing a delay station before hitting the THz emitter, e.g., a biased GaAs-based photoconductive emitter. The THz beam is guided via parabolic mirrors through the sample holder and afterwards to the ZnTe crystal for THz detection. The THz as well as the probe beams are propagating collinearly through the crystal. There, the THz beam induces a birefringence, which can be read out by the linearly polarized probe beam. By this, the linear polarization is rotated by the THz beam. Together with a λ/4-wave plate, a beam splitting polarizer and a set of balanced photodiodes, the THz beam’s amplitude can be mapped by monitoring the rotation of the probe beam’s polarization at a variety of delay times. Therefore, the LC properties can be determined by a differential measurement method, comparable to the resonator-based characterization. Moreover, it is a highly interesting feature of THz TDS, to be able to get full information of the electric field, i.e., amplitude and delay. One of the characterized LCs was the mixture GT3-23001, which shows an almost constant permittivity over a wide frequency range between 200 GHz to 1.5 THz, see Figure 9. The permittivities of ε ⊥ = 2.34 and ε k = 3.19 fit well (5.1% deviation for the perpendicular state and 2.8 % for the

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parallel state) with the values measured with the previously mentioned resonator setup at 19 GHz. The dielectric losses are in the expected order of magnitude, but could not be determined precisely, due to August the low range the TDS system. Further information and a more detailed description Version 24,dynamic 2018 submitted to of Crystals 9 of 26 can be found in [11].

State I ε ⊥ State II ε k

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9. THz characterization results for the LC GT3-23001 Merck KGaA. Germany. Figure Figure 9. THz characterization results for the LC GT3-23001 from Merckfrom KGaA, Darmstadt,

127 128 129 130 131 132 133 134 135 136 137 138 139 140 141

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Therefore, LC Together has been proven well suitable the realization ofand continuously tunable by the THz beam. with a as λ/4-wave plate,material a beamfor splitting polarizer a set of balanced components not the microwavebutcan also the (sub-)millimeter wave THz range. photodiodes, theonly THzinbeam’s amplitude beinmapped by monitoring theand rotation of the probe beam’s polarization at a variety of delay times. Therefore, the LC properties can be determined by a 4. Use Cases for LC-based Microwave Devices differential measurement method, comparable to the resonator-based characterization. Moreover, it Modern communication systems are to onbe the way generation The paradigm is a highly interesting feature of THz TDS, able to to getthe full5th information of (5G). the electric field, i. e. change includes three different main directions including increased data rate up to several gigabit per amplitude and delay. second, communication downthe to mixture milliseconds, and finally, increased densityconstant [12–15]. Onelow-latency of the characterized LCs was GT3-23001, which shows user an almost While most of the functionality will be covered by improved network architecture including e.g., permittivity over a wide frequency range between 200 GHz to 1.5 THz, see Figure 9. The permittivities network slicing, new deployed higher frequencies. Asand the ITU to of ε ⊥ = 2.34 andthere ε k =are 3.19 fit systems well (5.1to % be deviation foratthe perpendicular state 2.8 %plans for the standardize a new 60 GHz band at WRC’19, first providers demonstrated the potential at international parallel state) with the values measured with the previously mentioned resonator setup at 19 GHz. sportdielectric events such as are thein last Olympiad South Korea [16,17]. (mmWave) The losses thewinter expected order ofinmagnitude, but could Millimeter-wave not be determined precisely, communication at 60 GHz andofhigher frequencies is mainly limited by high free-space path loss and due to the low dynamic range the TDS system. Further information and a more detailed description atmospheric impairments, compare Figure 10. can be found in [11]. A second LC trend in mobile communications is characterized by worldwide ubiquitous Therefore, has been proven as well suitable material for the realization of continuously tunable communication for a long time but brought to the real world by and projects Google’s components not envisioned only in the microwavebut also in the (sub-)millimeter wave THz like range. LOOM and OneWeb. These concepts summarized in the New Space Initiative use flying platforms 4. Use for or LClow based Microwave Devices such asCases balloons earth orbit (LEO) satellites to provide communication services. Independently fromModern the finalcommunication system concepts, high-gain antennas required to establish a stable communication systems are on the wayare to the 5th generation (5G). The paradigm change link fromthree users’ terminals the network. Forincreased systems operated at to Kaseveral (26.5 GHz to 40 or includes different maintodirections including data rate up gigabit perGHz) second, K (12 GHz to 18 GHz) bands, high-gain antennas are required to bridge large distances between u low-latency communication down to milliseconds, and finally, increased user density [12–15]. While terminals, 600 km to will the LEO satellite, corresponds approximately 180 dBe.of loss most of thee.g., functionality be covered bywhich improved networktoarchitecture including g. path network at Ka -band. 60 GHz system free space path loss is alreadyAs reached a distance of 400 km. slicing, thereFor are anew systems to bethis deployed at higher frequencies. the ITUatplans to standardize a In practice, the reach is much lower due to mentioned atmospheric impairments such as absorption by new 60 GHz band at WRC’19, first providers demonstrated the potential at international sport events oxygen ≈ 14.6 and rain attenuation (≈ 5.8 dB/km for medium rain) for 60 GHz transmission, such as (the last dB/km) winter Olympiad in South Korea [16,17]. Millimeter-wave (mmWave) communication according ITU recommendations standard atmosphere at 60 GHz to and higher frequencies isfor mainly limited by high[18,19]. free-space path loss and atmospheric

impairments, compare Figure 10. A second trend in mobile communications is characterized by worldwide ubiquitous communication envisioned for long time but brought to the real world by projects like Google’s LOOM and OneWeb. These concepts summarized in the New Space Initiative use flying platforms such as balloons or low earth orbit (LEO) satellites to provide communication services. Independently from the final system concepts, high-gain antennas are required to establish a stable communication link from users’ terminals to the network. For systems operated at Ka (26.5 GHz to 40 GHz) or Ku (12 GHz to 18 GHz) bands, high-gain antennas are required to bridge large distances between terminals, e. g. 600 km to the LEO satellite, which corresponds to approximately 180 dB of path loss at Ka -band. For a 60 GHz system this free space path loss is already reached at a distance of 400 km. In practice, the reach is much lower due to mentioned atmospheric impairments such as absorption by

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Figure 10. 10. Channel Channel loss loss for for different different frequencies frequencies as as sum sum of of free free space space path path loss loss (dashed (dashed lines) lines) and and Figure the atmospheric atmospheric attenuation attenuation according according to to ITU ITU standard standard atmosphere atmosphere (solid (solid lines) lines) for for homogenous homogenous the medium between between transmitter transmitter and and receiver. receiver. As As reference reference the the properties properties of of aa geostationary geostationary satellite satellite for for medium broadcasting at Ku -band (12 GHz) 000 km 210 dB of210 insertion broadcasting GHz) with with 36 36,000 kmtransmission transmissionrange rangewith withapprox. approximately dB of loss are highlighted. insertion loss are highlighted.

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As the transmitter power is limited, thedB/km path loss be compensated by the gain of the oxygen (≈ 14.6 dB/km output and rain attenuation (≈ 5.8 formust medium rain) for 60 GHz transmission, antenna. gainrecommendations g of an antenna and beam width are connected accordingThe to ITU forthe standard atmosphere [18,19]. by As the transmitter output power is limited, the path4π loss must be compensated by the gain of the (13) g= η ·the D≈ η · width are connected by antenna. The gain g of an antenna and beam HPBWaz · HPBWel

4πwidth HPBW and HPBW in azimuth with the antenna’s efficiency η and half-power beam az el g =the η·D ≈η· (13) HPBW · HPBW el and elevation, respectively. Hence, to realize an antennaazwith enough gain (g > 60 dB), the width of the gets withefficiency values ofηless 0.1◦ very narrow. If youHPBW now want to establish a connection, withbeam the antenna’s andthan the half-power beam width az and HPBWel in azimuth and the antennas must be aligned in a precise way. elevation, respectively. Hence, to realize an antenna with enough gain (g > 60 dB), the width of the For thewith alignment be used. Among the very common ones are beam gets valuesof ofantennas, less than different 0.1◦ verymethods narrow. can If you now want to establish a connection, the mechanically steered antennas, where the antenna’s axes are motorized. This results in heavy systems antennas must be aligned in a precise way. whichFor additionally require maintenance Beside electronically steerable have the alignment of high antennas differentefforts. methods canthis, be used. Very common areantennas mechanically been developed andwhere explored the last centuries. These are based the array concept where which many steered antennas, the in antenna’s axes are motorized. Thison results in heavy systems antenna elements are combined with respect to phase and amplitude to synthesize a steered antenna additionally require high maintenance efforts. Beside this, electronically steerable antennas have beam pattern. For high-frequency high-bandwidth applications as on indicated byconcept previous scenarios, been developed and explored in the last centuries. These are based the array where many analog beam-steering concepts are a low-cost alternative to fully digital systems. Figure 11 gives an antenna elements are combined with respect to phase and amplitude to synthesize a steered antenna overview on the most important implementations. beam pattern. For high-frequency high-bandwidth applications as indicated by previous scenarios, LCsbeam-steering are the mostconcepts promising for applications GHz as their11dielectric analog are acandidates low-cost alternative to fully above digital 15 systems. Figure gives an loss is superior compared to other technologies such as ferroelectrics [20–25] and ferrites [26–28]. overview on the most important implementations. A direct competitor are micro-electro-mechanical systems (MEMS) [29,30], which utilize mechanical LCs are most promising candidates for applications above 15 GHz as their dielectric loss is changes the micro scale technologies to implementsuch phase shifter components for [26–28]. array antennas. superioron compared to other as ferroelectrics [20–25] required and ferrites A direct Researchers around the world develop new materials which can be used for tunable devices.changes Recent competitor are micro-electro-mechanical systems (MEMS) [29,30], which utilize mechanical examples are e.g., phase change materials [31,32] and electrochromic materials [33]. These have a great on the micro scale to implement phase shifter components required for array antennas. Researchers potential, and should be monitored consciously. around the world develop new materials which can be used for tunable devices. Recent examples are e. g. phase change materials [31,32] and electrochromic materials [33]. These have a great potential, and should be monitored consciously. To describe the requirements and properties of LC based array antennas in more detail, the use case of a Ka -band LEO satellite system is discussed. As the satellite is non-stationary, the antennas

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Possible implementation implementation of of the the weighting weighting functions functions w wmn for array array antennas. antennas. The The block block Figure 11. 11. Possible mn for Figure diagrams are are simplified simplified in in such such aa way way that that all all components components and and stages stages not not essential essential for for signal signal weighting weighting diagrams are neglected. neglected. are

describe the requirements properties of LC-based antennas in more use mustTo track the satellite all time. Theand orbit time of such a satellite array is around 120 min, so thedetail, phase the shifters ◦ case of a K -band LEO satellite system is discussed. As the satellite is non-stationary, the antennas a have to provide tuning speed of more than roughly 70 /min [34]. Hence, realization technology and must track the satellite all time. time of such a satellite 120length min, so material properties must matchThe theorbit requirements. The change is inaround electrical ofthe thephase phaseshifters shifter ◦ have to provide to tuning speed of more than roughly 70 of /min realization is proportional the dielectric contrast (anisotropy) the [34]. usedHence, LC mixture. The technology phase shift and of a material properties must match the requirements. The change in electrical length of the phase shifter transmission line phase shifter is defined by is proportional to the dielectric contrast (anisotropy) of the used LC mixture. The phase shift of a √ transmission line phase shifter is defined by ϕ = β · l ∝ ε · l, (14)

√ ϕ = β ·phase l ∝ εshift · l, is defined as with the physical line length l. The< differential

p √ with the physical line length l. The ϕ⊥ = β k phase − β ⊥ shift · l ∝is defined ∆ϕ = ϕk −differential ε k − as ε⊥ · l

183 184

(14)

(15) p √ β ⊥ · l ∝ lossε kbut − also = ϕanisotropy, ε ⊥ high · l viscosity are targeted. (15) Hence LC mixtures with∆ϕ high low k − ϕ⊥ = β k − dielectric Beside the material properties, the technological implementation is also challenging, as a hermetic Hence, LC mixtures with high anisotropy, lowfrom dielectric loss One but also high viscosity are targeted. liquid-proof must be realized to prevent the LC leaking. example for the realization of Beside the material properties, the technological implementation is also challenging, as a hermetic fully integrated planar LC based array antennas has been published in [35]. Here, the phase shifters liquid-proof mustas bemeandered realized to microstrip prevent the LCphase from shifters leaking.directly One example realization of are implemented line coupledfor to the an array of patch fully integrated planar array antennas has been published inthe [35]. Here, theofphase shifters antennas, see Figure 12. LC-based While planar implementations are well suited at lower end the frequency are implemented as meandered microstrip line phase shifters directly coupled to an array of patch band, ohmic losses of the microstrip line topology increase with frequency. While the planar topology antennas, see Figure 12. While planar implementations are well suited at the lower end of the frequency is meanwhile followed by the company Alcan Systems, research is focusing on other implementations band, ohmic losses of the microstrip line topology increase frequency. While planar topology such as high-performance rectangular waveguide basedwith implementations. Tothe compare different is meanwhile followed by the company Alcan Systems Darmstadt, Germany, research is focusing implementations the phase shifter Figure of Merit (FoM) is defined as on other implementations such as high-performance rectangular waveguide based implementations. ∆Φ To compare different implementations the phase FoM =shifter Figure of Merit (FoM) is defined as (16) max IL ∆Φ FoMphase = shift ∆Φ divided by the highest insertion loss(16) which is the ratio of the maximally achievable for max IL all tuning states max IL. which is the ratio of the maximally achievable phase shift ∆Φ divided by the highest insertion loss for all tuning states max IL.

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Figure 12. Planar Antenna with LC enabled beam steering. (Left) early demonstrator of an 2 × 2 array published in [35] with (top left) the measured beam pattern at 17.5 GHz with a antenna gain of 6 dB realized by phase shifters with a FoM of 90◦ /dB. (Right) prototype of an 8 × 8 sub-array realized in thin film LC technology. The complete antenna consists of 16 panels and shows a gain of 30 dB with a steering range of ±55◦ .

5. Metallic Rectangular Waveguides

Figure 12. Planar Antenna with LC enabled beam steering. (Left) early demonstrator of an 2 × 2 array A very common approach at measured Ka -band beam frequencies around 30with GHz for thegain realization published in [35] with (top left) the pattern at 17.5 GHz a antenna of 6 dB of ◦ tunable LC-based high-performance RF components is the dielectrically filled metallic waveguide. realized by phase shifters with a FoM of 90 /dB. (Right) prototype of an 8 × 8 sub-array realized in The dielectric filling contains cavity, in whichconsists the tunable LC material is afilled. theacross thin film LC technology. Theacomplete antenna of 16 panels and shows gain ofHence, 30 dB with section of therange rectangular steering of ±55◦ . waveguide is (partially) filled with tunable LC, see Figure 13. Brass

Hostaphan

Electrode connection terminal

Rexolite

LC

Biasing electrodes

Figure

Figure 13. Schematic cross section of the metallic rectangular waveguide LC phase shifter with partial LC filling. Figure 13. Schematic cross section of the metallic rectangular waveguide LC phase shifter with partial LC filling. 185

186 187 188 189 190 191 192 193 194 195 196 197 198

5. Metallic Rectangular Waveguides By controlling its effective permittivity, the electrical length of the waveguide can be varied, A very common approach at Ka -band frequencies around 30 GHz for the realization of tunable LC resulting in a tunable delay line. In combination with the intrinsic low ohmic loss of the waveguide, based high-performance RF components is the dielectrically filled metallic waveguide. The dielectric this leads to an easy realization of phase shifters with a high figure-of-merit (FoM) of more than filling contains a cavity, in which the tunable LC material is filled. Hence, the cross section of the 120◦ /dB [36]. The LC orientation is controlled by means of six pairs of electrodes, which need to be rectangular waveguide is (partially) filled with tunable LC, see Figure 13. included into the waveguide, due to its electrical boundaries. A high complexity of these biasing By controlling its effective permittivity, the electrical length of the waveguide can be varied, electrodes is required for avoiding the propagation of parasitic stripline modes between the electrodes resulting in a tunable delay line. In combination with the intrinsic low ohmic loss of the waveguide, and the waveguide’s walls. An example for the parasitic modes as well as the biasing electrodes’ this leads to an easy realization of phase shifters with a high figure-of-merit (FoM) of more than 120 ◦ /dB [36]. The LC orientation is controlled by means of six pairs of electrodes, which need to be included into the waveguide, due to its electrical boundaries. A high complexity of these biasing electrodes is required for avoiding the propagation of parasitic stripline modes between the electrodes and the waveguide’s walls. An example for the parasitic modes as well as the biasing electrodes’ complexity is given in Figure 14.The electrodes were processed on a thin Hostaphan film of 50 µm thickness, being lead to the outside of the metallic waveguide for contacting to the voltage supply.

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complexity is given in Figure 14. The electrodes were processed on a thin Hostaphan film of 13 50ofµm 26 thickness, being lead to the outside of the metallic waveguide for contacting to the voltage supply.


Stripline mode with high field components max

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with low field components


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Figure 14. (Left) (Left) Field Field intensity intensity of of the the guided guided modes modes within the dielectrically filled waveguide phase shifter for the case of straight biasing electrodes. Most power is confined in the TEM stripline mode, mode while nearly no power is inStripline the fundamental TE10 waveguide mode. (Right) Biasing electrodes with with high field components stepped-impedance structures and λ/4-stub lines for max strip mode suppression. stub lines stepped impedance

Aʹ horn Such phase shifters are used in Alightweight antenna arrays for satellite communications. 0 LC Although the LC’s response time is comparatively high, it is already enough for specific applications, PTFE such asl the tracking of a low earth orbit (LEO) Brass satellite from a geostationary (GEO) satellite, where a ◦ steering speed of maximum 73 /min is required [11,34,37]. line width ~60 µm A The concept of a dielectrically filled rectangular waveguide can also be adapted to W-band Aʹ Waveguide mode frequencies around 100 GHz [38,39]. There, the waveguide was realized in split-block technology with low field components made of brass, where the bottom part has a u-shape, being sealed with a top lid, see Figure 15. The waveguide is tapered in width andguided heightmodes within the dielectrically filled section to still provide Figure 14. (Left) Field intensity of the within the dielectrically filled waveguide phase shifter for the case of straight electrodes. power of is confined in the TEM stripline a single mode propagation. Thebiasing tapering is doneMost by means λ/4 transformer steps. Themode, dielectric h while nearlyof noPTFE power(εisr in fundamental waveguide mode. (Right) Biasing with cavity is made = the 2.06 and tan δ TE =w10 0.000 222 [40]), being tapered in aelectrodes triangular shape. stepped-impedance structures and λ/4-stub lines for strip mode suppression. The PTFE cavity is also designed in a u-shape, which is sealed by pressing the metallic lid on top. Aʹbiased W-band rectangular waveguide phase shifter A Figure 15. Split-block design of the magnetically LC based on LC. The dimensions of the split-block are (l × w × h) = 31.6 mm × 30.0 mm × 33.0 mm.

l

199 200 201 202 203 204 205 206 207 208 209 210 211 199 212 200 213 201 214 202 215 203 216 204 217 205 218 206 207 208

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Such phase shifters are used in lightweightAhorn antenna arrays for satellite communications. Although the LC’s response time is comparatively high, itAʹis already enough for specific applications, such as the tracking of a low earth orbit (LEO) satellite from a geostationary (GEO) satellite, where a steering speed of maximum 73 ◦ /min is required [11,34,37]. The concept of a dielectrically filled rectangular waveguide can also be adapted to W-band h frequencies around 100 GHz [38,39]. There, the waveguide was realized in split-block technology w made of brass, where the bottom part has a u-shape, being sealed with a top lid, see Figure 15. The waveguide is tapered in width and height within the dielectrically filled section to still provide a single mode propagation. The tapering is done by means of λ/4 transformer steps. The dielectric cavity is Figure 15. 15. Split-block Split-block design design of of the the magnetically magnetically biased biased W-band W-band rectangular rectangular waveguide waveguide phase phase shifter shifter Figure made of PTFE (ε r = 2.06 and tan δ = 0.000 222 [40]), being tapered in a triangular shape. The PTFE based on LC. The dimensions of the split-block are ( l × w × h ) = 31.6 mm × 30.0 mm × 33.0 mm. based on LC. The dimensions of the split-block are (l × w × h) = 31.6 mm × 30.0 mm × 33.0 mm. cavity is also designed in a u-shape, which is sealed by pressing the metallic lid on top. After sealing, the LC is filled into the dielectric cavity through filling holes on the top part of the split-block. the help of a syringe the LC is flushed through thefor cavity, by which the amount Such phaseWith shifters are used in lightweight horn antenna arrays satellite communications. of air bubbles within the RF path can be significantly reduced, see Figure 16. The LC section within Although the LC’s response time is comparatively high, it is already enough for specific applications, the dielectric cavity of hasa alow width 1.0 mm a height of 0.6 mm and a length of 14.6 mm, providing such as the tracking earthoforbit (LEO) satellite from a geostationary (GEO) satellite, where a ◦ according ◦ /min is required phase shift of more than 36073 to the simulation steering speed of maximum [11,34,37]. carried out with CST Studio Suite. This demonstrator canof only be biased magnetically, since no biasing electrodes be included to this The concept a dielectrically filled rectangular waveguide can also can be adapted to W-band design. Hence, rare-earth magnets wereThere, placedthe outside the split-block duringinthe measurements. They frequencies around 100 GHz [38,39]. waveguide was realized split-block technology are generating magnetic field of part around T. The measurements wereacarried an15. Aritsu made of brass, awhere the bottom has0.7 a u-shape, being sealed with top lid,out seeusing Figure The 37397C vector network two 3740A-EW extensions for frequencies in waveguide is tapered in analyzer width and(VNA) heightcombined within thewith dielectrically filled section to still provide a single mode propagation. The tapering is done by means of λ/4 transformer steps. The dielectric cavity is made of PTFE (ε r = 2.06 and tan δ = 0.000 222 [40]), being tapered in a triangular shape. The PTFE

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After sealing, the LC is filled into the dielectric cavity through filling holes on the top part of the split-block, see Figure 16. With the help of a syringe the LC is flushed through the cavity, by which the amount of air bubbles within the RF path can be significantly reduced. The LC section within the dielectric cavity has a width of 1.0 mm a height of 0.6 mm and a length of 14.6 mm, providing a differential phase shift of more than 360◦ according to the simulation carried out with CST Studio Suite. This demonstrator can only be biased magnetically, since no biasing electrodes can be included to this design. Hence, rare-earth magnets were placed outside the split-block during the measurements, see Figure 16. They are generating a magnetic field of around 0.7 T. The measurements were carried out using an Aritsu 37397C vector network analyzer (VNA) combined with two 3740A-EW extensions for frequencies in the extended W-band. These first proof-of-concept measurements revealed a good matching of −10 dB over an 8 % bandwidth. This limitation in bandwidth is due to the narrow-banded quarter-wave transformer steps. A differential phase shift between 300◦ to 330◦ was achieved, being 60◦ to 100◦ less than simulated. This is due to the flexible PTFE material which makes an exact processing of the LC cavity difficult. Further, the press-fit sealing causes a narrowing of the LC section. Accompanied with an insertion loss of less than 3 dB, this results in a phase shifter FoM between 105◦ /dB to 148◦ /dB from 90 GHz to 107 GHz, as can be seen from Figure 17. The deviation of the insertion loss has its origin in the interruption of the current walls, due to a non-perfectly closed Version August 24, 2018 submitted to Crystals 14 of 26 split-block.

Rare-earth m

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Figure16. 16.(Left) (Left)Cross Crosssection sectionand and(Right) (right) measurement measurementsetup setupfor forthe themagnetically magneticallybiased biasedW-band W-band Figure rectangular waveguide phase shifter. rectangular waveguide phase shifter.

219 220 221 222 223 224 225 226 227 228 229 230 231 232 233 234 235 236 237 238 239 240 241

However,W-band. it is veryThese challenging to implement an electrical biasing system due to the limited the extended first proof-of-concept measurements revealed a good matching of −space 10 dB as wellanas8 the electrical boundary conditions. Within theislimited of 1.6 mm in width for the over % bandwidth. This limitation in bandwidth due to space the narrow-banded quarter-wave ◦ was ◦ to 100◦ material, dielectrically filled section, several pairs of electrodes toachieved, be placedbeing on the60substrate transformer steps. A phase shift between 300◦ to 330need less than including stepped structures formaterial striplinewhich modemakes suppression. simulated. This is impedance due to the flexible PTFE an exact processing of the LC cavity Simulations in-house director dynamics simulation [8] proved, that two difficult. Further,with the the press-fit sealing causes a narrowing of thetool LC SimLCwg section. Accompanied with an ◦ /dBin pairs of electrodes arethan sufficient butresults not ideal a proper LCFoM alignment, shown 18 [41]. insertion loss of less 3 dB, this in afor phase shifter betweenas105 to Figure 148 ◦ /dB from There it can be seen, a small region with a perpendicular LC alignment will stillloss be left the center 90 GHz to 107 GHz,that as can be seen from Figure 17. The deviation of the insertion hasinits origin in ofthe theinterruption waveguide of when LC iswalls, supposed toabenon-perfectly aligned parallel. However, an odd number of pairs of the the current due to closed split-block. electrodes is notitpossible, since the centered fillingan system of the split-block combination withspace the However, is very challenging to implement electrical biasing systemindue to the limited press-fit PTFE container should be kept inWithin the design. Also, the limited combination as well sealed as the electrical boundary conditions. the limited space of 1.6space mm in width for the with lithography challenges limiting theofrealization moretothan three pairs of substrate electrodes.material, dielectrically filled section, are several pairs electrodesofneed be placed on the including stepped impedance structures for stripline mode suppression. Simulations with the in-house director dynamics simulation tool SimLCwg [8] proved, that two pairs of electrodes are sufficient but not ideal for a proper LC alignment, as shown in Figure 18 [41]. There it can be seen, that a small region with a perpendicular LC alignment will still be left in the center of the waveguide when the LC is supposed to be aligned parallel. However, an odd number of pairs of electrodes is not possible, since the centered filling system of the split-block in combination with the press-fit sealed PTFE container should be kept in the design. Also, the limited space in combination with lithography challenges are limiting the realization of more than three pairs of electrodes. The substrate material on which the electrodes were processed had to be as thin as possible. By this, the impedance of the parasitic microstrip line consisting of the biasing electrodes and the waveguide walls is decreased. Hence, the excitation of the unintended stripline modes is suppressed. For a first proof, a 20 µm thin PET film was used. As metallization, chromium was evaporated on the

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VVB B + +

VVB B + +

Figure Figure 18. 18. Simulated Simulated field field distribution distribution of of the the electrically electrically biased biased W-band W-band rectangular rectangular waveguide waveguide phase phase shifter shifter based based on on LC. LC.

Biasing electrodes Contact pads terminal The substrate material on which the electrodes were processed thin as possible. Biasing electrodes Contact pads had to be asBias Bias terminal By this, the impedance of the parasitic microstrip line consisting of the biasing electrodes and the waveguide walls is decreased. Hence, the excitation of the unintended stripline modes is suppressed. For a first proof, a 20 µm thin PET film was used. As metallization, chromium was evaporated on the PET film. After this, the electrodes including their stepped impedance structures are processed on the substrate by means of lithography steps. Further, the split-block design was changed in the way that the electrodes can be guided to the outside. There, the voltage source can be connected to the biasing lines with the help of spring-probe-pins. The design of the PTFE container was changed as well. Due to the mounting of the electrodes, the taper was chosen as discrete taper instead of a triangular taper. The processed biasing electrodes, the re-designed split-block as well as PTFE container are given in Figure 19. λ/4 PTFE λ/4 transformer transformer steps steps PTFE container container Figure Figure 19. 19. Split-block Split-block design design of of the the electrically electrically biased biased W-band W-band rectangular rectangular waveguide waveguide phase phase shifter shifter based based on on LC. LC.

Figure 18. Simulated field distribution of the electrically biased W-band rectangular waveguide phase

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Biasing electrodes

λ/4 transformer steps

Contact pads

Bias terminal

PTFE container

Figure Figure19. 19. Split-block Split-blockdesign designof ofthe theelectrically electricallybiased biasedW-band W-bandrectangular rectangularwaveguide waveguidephase phaseshifter shifter based basedon onLC. LC.

However, it was not possible to characterize the electrically biased phase shifter, since the electrodes did not work properly. A microscopical investigation of the electrodes revealed cracks within the metallization as well as the substrate material itself, as can be seen in Figure 20. These cracks are originating both from the expansion during the evaporation process as well as the lithography treatments, e.g., etching. The high chromium vacuum melting temperature of 1920 ◦C caused an expansion of the substrate as well as a shrinking while cooling after the evaporation. Hence, microscopic cracks are occurring and no continuous electrical contact could be provided all over the biasing lines. Additionally, the thin PET substrate cannot withstand the contacting of the Version August 24, 2018 submitted to Crystals 16 of 26 spring-probe-pins, which are damaging the substrate irreversibly.

Figure Figure 20. 20. Microscopic Microscopic view view on on an an evaporated evaporated as as well well as as photolithographically photolithographically processed processed Chromium Chromium electrode on a 20 µm thick PET substrate. electrode on a 20 µm thick PET substrate.

|S11 | (dB)

|S21 | (dB)

I simulated At least the simulations can give an State insight into the possible functionality of the electrically biased State II simulated LC metallic waveguide phase shifter in Differential split-block technology, simulated see Figure 21. The matching could be improved compared with the simulation results of the magnetically biased phase shifter in Figure 17. 0 0 However, the insertion loss is increased by 1 dB due to the biasing system, resulting in a phase shifter FoM of around 115◦ /dB. This still includes the ideal LC alignment assumed by CST Studio Suite. −10 −2 anymore and the phase shift, and therefore, In a real measurement, the alignment would not be ideal the phase shifter FoM would degrade.

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256 257 258 259 260 261 262 263 264 265 266 267 268 269

it is obvious that the realization of PET a metallic waveguide based W-band LC phase shifter overHence, the biasing lines. Additionally, the thin substrate cannot withstand the contacting of the isspring-probe-pins, challenging. Further, the integration of a biasing system into the dielectrically filled waveguide is which are damaging the substrate irreversibly. always introducing additional loss. Thus, a new technology needed to be found to easily overcome all At least the simulations can give an insight into the possible functionality of the electrically biased these challenges. This technology was in found in the realization tunable dielectric based LC metallic waveguide phase shifter split-block technology,ofsee Figure 21. The waveguides matching could be on LC, being discussed in the following. improved compared with the simulation results of the magnetically biased phase shifter in Figure 17. Not only the realization of the by biasing electrodes is very system, challenging and their integration However, the that insertion loss is increased 1 dB due to the biasing resulting in a phase shifter ◦ into metallic waveguides introduces additional losses, but also RF leakage due to unintended FoM of around 115 /dB. This still includes the ideal LC alignment assumed by CST Studio Suite.slits In a in themeasurement, split-block design is playingwould a dominant the metallic are therefore, not perfectly real the alignment not berole. idealIfanymore andinterconnections the phase shift, and the sealed in the areas where the wall currents are flowing perpendicularly to these slits, RF leakage occurs. phase shifter FoM would degrade. This effect asitwell as the additional losses from integrated biasingbased electrodes play increasing, Hence, is obvious that the realization of athe metallic waveguide W-band LCanphase shifter non-negligible role with increasing frequency. Further, the higher the frequency, the higher the impactis is challenging. Further, the integration of a biasing system into the dielectrically filled waveguide of the skin depth lossadditional of metallic components. always introducing loss. Thus, a new technology needed to be found to easily overcome all

these challenges. This technology was found in the realization of tunable dielectric waveguides based 6. Dielectric Waveguides on LC, being discussed in the following. Dielectric waveguides are predestinated to overcome aforementioned drawbacks of metallic rectangular waveguides. They offer a low-cost fabrication and low-loss propagation. For example, 3D-printing or injection molding fabrication can be used, and no metallic component is required for the waveguide itself. In addition, the design of LC-based dielectric waveguides is comparatively simple. Due to their electrically unshielded design, electrodes can be placed around the dielectric waveguide without disturbing the propagating wave. Their guiding characteristic can be approximated by the geometrics approach, see Figure 22. Assuming a dielectric slab with a core material, ε 1 surrounded by a cladding material ε 2 , with ε 1 > ε 2 . Internal reflections at the boundary layer of core and cladding occur according to Snell’s law if the angle θ is below the critical angle

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ε2 . (17) ε1 With With the the internal internal reflections, reflections, the the propagating propagating wave wave is is confined confined in in the the dielectric dielectric core. core. Two Two types types of dielectric waveguides are the most prominent ones, the step-index fiber and subwavelength of dielectric waveguides are the most prominent ones, the step-index fiber and subwavelength fiber, fiber, see see Figure Figure 22. 22. cos θC =

ε0 ε2

ε1

ε0 ε1

ε1

Θ

ε2

ε2

|E|

|E|

Figure 22. 22. (Left) (Left) Schematic Schematic cross-section cross-section of of aa DW DW used used for for the the geometric geometric approach, approach, with with εε11 > > εε22 > > εε00.. Figure Cross-section of of two two common common dielectric dielectric waveguide waveguide types: types: (Center) (Center) Step-index Step-index fiber fiber with with εε11 > > εε22 and and Cross-section (Right) Subwavelength Subwavelength fiber. fiber. The The principle principle E-field E-field distributions distributions for for both both types types are are depicted. depicted. (Right)

Rexolite Rexolite

LC core core LC

LC filling filling hole hole LC

Electrodes Electrodes

1.6 mm 1.6 mm 550 µm 550 µm 2.2 mm 2.2 mm 550 µm 550 µm 1.6 mm 1.6 mm

35 mm 35 mm 1.6 mm 1.6 mm ey ey

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ez ez

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ex ex

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Rogers 3003 3003 Rogers

22 mm 5555mm 00 µ 1.1. µm 44 m 5555mm m m 2 20 µ0 µ mmmm mm

Whereas the subwavelength fiber consists of only one single dielectric material, the step-index fiber comprises of two dielectrics. In both cases, evanescent fields are propagating outside the dielectric core. Since the subwavelength fiber only consists of one single dielectric core, the evanescent field propagates outside the waveguide in air, which leads to a low-loss propagating. On the other hand, these fields are sensitive to discontinuities or external influences, such as metallic structures. Using a cladding as for the step-index fiber, all evanescent fields are already decaying inside the dielectric waveguide. These two characteristics are very important for LC LC-based tunable dielectric dielectric waveguides, waveguides, based tunable which will be discussed in the following. The design of asubmitted tunable step-index Version August 24, 2018 to Crystals fiber designed for W-band frequencies which has been recently 18 of 26 published in [42] is shown in Figure 23.

6.5 mm 6.5 mm

Figure 23. 23. Cross Cross section section of of the the step-index step-index fiber fiber phase phase shifter shifter with with the the LC LC cavity cavity highlighted highlighted in in red. red. Figure

The presented design uses Rogers 3003 as non-tunable core material with a permittivity of ε = 3 The presented design uses Rogers 3003 as non-tunable core material with a permittivity of ε rr = 3 and dissipation factor of tan δ ≤ 0.05. As cladding material Rexolite 1422 from C-Lec Plastics Inc., and dissipation factor of tan δ ≤ 0.05. As cladding material Rexolite 1422 from C-Lec Plastics Inc. is Philadelphia, PA, USA is used, with ε = 2.53 and tan δ = 0.000 66. To achieve tunability, LC is used used, with ε r = 2.53 and tan δ = 0.000r66. To achieve tunability, LC in used as core material instead as core material instead of Rogers 3003. The core is untunable and still manufactured from Rogers of Rogers 3003 which is used as core taper, see Figure 23. The phase constant of the propagating 3003, see Figure 23. The phase constant of the propagating wave can now be adjusted by means of wave can now be adjusted by means of the LCs permittivity. As the perpendicular permittivity is the LCs permittivity. As the perpendicular permittivity is below the permittivity of the surrounding below the permittivity of the surrounding cladding, RF leakage would occur, as the propagation cladding, RF leakage would occur, as the propagation condition is not fulfilled anymore. To prevent condition is not fulfilled anymore. To prevent RF leakage, air gaps are included in the cladding to RF leakage, air gaps are included in the cladding to reduce its effective permittivity. With this design reduce its effective permittivity. With this design step, the propagation condition is still fulfilled, even for the lowest LC permittivity, and the evanescent fields are still decaying inside the effective cladding. LC The main advantage of the step-index fiber is that electrodes can be directly attached on the cladding.

Rogers 3003

Copper adhesive electrodes

Figure 24. Fabricated phase shifter and measurement setup of the step-index phase shifter. Two of four

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2 m 55 m 0 1. µm 4 55 mm 2 0µ m m m

step, the propagation condition is still fulfilled, even for the lowest LC permittivity, and the evanescent fields are still decaying inside the effective cladding. The main advantage of the step-index fiber is that electrodes can be directly attached on the cladding. Therefore, the design of the biasing electrodes can be kept very simple. For characterization purposes, transitions were designed to provide a smooth mode conversion from the fundamental TE10 mode of the rectangular metallic waveguide to the Ey11 mode of the dielectric waveguide and vice versa [42,43]. Several tapers, such as for the cladding and core are used to achieve mode matching and for keeping losses due to radiation on a low level. The measurement setup is depicted in Figure 24. It consists of the fabricated dielectric phase shifter including two WR10-to-subwavelength fiber transitions for connection to a VNA. The phase shifter was fabricated in two parts milled out of Rexolite, being glued together by using an UV glue. By this, LC leakage can be easily avoided and two filling holes are left for later LC filling. To show the simplicity of the biasing system, adhesive copper tape was used, being directly stuck on the cladding. Furthermore, a voltage of ±550 V was applied to the parallel plate biasing configurations. The achieved S-parameters as well as the phase shift and FoM are given in Figure 25. It can be observed that the phase shifter is well matched with |S11 | < −15 dB over the whole frequency range. The insertion loss is in a range of 3 dB to 5 dB. In comparison to the simulation, a deviation can be Version August 24, 2018 submitted to Crystals 18 of 26 observed. The reason was found in the splice, which was not included in the simulation. Further, the dielectric loss of the glue itself was unknown at the time of the simulation, but it is expected to be higher than that of Rexolite. Nevertheless, with the LC GT5-26001 an FoM of around 90◦ /dB was ◦ ◦ achieved between to 105 GHz, accompanied with a differential LC core LC filling hole Electrodesphase shift of 220 to 420 , see Rogers 3003 94 GHz Rexolite Figure 25. It is noticeable, that the simulated phase shift was not achieved in the measurement. This is 1.6 mm 35 mm due to fabrication tolerances in the milling process. With a precision of 30 µm, the fabricated LC cavity 550 µm is slightly reduced, resulting in a reduced phase shift. Furthermore, the glue used in the assembling 2.2 mm 1.8 mm 1.6 mm can flow into the cavity and can further decrease the LC volume. In addition, the LC might 550not µm be 1.6 mm e e perfectly aligned. However, both loss and phase 19 insertion mm 6.28 shift mm show the same behavior independence e e of the biasing concept. This is an outstanding result for tunable devices based on 6.5LC, mm especially for in comparison to metallic waveguides. Here, electrical biasing systems always introduce additional 23. Cross section of LC the as step-index fiber phase shifter with thebiasing. LC cavity highlighted in red. losses Figure and could not align the properly than for the magnetic y

y

z

x

LC Rogers 3003

Copper adhesive electrodes

Figure 24. Fabricated phase shifter and measurement setup of the step-index phase shifter. Two of four electrodes are removed to enable a view of the LC core.

291 292 293 294 295 296 297 298 299 300 301

The presented design uses Rogers 3003 as non-tunable core material with a permittivity of ε r = 3 and dissipation factor of tan δ ≤ 0.05. As cladding material Rexolite 1422 from C-Lec Plastics Inc. is used, with ε r = 2.53 and tan δ = 0.000 66. To achieve tunability, LC in used as core material instead of Rogers 3003 which is used as core taper, see Figure 23. The phase constant of the propagating wave can now be adjusted by means of the LCs permittivity. As the perpendicular permittivity is below the permittivity of the surrounding cladding, RF leakage would occur, as the propagation condition is not fulfilled anymore. To prevent RF leakage, air gaps are included in the cladding to reduce its effective permittivity. With this design step, the propagation condition is still fulfilled, even for the lowest LC permittivity, and the evanescent fields are still decaying inside the effective cladding. The main advantage of the step-index fiber is that electrodes can be directly attached on the cladding. Therefore, the design of the biasing electrodes can be kept very simple. For characterization purposes,

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Figure 25. 25. Measured (dashed) results of the step-index phase shifter. For Figure Measured(solid) (solid)and andsimulated simulated (dashed) results of the step-index phase shifter. simplification, s-parameter plots show perpendicular orientation only. For simplification, S-parameter plots show perpendicular 100orientation only. 300

Rexolite

LC core

With the design of the step-index fiber, a high-performance but85 bulky phase shifter was presented, 26 mm75 80 90 95 100 105 110 75 80 85 90 95 100 105 110 which also allows a feasible phase shifter design for terahertz However, 0.6 mmfrequencies. Frequency f (GHz) for the integration Frequency f (GHz) of a dielectric waveguide phase shifter, e.g., in phased arrays, a more compact design is needed. 1.8 mm As shown in 25. Figure 26, a subwavelength fiber is(dashed) well suited for of such application, due to its smaller Figure Measured (solid) and simulated results thean step-index phase shifter. For dimensions. In [43] a phase shifter design was presented. simplification, s-parameter plots show perpendicular orientation only. Figure 26. Cross-section of the phase shifter designed as subwavelength fiber as presented in [43]. Rexolite

LC core

26 mm 0.6 mm

315 316 317 318 319 320 321 315 322 316 323 317 324 318 325 319 326 320 327 321 328 322 329 323 330 324 331 325 332 326 333 327 328 329 330 331 332

found in the splice, which was not included in the simulation. Further, the dielectric loss of the glue 1.8 mm itself was unknown at the time of the simulation, but it is expected to be higher than that of Rexolite. Nevertheless, with the LC GT5-26001 an FoM of around 90 ◦ /dB was achieved between 94 GHz to 105 GHz, accompanied with a differential phase shift of 220◦ to 420◦ , see Figure 25. It is noticeable, that Figure26. 26. Cross-section thephase phaseshifter shifter designed assubwavelength subwavelength fiber aspresented presented [43]. the simulated phase shift was not achieved indesigned the measurement. This isfiber dueas to fabrication tolerances Figure Cross-section ofofthe as inin[43]. in the milling process. With a precision of 30 µm, the fabricated LC cavity is slightly reduced, resulting in a The reduced phase shift. Furthermore, the glue usedisin the assembling can flow the cavity and of the presented phase shifter depicted in Figurethe 26.dielectric The into mainloss part found incross-section the splice, which was not included in the simulation. Further, ofconsists the glue can decrease thefiber LC volume. In addition, the LC might nottunability, be perfectly aligned. However, of a further quadratic Rexolite with an edge lengthbut of 1.8 For a than 0.6 mm LC itself was unknown at the time of the simulation, it ismm. expected to be higher thatcircular of Rexolite. both insertion loss and phase shift show the same behavior independent of the biasing concept. This ◦ /dB core is included. Like metal plates 90 were used the electric biasing, being Nevertheless, with thethe LCstep-index GT5-26001design, an FoM of around wasfor achieved between 94 GHz to is an outstanding result distance for tunable devices based on LC,◦ the especially for infields. comparison to metallic ◦ placed in a large enough to the fiber, to not disturb evanescent The measurement 105 GHz, accompanied with a differential phase shift of 220 to 420 , see Figure 25. It is noticeable, that waveguides. Here, biasingsystem systems always introduce additional losses and not align setup, including theelectrical electric biasing in Figure 27. With a distance ofcould the electrodes the simulated phase shift was not achieved is in shown the measurement. This is due to fabrication tolerances the LC as properly than for the magnetic biasing. of mm from the edgeWith of the Rexolite body, the the metallic platesLC are not disturbing the propagating in3the milling process. a precision of 30 µm, fabricated cavity is slightly reduced, resulting With the design of the step-index fiber, a high-performance but bulky phase shifter was be wave. To ensure a proper position, Rohacell 31HF, from Evonik Industries, Darmstadt, Germany is in a reduced phase shift. Furthermore, the glue used in the assembling can flow into the cavity and presented, which also allows a feasible phase shifter design for terahertz frequencies. However, for the acan special highdecrease performance foam suitable for high frequency applications. Its permittivity around further the LC volume. In addition, the LC might not be perfectly aligned. isHowever, integration of a dielectric waveguide phase shifter, e. g. GT5-26001 in phased arrays, a more compact design is 1.05 with a dissipation factor of 0.01 at 27 GHz. As LC, was used. For characterization, both insertion loss and phase shift show the same behavior independent of the biasing concept. This needed. As shown in Figure 26, a subwavelength fiber is well suited for such an application, due to its the same WR10-to-subwavelength fiber transition used as introduced before. The measured is an outstanding result for tunable devices basedwere on LC, especially for in comparison to metallic smaller dimensions. In [43] a phase shifter design was presented. S-parameters shown in Figure 28. It can be always noticed,introduce that the phase shifterlosses is welland matched the waveguides. are Here, electrical biasing systems additional could over not align The cross-section of the presented phase shifter is depicted in Figure 26. The main part consists whole W-band. the LC as properly than for the magnetic biasing. of a quadratic Rexolite fiber with an edge length of 1.8 mm. For tunability, a 0.6 mm circular LC core With the design of the step-index fiber, a high-performance but bulky phase shifter was be presented, which also allows a feasible phase shifter design for terahertz frequencies. However, for the integration of a dielectric waveguide phase shifter, e. g. in phased arrays, a more compact design is needed. As shown in Figure 26, a subwavelength fiber is well suited for such an application, due to its smaller dimensions. In [43] a phase shifter design was presented. The cross-section of the presented phase shifter is depicted in Figure 26. The main part consists

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WR10 transition

Electrode Figure 27. Measurement setup of the subwavelength fiber phase shifter.


339 340 341 342 343 344 345 346 347 348 349 350 351 352 353 354 355 356 357 358 359 360 361

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The achieved is in a design, range ofmetal 60◦ /dB, aswere the phase shifter reachesbiasing, a differential phase is included. Like theFoM step-index plates used for the electric being placed ◦ ◦ shift between 90 to 160 accompanied with an insertion loss of 1.5 dB to 2.9 dB. Again, the results in a large enough distance to the fiber, to not disturb the evanescent fields. The measurement setup, for magnetic electric biasing areisinshown a goodinagreement highlights the advantage of including the and electric biasing system Figure 27. which With a again distance of the electrodes of 3 mm this technology. from the edge of the Rexolite body, the metallic plates are not disturbing the propagating wave. To ensure a proper position, Rohacell 31HF, from Evonik Industries was used as spacer material. It is a Magnetic Biasing applications. Simulation Electric Biasing special high performance foam suitable for high frequency Its permittivity is around 1.05 −10 factor of 0.01 at 27 GHz. As LC, GT5-26001 0 with a dissipation was used. For characterization, the same WR10-to-subwavelength fiber transition were used as introduced before. The measured S-parameters −15 −1 are shown in Figure 28. It can be noticed, that the phase shifter is well matched over the whole W-band. ◦ The achieved FoM is in a range of 60 /dB, as the phase shifter reaches a phase shift between 90◦ −20 −2 to 160◦ accompanied with an insertion loss of 1.5 dB to 2.9 dB. Again, the results for magnetic and −25are in a good agreement which again highlights −3 electric biasing the advantage of this technology. Another application could be the integration of the subwavelength fiber phase shifter in a fully −30 −4 75 array. 80 In 85 [44], 90 a first 95 fully 75 80 95 100 see 100 105 110 105Figure 110 29 . It dielectric phased dielectric antenna array85was90presented, f (GHz)which is also fed by a special f (GHz)interference (MMI) Frequency consists of four dielectricFrequency rod antennas, multimode power divider. Within this divider, multimode propagation is used to provide power splitting in one single step. Multiple maxima are formed due to the 100superposition and the different propagation 150 constants of all propagable modes inside the waveguide. This interference phenomenon is visualized 80 in Figure 30. With the input position on the left, the interference60phenomenon occurs in the broad waveguide 100 section. It can be seen, that multiple field maxima are formed, with each single maximum feeding a rod antenna. This power divider allows a very compact40design, where power splitting is achieved in 75 and 80 further, 85 90parasitic 95 100 105 110The next one single step.75On 80 one side, this reduces loss, radiation. 85 90 95 100 the 105 dielectric 110 Frequency f (GHz) Frequency f (GHz) step will be the integration of LC based subwavelength fiber phase shifters in the presented antenna array design. For this, LC can be directly inserted in the waveguide section before the rod antennas, Figure 28. 28. Measured Measured (solid) (solid) and simulated (dashed) (dashed) results results of of the the subwavelength subwavelength fiber phase shifter. shifter. Figure simulated phase see Figure 29. Although the LCand integration can be realized quite easily, the main fiber challenge will be the For simplification, s-parameter plots show perpendicular orientation only. For simplification, S-parameter plots show perpendicular orientation only. integration of the required biasing electrode as they can disturb the propagating electromagnetic wave. Furthermore, high permittivity plastics are in the focus of research for minimization of the antenna. Another application could be the integration of the subwavelength fiber phase shifter in a fully 7. Conclusion dielectric phased array. In [44], a first fully dielectric antenna array was presented, see Figure 29 . It consists of paper four dielectric rod antennas, whichonis latest also fed by a special multimode interference (MMI) In this a comprehensive review development of microwave liquid crystal power divider. Within this divider, multimode propagation is used to provide power splitting in technology has been given. Starting from the material properties and their characterization, one single step.for Multiple maxima are formed due to the have superposition and the different requirements mmWave communication systems been introduced. From propagation these basic constants of all propagable modes inside the waveguide. This interference phenomenon visualized considerations the necessity for electronic steerable antennas systems is derived. isAn energy in Figure 30. efficient and high performance realization uses liquid crystals as tunable dielectric. Above 60 GHz Differential phase shift ∆ϕ (◦ )

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16.4 mm

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Multimode interference Rod antennas power divider

Figure 29. Fully Fullydielectric dielectricantenna antenna array presented in [44]. It of consists of antennas four rod which antennas array presented in [44]. a) It (Left) consists four rod are which are fed by a multimode interference power divider; (Right) Further development could be the fed by a multimode interference power divider. b) Further development could be the integration of Version August 24, 2018 submitted to Crystals 22 of 26 integration of the subwavelength subwavelength fiber phase shifter.fiber phase shifter.

max

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10 mm

0 V/m Multimode interference power divider

Rod antennas

Figure Figure 30. 30. Field Field plot plot of of the the fully fully dielectric dielectric antenna antenna array. array. 3. Comparison of the different phase shifter technologies at W-band. With the inputTable position on the left, interference phenomenon occurs in the broad waveguide section. It can be seen, that multiple field maxima are formed, with each single maximum feeding a f where ∆ϕ FoM rod antenna. This power divider allows a very compact design, power IL splitting is achieved Technology Implementation REF in (GHz) (◦ ) (dB) (◦ /dB) one single step. This reduces the dielectric loss, and further, parasitic radiation. The next step will be GaN 2-bit RTPS 80 in the82.5 3.7antenna 17 array[47] the integration of LC-based subwavelength fiber phase shifters presented design. 4-bit loaded line 60 250 3 83 [48] For this, LC can be directly inserted in the waveguide section before the rod antennas, see Figure 29. MEMS Waveguide MEMS 106 47 3.4 13 [49] Although the LC integration can be realized quite easily, the77main challenge will 3-bit switched line 315 5.7 be the 55 integration [45] of the required as they0.18 canµm) disturb the propagating electromagnetic MEMS &biasing CMOSelectrode RTPS (CMOS 65 144 3.2 wave. 42 Furthermore, [50] Ridged waveguide 96 500 of the antenna. 7.2 70 [51] high permittivity plastics are in the focus of research for minimization Metallic waveguide (magnetic bias) Subwavelength fiber Step-index fiber In this paper, a comprehensive review of the latest LC & MEMS Loaded line LC

7. Conclusions

368 369 370 371 372 373 374 375 376 377 378

100 309 2.3 135 [39] 103 140 2.3 60 [43] 103 390 3.9 100 [42] developments crystal 93 190 in microwave 4.5 42 liquid[52]

technology has been given. Starting from the material properties and their characterization, requirements for mmWave communication systems have been introduced. From these basic metallic rectangular waveguide can usually providesystems highest performance. Dueenergy to the considerations, the necessity forcomponents electronic steerable antennas is derived. An implementation of biasing electrodes, the performance degrades substantially. To overcome this efficient and high-performance realization uses liquid crystals as tunable dielectric. Above 60 GHz problem,rectangular the dielectric waveguidecomponents technology, commonly from optics, is adapted toDue millimeter metallic waveguide can usuallyknown provide highest performance. to the waves. With this, an easy integration of biasing electrodes is possible. The first demonstrators presented implementation of biasing electrodes, the performance degrades substantially. To overcome this in this paper show promising results. Table 3 shows an overview on different phase shifter realizations problem, the dielectric waveguide technology, commonly known from optics, is adapted to millimeter at W-band. The high potential of LC-based devices is obvious. All given phase shifters are line phase shifter, which make them operate in a wide frequency band. Usually, MEMS and semiconductor phase shifters have response times in the range of µs, as exemplarily 15 µs in [45]. LC devices are suffering from comparatively high response times, as for example 340 ms in the best case for a loaded CPW line [46] and approx. 1 min for the presented dielectric waveguide phase shifters. Further work includes optimization of components based on

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waves. With this, an easy integration of biasing electrodes is possible. The first demonstrators presented in this paper show promising results. Table 3 shows an overview on different phase shifter realizations at W-band. The high potential of LC-based devices is obvious. All given phase shifters are line phase shifter, which make them operate in a wide frequency band. Usually, MEMS and semiconductor phase shifters have response times in the range of µs, as exemplarily 15 µs in [45]. LC devices are suffering from comparatively high response times, as for example 340 ms in the best case for a loaded CPW line [46] and approximately 1 min for the presented dielectric waveguide phase shifters. Further work includes optimization of components based on dielectric waveguides. Specific focus is on the fabrication of the devices and on LC integration. Here, modern additive manufacturing techniques can be implemented. Table 3. Comparison of different phase shifter technologies at W-band. Technology

Implementation

f (GHz)

GaN

2-bit RTPS

80

MEMS

4-bit loaded line Waveguide MEMS 3-bit switched line

MEMS & CMOS

RTPS (CMOS 0.18 µm)

LC

Ridged waveguide Metallic waveguide (magnetic bias) Subwavelength fiber Step-index fiber

LC & MEMS

Loaded line

∆ϕ (◦ )

IL (dB)

82.5

3.7

17

[47]

FoM (◦ /dB)

REF

60 106 77

250 47 315

3 3.4 5.7

83 13 55

[48] [49] [45]

65

144

3.2

42

[50]

96 100 103 103

500 309 140 390

7.2 2.3 2.3 3.9

70 135 60 100

[51] [39] [43] [42]

93

190

4.5

42

[52]

The tunable dielectric waveguide technology is one example for research beside LC material engineering. Material optimization is ongoing [9] and will stay a major task in future. The development of new tunable structures is the second major field of research. Recently, planar structures utilizing plasmonic or slow wave effects have been presented [53–55]. These structures show a huge potential, as they faciitate integration. Unfortunately, these types of components usually show high insertion losses. The main advantages compared to metallic rectangular and dielectric waveguides are much faster response times of the components. This is due to a much thinner LC layer which can be tuned faster with lower voltages. Also, substrate integrated waveguides (SIW) [56,57] will probably play a significant role in the future. The right technology has always to been chosen based on a tradeoff. Researchers in academia and industry are working hard iso that LC-based devices can manifest their large potential in many applications. Author Contributions: Conceptualization: H.M., M.J., R.R. and E.P.; Investigation: M.J., R.R. and E.P.; Writing: H.M., M.J. and R.R.; Review & Editing, M.N., E.P. and R.J.; Supervision, H.M. and R.J.; Funding Acquisition, H.M. and M.J. Funding: This research was partially funded by DFG grants JA921/68-1, MA6281/4-1, JA921/29-2 and BMWi/DLR under contract 50YB1113. Acknowledgments: The authors gratefully acknowledge Merck KGaA and CST AG for supplying Liquid Crystal mixtures and CST Microwave Studio, respectively. We acknowledge support by the German Research Foundation and the Open Access Publishing Fund of Technische Universität Darmstadt. Conflicts of Interest: The authors declare no conflict of interest.

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Abbreviations The following abbreviations are used in this manuscript: 5G D DC DW E ε eff εk ε⊥ FoM GEO HPBW IL IP3 ITU Ka -band Ku -band LC LCD LEO MMI mmWave OLED PIM PET PTFE RF SIW TDS TN UV VNA W-band WRC

System for Mobile Communication of the 5th generation Displacement field Direct current Dielectric waveguide Electric field effective permittivity effective permittivity of an LC volume for parallel alignment effective permittivity of an LC volume for perpendicular alignment Figure of Merit geostationary orbit (satellite) Half-power beam width Insertion loss (IL = −|S21 |) Third-order intercept point International Telecommunication Union Communications band from 26.5 GHz to 40 GHz Communications band from 12 GHz to 18 GHz Liquid Crystal Liquid crystal display low earth orbit (satellite) Multimode interference Millimeter wave Organic LED Passive intermodulation Polyethylenterephthalat Polytetrafluorethylene (Teflon) Radio frequency Substrate integrated waveguide time-domain spectroscopy twisted nematic Ultra violet Vector network analyzer Communications band from 75 GHz to 110 GHz World Radio Conference

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