New CFOA-Based Single-Element-Controlled Sinusoidal Oscillators

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IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 55, NO. 6, DECEMBER 2006

New CFOA-Based Single-Element-Controlled Sinusoidal Oscillators D. R. Bhaskar and Raj Senani

Abstract—New single-element-controlled sinusoidal oscillator (SECO) circuits using only two current-feedback operational amplifiers (CFOAs) and only five passive elements [two/three grounded capacitors (GC) and three/two resistors] are presented, which not only enlarge the previously known class of twoCFOA-GC SECOs but also provide new configurations that possess properties not available in the previously known circuits. The new oscillators are useful from the viewpoint of applications in several instrumentation and measurement situations such as oscillator-based capacitance measurement schemes, realization of very low frequency oscillators, and the design of voltage-controlled oscillators. Experimental results based upon commercially available AD844-type CFOAs are included, which confirm the practical workability of the new oscillator configurations. Index Terms—Current-feedback operational amplifiers (CFOAs), sinusoidal oscillators, voltage-controlled oscillators (VCOs).

I. I NTRODUCTION

S

INGLE-ELEMENT-CONTROLLED sinusoidal oscillators (SECOs) have numerous applications in instrumentation and measurement systems (see [1]–[5] and the references cited therein). Thus, sinusoidal oscillators may be used as or in test oscillators or signal generators for testing of radio receivers, measurement of standing-wave ratio, signal-to-noise ratio, etc., transducer oscillators (in conjunction with resistive/capacitive transducers), measurement of an unknown capacitance in some oscillator-based schemes, realization of very low frequency (VLF) oscillations (i.e., 1 Hz and lower) for biomedical, geophysical, or control applications, and realization of voltagecontrolled oscillators (VCOs) useful for analog phase-locked loops, A/D converters, frequency response display systems, spectrum analyzers, etc. Traditionally, the IC operational amplifier (op amp) has been the work horse of sinusoidal oscillators (for example, see [1], [2], and [6] and the references cited therein). However, lately, the current-feedback operational amplifier (CFOA), particularly the four-terminal type such as AD844 providing an externally accessible compensation terminal, is attracting prominent attention as an alternative building block for analog circuit designs because of several advantages that it offers over Manuscript received October 23, 2005; revised August 6, 2006. D. R. Bhaskar is with the Department of Electronics and Communication Engineering, Faculty of Engineering and Technology, Jamia Millia Islamia, New Delhi 110 025, India. R. Senani is with the Division of Electronics and Communication Engineering, Netaji Subhas Institute of Technology, New Delhi 110 075, India (e-mail: [email protected]). Color versions of Figs. 3–5, 7, and 8 are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIM.2006.884139

the traditional voltage-mode op amp (VOA)—such as nearly constant bandwidth independent of gain, much higher slew rates, and ease of designing various functional circuits with the least possible number of external components and without requiring any component matching conditions (see [7]–[11] and the references cited therein). However, CFOAs usually have dc offset and gain errors and are, therefore, not favored for dc/VLF applications (for instance, see [29] and the references cited therein). Interest in designing sinusoidal oscillators using CFOAs grew when it was realized that CFOA-based oscillators can offer improved performance in terms of frequency accuracy, dynamic range, distortion level, and frequency span as compared to their VOA-based counterparts [12]. Consequently, a number of such circuits have been evolved, which can be practically implemented using commercially available AD844type CFOAs [13]–[26]. The use of grounded capacitors (GCs) is advantageous from the viewpoint of IC implementation as well as ease of incorporating the effect of parasitic capacitances of the CFOAs, which can be absorbed in them. Although a number of single CFOA-based oscillators have been evolved [12], [13], [15]–[18], [21], [24], [25], none of them employ both GCs. However, given two CFOAs, sinusoidal oscillators providing noninteracting controls of the condition of oscillation (CO) as well as the frequency of oscillation (FO) through separate resistors as well as employing GCs appear feasible, as is evident from the publication of a number of two-CFOA-based singleresistance-controlled oscillators (SRCO) in the past as in [14], [19], [20], [22], and [23]. The complete family of 14 SRCOs employing only two CFOAs, two GCs, and three resistors was systematically derived in [20], [22], and [23]. These SRCOs provide the following features: 1) use of two GCs; 2) use of two CFOAs; and 3) independent control of CO and FO through two separate resistors. The 14 SRCOs possessing the above features were all based upon the following tuning laws: CO : FO :

R1 = R3
1 ω0 = C1 C2 R2 R3

(1) (2)

which gives the resulting circuits, the control of CO by R1 , and that of FO by R2 . Although, from a purely circuit theoretic or aesthetic viewpoint, major attention has been received by sinusoidal oscillators providing control of both FO and CO through independent resistors in accordance with the tuning laws of (1) and (2), oscillators governed by other types of tuning laws, which thereby provide CO control through a capacitor or FO control through

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BHASKAR AND SENANI: NEW CFOA-BASED SINGLE-ELEMENT-CONTROLLED SINUSOIDAL OSCILLATORS

Fig. 1.

2015

Proposed new single-element-controlled oscillators.

a single capacitor or provide an expression for FO containing a difference term, are not necessarily outclassed when considered from the perspective of instrumentation and measurement applications. For instance, oscillators providing single element control (SEC) of oscillation frequency through a capacitor can be used as transducer oscillators in conjunction with capacitive transducers; those providing CO control through a capacitor can be used in some capacitance measurement schemes such as those mentioned in [3]–[5]; and, finally, those having a difference term in the expression for FO may be usefully employed as VLF oscillators [6]. It is these kinds of oscillators with which this paper is concerned. Therefore, when viewed from this angle, the catalog of 14 two-CFOA oscillators presented in [23] does not really exhaust all possible GC SECOs realizable from two CFOAs and is, therefore, far from being complete! Thus, given two CFOAs and using only GCs along with two or three resistors, a number of new sinusoidal oscillator configurations should be possible (beyond those of [23]), which may have tuning laws different from those of (1) and (2) and yet satisfy the single-element-controllability conditions. The objective of this paper is, therefore, to present a number of such new two-CFOA-GC SECOs as an addition to those of [23], which are useful from the viewpoint of applications in instrumentation and measurement.

II. N EW O SCILLATOR C ONFIGURATIONS The sinusoidal oscillator circuits to be presented here have been derived by framing new tuning laws different from those of (1) and (2), determining the required [A] matrices, converting the [A] matrices into node equations, and, finally, synthesizing the resulting node equations by physical circuits using CFOAs and RC elements, as per the procedure outlined earlier in [20], [22], and [23]. A brief outline of the methodology, the definition of the [A] matrix, and the specific [A] matrices considered for the derivation of the new circuits, as well as an illustrative example, are given in the Appendix. The resulting new circuit configurations are shown in Fig. 1, where each CFOA has input terminals x, y and output terminals z, w, is characterized by the instantaneous terminal equations iy = 0, vx = vy , iz = ix , and vw = vz (see CFOA-1 in oscillator 1). Fig. 1 also shows the CO and the FO for all the circuits. The following features of the new circuits may now be noted: 1) Circuit 8 still qualifies for feature 3) although it uses three GCs. 2) Circuit 7 also qualifies for condition 3), but frequency control is through a (grounded) capacitor; it is independently controllable with either C1 or C2 .

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IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 55, NO. 6, DECEMBER 2006

3) Circuits 1–5 have tuning laws that do not conform to (1) and (2), and yet, these circuits do possess features 1) and 2), which are outlined in Section I. 4) Circuit 6 presented in Fig. 1 is the only oscillator circuit realizable with a bare minimum of only four passive elements, and yet, it has both capacitors grounded (this circuit is derivable from circuit 5 by deleting resistor R2 ). It may be pointed out that although a similar oscillator using second generation current conveyors has recently appeared in [27], in comparison to the circuit of [27], the new circuit 6 offers the advantage of a buffered voltagemode output. It is useful to examine the effect of various CFOA nonideal parameters on the FO of various circuits. Taking into consideration the finite X-terminal input resistance rx and the parasitic impedance at the Z-terminal, consisting of a resistance Rp in parallel with a capacitance Cp , the various nonideal expressions for the oscillation frequencies ω0i , i = 1, . . . , 8, for the circuits shown in Fig. 1 are shown in (3)–(10) at the bottom of the page.

From these expressions, it is seen that the influence of CFOA parasitics on the performance of the oscillators can be reduced by choosing external resistances to be much greater than rx and much smaller than Rp and selecting external capacitors to be much larger than Cp . Note that with rx
Ri
Rp and Ci  Cp , i = 1–3, all expressions would approximate to their ideal values. On the other hand, in view of different natures of nonideal expressions, it can be seen that there will be a different upper bound on the frequency, which can be generated from a chosen oscillator. From these expressions, the nonideal values of the oscillation frequencies with component values chosen according to the desired frequency and with known values of the parameters rx , Rp , and Cp for the CFOA employed can be easily estimated. To give a quantitative idea about the influence of these parasitics, we have calculated the percentage of errors in FO for all the eight oscillators that are designed for a frequency of 10 kHz with capacitor values selected in the range of 1–2 nF, resistor values in the range of 5.6–25.35 kΩ, and assuming that rx = 50 Ω, Cp = 4.5 pF, and Rp = 3 MΩ. These

         Rx R3 Rx R2 R1 R2 R1 R3  1− 1 − 1 + + 1 + − + R2 −R1 Rp Rp Rp Rp Rx Rp Rx R2 − R1       2 C C R R R p p C1 C2 R1 R2 R3 x 1 + C1 1 + C2 1 + Rx1 + Rx3 + R1 R 3     R1 R2 R2 Rx   1 + R3 Rp + R1 Rp − 1 R2 − R1 1   Rp R2−R     Cp Cp Rx C1 C2 R1 R2 R3 1 + C1 1 + C2 1 + R1      R1 R1 R2   x 1 + R2R−R 1 + + R R R R2 − R1 1 p p x    2 2Cp Rx Rx C1 C2 R1 R2 R3 x 1 + C1 1 + R2 + R3 + RR2 R 3      2 Rx R1 R2 R1 R3 Rx R3 R1 Rx Rx   1 + + 1 + + + + + 2 2 2 2 Rp R1 Rp Rp Rp Rp 1   Rp      Cp Cp Rx Rx R2 C1 C2 R1 (R2 + R3 ) 1+ 1+ 1+ 1+ 

2 ω01 =

2 ω02 =

2 ω03 =

2 ω04 =



2 = ω05

2 ω08

C2

R1

(4)

(5)

(6)

R1 (R2 +R3 )

R2 − R1 C1 C2 R1 R2 R3        2 2 R3 R1 R3 1 x x 1 R2 R3 x x x x 1+ RR 1+ R − RR2x−R 1− +R +R + RR1 R + RR1 R + RRp R − R13 + Rp (R 2 (R −R ) R R R R −R ) 2 1 1 2 3 2 3 1 p 3 2 1 p     × 2 2 C C Rx Rx Rx Rx x 1+ Cp1 1+ Cp2 1+ R R1 + R2 + R3 + R1 R2 + R1 R3 





1 C1 C2 R1 R3



1+

R3 Rx 2 Rp

2 Rx 2 Rp



+

Rx R3

  (7)



3 1 Rx + RR1 R + RR 2 2 p  p  C C Rx x 1 + Cp1 1 + Cp2 1+ R + R2 R3       2 Rx Rx 2Rx R3 R1 Rx 1 1   1 + R3 1 + R1 + + + + 1 + − Rp RP R2 R3 R2 R3 Rp R2 R3 R2 R3 1       = 2 Cp Cp Rx Rx 2Rx C1 C2 R1 R3 1 + C1 1 + C2 1 + R2 + R3 + R2 R3    Rx R2 Rx   1 + 1 + 2 Rp R2 1        = 2 Cp C Rx Rx Rx Rx 1 C1 C2 R1 R2 1 + C1 1 + Rp + R2 + Rp R2 + C3 Rx 1 + R2 1 + Cp3 C2 R1 1 +

2 ω06 =

2 ω07

C1



(3)

(8)

(9)  Cp C1

 +

1 C1 Rp

 (10)

BHASKAR AND SENANI: NEW CFOA-BASED SINGLE-ELEMENT-CONTROLLED SINUSOIDAL OSCILLATORS

2017

TABLE I FREQUENCY STABILITY FACTORS FOR THE NEW OSCILLATOR CIRCUITS DEPICTED IN FIG. 1

errors are found to be −1.15%, −1.33%, −0.074%, −0.89%, −0.786%, −1.88%, −0.82%, and −0.622%, respectively, for oscillators 1–8 and are, thus, indeed very low. In general, the applicability of such CFOA-based oscillators is usually limited to a few hundred kilohertz [28]. Frequency stability may be considered to be another important figure of merit on the basis of which various oscillators can be compared. The frequency stability factor S F is defined as S F = dφ(u)/du, where u = ω/ω0 , and φ(u) is the phase of the open-loop transfer function of the oscillator circuit and for oscillators. The expressions for the frequency stability factors for all the new oscillators have been derived and are shown in Table I. From these values, it is seen that oscillator 7 has the least value of S F , and the next higher value is attained by oscillator 3. In oscillator 4, S F can be made as large as desired by keeping n small. On the other hand, in the remaining oscillators 1, 2, 5, 6, and 8, S F can be made as large as desired by keeping n larger than unity. Thus, oscillators 1, 2, 4, 5, 6, and 8 enjoy the highest possible value of S F and are in the same class as the earlier SRCOs of [8], [26], and [28]. III. S IGNIFICANCE OF THE N EW O SCILLATORS We now explain how the new circuits may be useful in some instrumentation and measurement situations. An inspection of Fig. 1 reveals that the new oscillators 1, 2, 3, and 5 contain a difference term in the expression for FO, which is of the type √ 1−n ω0 = RC

(11)

where n is the frequency-controlling resistor ratio and, thus, qualify to be used for generating VLF oscillations (i.e., 1 Hz and lower) in the manner of [6] by choosing “n” such that (1 − n) can be made small so that lower values of f0 are achievable.

Oscillator circuits 6 and 8 appear suitable for the measurement of an unknown capacitance by the method of [3]–[5]. For this, the unknown capacitance (whose value is to be determined) can be connected as C1 , and then, the known variable capacitance C2 is to be varied until the circuit just starts (or stops) to oscillate, as has been done in [3]–[5]. We now show how three of the new oscillator circuits can be easily modified to realize VCOs by incorporating a nonlinearity-cancelled FET/MOSFET [8] without requiring any additional CFOAs. The new VCOs realizable from three of the new SECOs are shown in Fig. 2. To conserve space, we explain here the mechanism of VCO realization only for the oscillator circuit 1, which is shown in Fig. 2 as circuit A. In this circuit, resistance R1 has been replaced by a FET, and its gate voltage has been made equal to (Vc + V01 )/2 by using two equal-valued resistors connected to Vc and V01 , respectively, with their common junction connected to the gate terminal. Note that the topology of this oscillator (circuit 1 in Fig. 1) is structured such that it permits utilizing the unused voltage-output terminal W1 of CFOA1 easily from where a voltage exactly equal to the drain voltage of the FET is available via the route X2 -Y2 -Z1 -W1 that can be easily tapped without disturbing any other voltages/currents in the circuit. Thus, in the VCO shown as circuit A in Fig. 2, the equivalent resistance of the FET rDS (replacing R1 ) is now determined from the following equations: iD =



2 IDSS VDS − V )V − (V GS p DS Vp2 2

1 VGS = (Vc + V01 ) 2 VDS = V01

(12)

(13)

and turns out to be rDS =

2Vp2 V01 = iD IDSS (Vc − 2Vp )

(14)

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IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 55, NO. 6, DECEMBER 2006

Fig. 3. Variation of oscillation frequency with resistance R1 of circuit 4 in Fig. 1.

Fig. 2. VCOs realizable from oscillators 1, 5, and 6 in Fig. 1.

where Vp is the pinchoff voltage, IDSS is the reverse saturation current, and rDS is the equivalent (linearized) drain-to-source resistance [8]. 2 in (12) is Thus, the nonlinearity term proportional to VDS effectively cancelled, without requiring any additional CFOA, and the overall circuit thus realizes a VCO using a nonlinearitycancelled voltage-controlled resistance (VCR). A careful check of the circuits in Fig. 1 reveals that, apart from circuit 1, the topologies of the circuits 5 and 6 also permit conversion into VCOs in this manner. This results in two more VCOs, which are shown in Fig. 2 as circuits B and C. Note that the conversion of the SRCOs into VCOs in the manner shown here has not been known in any other earlier publications on CFOA-based oscillators.

Fig. 4. Variation of percentages of error between the nonideal and ideal frequencies of oscillation for oscillator 4.

IV. E XPERIMENTAL R ESULTS The workability of all the new sinusoidal oscillator circuits has been confirmed in hardware using a commercially available AD844-type of CFOA and 1% tolerance RC components. All the circuits have been found to work as expected. Some sample experimental results are given in Figs. 3–8. The variability of oscillation frequency with resistor R1 for the oscillator circuit 4 in Fig. 1 is shown in Fig. 3. The component values were C1 = 2 nF, C2 = 1 nF, R2 = R3 = 4.7 kΩ, and ±VCC = ±15 V dc,

Fig. 5.

Typical waveform (49.9 kHz) from circuit 8 in Fig. 1.

and R1 was varied from 100 Ω to 100 kΩ. Fig. 4 shows that over the frequency range of interest, the percentage of errors in the frequencies due to the effect of CFOA parasitics is not more than 2.5%, which explains the close correspondence between the experimental data with the nonideal curve in Fig. 3.

BHASKAR AND SENANI: NEW CFOA-BASED SINGLE-ELEMENT-CONTROLLED SINUSOIDAL OSCILLATORS

2019

of R1 ranges from 2.11% to 3.08% as opposed to 0.56% to 1.54% without JFET (with the circuit designed to produce the same value of frequencies in both cases). Thus, it is confirmed that the distortion introduced by the JFET is indeed small due to the cancellation of the nonlinearity. Fig. 8 shows a typical waveform [f0 = 1.098 Hz and 12.8 V (p-p)] generated by oscillator 1 in VLF mode. The experimental results presented here thus establish the workability of the new propositions. Fig. 6.

V. C ONCLUDING R EMARKS

Transient response of oscillator 8 in Fig. 1.

Fig. 7. Variation of oscillation frequency with control voltage Vc for circuit A in Fig. 2.

Eight new sinusoidal oscillator circuits, each employing only two CFOAs, all GCs, and two/three resistors, have been introduced in this paper. The new circuits have been derived by applying the methodology of [20], [22], and [23] in conjunction with several new types of tuning laws not considered in these papers. Consequently, several of the resulting SECOs were seen to possess attractive features such as realizability of SRCOs using a bare minimum of only four passive components, suitability for realization of VLF oscillations, the possibility of being used in capacitance measurement schemes, and easy convertibility into VCOs using nonlinearity-cancelled VCRs. The practicability of the new oscillator circuits has been demonstrated by experimental results based upon the commercially available AD844-type CFOAs. It is believed that the proposed oscillators provide new and interesting possibilities for the aforementioned instrumentation and measurement applications, which can now be explored in more detail further. A PPENDIX M ETHOD OF S YNTHESIZING THE P ROPOSED C IRCUITS A canonic second-order (i.e., employing only two capacitors) oscillator can, in general, be characterized by the following autonomous state equation:

a11 a12 x1 x˙ 1 = (15) x˙ 2 a21 a22 x2 where



a11 [A] = a21

a12 . a22

(16)

From (15), the characteristic equation Fig. 8. Typical waveform generated from oscillator 1 in VLF mode [f0 = 1.098 Hz and V0 = 12.8 V (p-p)].

s2 − (a11 + a22 )s + (a11 a22 − a12 a21 ) = 0

(17)

gives the CO and FO as A typical waveform generated by circuit 8 in Fig. 1 with component values C1 = C2 = C3 = 1 nF, R1 = 1 kΩ, R2 = 10 kΩ, and ±VCC = ±15 V dc is shown in Fig. 5. For the waveform shown in Fig. 5, the percent total harmonic distortion (% THD) was found to be 1.1%. The transient response of this oscillator showing the buildup of the oscillations (for the same component values) is shown in Fig. 6. Variation of oscillation frequency with control voltage Vc for the VCO circuit A in Fig. 2 using component values C1 = C2 = 10 nF, R2 = R3 = 2.2 kΩ, and R = 10 kΩ and employing BFW10 FET is shown in Fig. 7. The % THD in the output waveform with JFET circuitry included in place

(a11 + a22 ) = 0  ω0 = (a11 a22 − a12 a21 ).

(18) (19)

The proposed methodology involved selecting the parameters aij , i = 1, 2 and j = 1, 2, in accordance with the required features, converting the [A] matrices into node equations, and, finally, synthesizing the resulting node equations by physical circuits using CFOAs and RC elements. In [20], [22], and [23], a total of 14 different choices of [A] matrices have been found, which lead to 14 different CFOAbased oscillators, all characterized by tuning laws of type (1)

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and (2). In this paper, we have considered the following eight choices of [A] matrices (not covered in [20], [22], and [23]), which lead to the eight new oscillators shown in Fig. 1:  1  −1 [A1 ] =

C1 R3 C1 R3 1 −1  C2 R1 C2 R2  1 1 1 C1 R2 − R3

[A2 ] =   [A3 ] =  [A4 ] =

1 C1





−1 C1 R1

1 R3

−

1 R1

1 R3

−

1 R1





−1 C2 R3 1 C1



 

1 R2

R1 +R2 C1 R1 (R2 +R3 )

−1 C2 R3 −1 C1 R3



 [A6 ] =  [A7 ] = 

1 C2





1 1 R3 + R1 1 C1 R3 1 R1

+

0 1 C2 R1 C3 C2 C1 R1 1 C2 R1

1 R3



 C2 R3 

1 C2

−

−1

1 C2 R3

R2 −1 C2 R C2 R1 (R 1 2 +R3 )  1 1 1 −1 C1 R2 + R3 C1 R3

[A5 ] = 

[A8 ] =

1 C2

−1 C1 R3

1 C2

1 R3



−





−1 C2 R3 −1 C1 R3

 

1 R3

−

1 R1

C3 C2 C1 R1

+

1 C1 R2

1 R2





+

−1 C2 R1

 .

To illustrate the procedure outlined earlier, consider the case of matrix [A6 ]. The node equations, which were obtained from [A6 ], can be written as C1

dx1 (x1 − x2 ) = dt R3

(20)

C2

x1 dx2 (x1 − x2 ) = + dt R1 R3

(21)

where the state variables x1 and x2 are the voltages across C1 and C2 , respectively. We now show that oscillator 6 is, in fact, an embodiment of the node equations (20) and (21). In oscillator 6, it may be seen that the current in capacitor C1 , which is represented by C1 (dx1 /dt), flowing out of the Z1 -terminal of CFOA-1, is equal to the current in R3 , which is given by (x1 − x2 )/R3 , flowing out of the X1 -terminal, thus, satisfying (20). Similarly, the current in C2 , which is represented by C2 (dx2 /dt), flowing out of the Z2 -terminal of CFOA-2 is the sum of the current in R3 , which is given by (x1 − x2 )/R3 , and the current going out of the X2 -terminal, which is given by x1 /R1 , thus, satisfying (21). The remaining oscillators have been similarly derived from the corresponding [A] matrices. ACKNOWLEDGMENT The authors would like to thank D. Prasad, R. K. Sharma, and A. K. Singh for their assistance in the preparation of the manuscript and the anonymous reviewers for their constructive feedback.

R EFERENCES [1] R. Senani, “New types of sine wave oscillators,” IEEE Trans. Instrum. Meas., vol. IM-34, no. 3, pp. 461–463, Sep. 1985. [2] R. Senani and D. R. Bhaskar, “Single op-amp sinusoidal oscillators suitable for generation of very low frequencies,” IEEE Trans. Instrum. Meas., vol. 40, no. 4, pp. 777–779, Aug. 1991. [3] S. Natarajan, “Measurement of capacitances and their loss factors,” IEEE Trans. Instrum. Meas., vol. 38, no. 6, pp. 1083–1087, Dec. 1989. [4] W. Ahmad, “A new simple technique for capacitance measurement,” IEEE Trans. Instrum. Meas., vol. IM-35, no. 4, pp. 640–642, Dec. 1986. [5] S. S. Awad, “Capacitance measurement based on operational amplifier circuit: Error determination and reduction,” IEEE Trans. Instrum. Meas., vol. 37, no. 3, pp. 379–382, Sep. 1988. [6] A. S. Elwakil, “Systematic realization of low-frequency oscillators using composite passive–active resistors,” IEEE Trans. Instrum. Meas., vol. 47, no. 2, pp. 584–586, Apr. 1998. [7] C. Toumazou and F. J. Lidgey, “Current feedback op-amps: A blessing in disguise?” IEEE Circuits Devices Mag., vol. 10, no. 1, pp. 34–37, Jan. 1994. [8] R. Senani, “Realization of a class of analog signal processing/signal generation circuits: Novel configurations using current feedback op-amps,” Frequenz: J. Telecommun., vol. 52, no. 9/10, pp. 196–206, 1998. [9] A. M. Soliman, “Applications of the current feedback amplifier,” Analog Integr. Circuits Signal Process., vol. 11, no. 3, pp. 265–302, Nov. 1996. [10] J. W. Horng, “New configuration for realizing universal voltage-mode filter using two current feedback amplifiers,” IEEE Trans. Instrum. Meas., vol. 49, no. 5, pp. 1043–1045, Oct. 2000. [11] S. I. Liu and D.-S. Wu, “New current feedback amplifier based universal biquadratic filter,” IEEE Trans. Instrum. Meas., vol. 44, no. 4, pp. 915–917, Aug. 1995. [12] S. Celma, P. A. Martinez, and A. Carlosena, “Current feedback amplifiers based sinusoidal oscillators,” IEEE Trans. Circuits Syst. I, Fundam. Theory Appl., vol. 41, no. 12, pp. 906–908, Dec. 1994. [13] S. I. Liu, C. S. Shih, and D. S. Wu, “Sinusoidal oscillators with single element control using a current-feedback amplifier,” Int. J. Electron., vol. 77, no. 6, pp. 1007–1013, 1994. [14] S. I. Liu and J.-H. Tsay, “Single-resistance-controlled sinusoidal oscillator using current-feedback amplifiers,” Int. J. Electron., vol. 80, no. 5, pp. 661–664, 1996. [15] R. Senani and V. K. Singh, “Synthesis of canonic single-resistancecontrolled-oscillators using a single current-feedback-amplifier,” Proc. Inst. Electr. Eng.—Circuits, Devices Syst., vol. 143, no. 1, pp. 71–72, 1996. [16] M. T. Abuelma’atti, A. A. Farooqi, and A. M. Al-shahrani, “Novel RC oscillators using the current-feedback operational amplifier,” IEEE Trans. Circuits Syst. I, Fundam. Theory Appl., vol. 43, no. 2, pp. 155–157, Feb. 1996. [17] P. A. Martinez, S. Celma, and J. Sabadell, “Designing sinusoidal oscillators with current-feedback amplifiers,” Int. J. Electron., vol. 80, no. 5, pp. 637–646, 1996. [18] M. T. Abuelma’atti and A. M. Al-shahrani, “New CFOA-based sinusoidal oscillators,” Int. J. Electron., vol. 82, no. 1, pp. 27–32, 1997. [19] P. A. Martinez, J. Sabadell, and C. Aldea, “Grounded resistor controlled sinusoidal oscillator using CFOAs,” Electron. Lett., vol. 33, no. 5, pp. 346–348, Feb. 1997. [20] R. Senani and S. S. Gupta, “Synthesis of single resistance controlled oscillators using CFOAs: Simple state-variable approach,” Proc. Inst. Electr. Eng.—Circuits, Devices Syst., vol. 144, no. 2, pp. 104–106, Apr. 1997. [21] M. T. Abuelma’atti and A. M. Al-shahrani, “Novel CFOA-based sinusoidal oscillators,” Int. J. Electron., vol. 85, no. 4, pp. 437–441, 1998. [22] S. S. Gupta and R. Senani, “State variable synthesis of single resistance controlled grounded capacitor oscillators using only two CFOAs,” Proc. Inst. Electr. Eng.—Circuits, Devices Syst., vol. 145, no. 2, pp. 135–138, Apr. 1998. [23] ——, “State variable synthesis of single resistance-controlled grounded capacitor oscillators using only two CFOAs: Additional new realizations,” Proc. Inst. Electr. Eng.—Circuits, Devices Syst., vol. 145, no. 6, pp. 415– 418, Dec. 1998. [24] A. Toker, O. Cicekoglu, and H. Kuntman, “On the oscillator implementations using a single current feedback op-amp,” Comput. Electr. Eng., vol. 28, no. 5, pp. 375–389, Sep. 2002. [25] E. O. Gunnes and A. Toker, “On the realization of oscillators using state equations,” AEU, vol. 56, no. 5, pp. 1–10, 2002. [26] D. R. Bhaskar, “Realisation of second-order sinusoidal oscillator/filters with non-interacting controls using CFAs,” Frequenz: J. Telecommun., vol. 57, no. 1/2, pp. 12–14, 2003.

BHASKAR AND SENANI: NEW CFOA-BASED SINGLE-ELEMENT-CONTROLLED SINUSOIDAL OSCILLATORS

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D. R. Bhaskar received the B.Sc. degree from Agra University, Agra, India, the B.Tech. degree from the Indian Institute of Technology (IIT), Kanpur, India, the M.Tech. degree from IIT, Delhi, India, and the Ph.D. degree from the University of Delhi. He was an Assistant Engineer with Delhi Electric Supply Undertaking (DESU) from June 1981 to January 1984 and a Lecturer from 1984 to 1990 and a Senior Lecturer from 1990 to 1995 with the Department of Electrical Engineering, Delhi College of Engineering. He joined the Department of Electronics and Communication Engineering, Faculty of Engineering and Technology, Jamia Millia Islamia, New Delhi, India, in July 1995, as a Reader and later became a Professor in January 2002. He served as the Head of the department between 2002 and 2005 under the rotational headship prevalent at Jamia Millia Islamia. His teaching and research interests are in the areas of bipolar and CMOS analog integrated circuits and systems, current-mode signal processing, communication systems, and electronic instrumentation. He has authored or coauthored 29 research papers in various international journals. He has acted/has been acting as a Reviewer for several IEEE journals (USA), the Institution of Electrical Engineers (U.K.), as well as a number of other international journals. His biography is listed in the 2005 edition of Marquis’ Who’s Who (New Jersey).

2021

Raj Senani received the B.Sc. degree from the University of Lucknow, Lucknow, India, the B.Sc.Eng. degree from the Harcourt Butler Technological Institute, Kanpur, India, the M.E.(Hons.) degree from the Motilal Nehru National Institute of Technology (MNNIT), Allahabad, India, and the Ph.D. degree in electrical engineering from the University of Allahabad. He was a Lecturer (1975–1986) and Reader (1987–1988) with the Department of Electrical Engineering, MNNIT. He joined the Department of Electronics and Communication Engineering, Delhi Institute of Technology, Delhi, India, in 1988 as an Assistant Professor. He became a Professor in 1990. Since then, he has served as Head, Department of Electronics and Communication Engineering (1990–1993, 1997–1998), Head, Applied Sciences (1993–1996), Head, Manufacturing Processes and Automation Engineering (1996–1998), Dean, Research (1993–1996), Dean, Academic (1996–1997), Dean, Administration (1997–1999), Dean, Post Graduate Studies (1997–2001), and Director, Netaji Subhas Institute of Technology (NSIT), New Delhi, India (June 1996 to September 1996, February 1997 to June 1997, and May 2003 to January 2004). He has been the Head of the Division of Electronics and Communication Engineering, NSIT, since 2000. Dr. Senani served as an Honorary Editor of the Journal of Research of the Institution of Electronics and Telecommunication Engineers (IETE), India, during 1990–1995 in the area of circuits and systems and has been a Member of the Editorial Board of the IETE Journal on Education since 1995. He has been functioning as the Editorial Reviewer for a number of IEEE (USA), Institution of Electrical Engineers (U.K.), and other international journals. He has been serving as an Associate Editor for the Journal on Circuits, Systems and Signal Processing since 2003. His teaching and research interests are in the areas of bipolar and CMOS analog integrated circuits, current-mode signal processing, electronic instrumentation, chaotic nonlinear circuits, and translinear circuits. He has authored or coauthored over 100 research papers in various international journals. He is listed in several editions of the Marquis’ Who’s Who series (New Jersey), several biographical publications of the International Biographical Centre (Cambridge), U.K. and a number of other international biographical directories.