IEEE1394, HDMI, DVI, PCI-Express, and S-ATA [1]. Since channels should maintain a frequency band covering maximum bit rates, the frequency range of ...
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Eye-Pattern Design for High-Speed Differential Links Using Extended Passive Equalization Ki Jin Han, Student Member, IEEE, Hayato Takeuchi, and Madhavan Swaminathan, Fellow, IEEE
Abstract—The performance of current high-speed consumer electronic systems is often compromised by degradation caused by distortion in eye patterns. This paper proposes a systematic method that uses the voltage transfer function for arbitrary source and load terminations to improve the eye patterns of high-speed differential links with passive components that minimize distortion. This approach is cost-effective since it only utilizes commercially available surface-mount components. The methodology has been validated by measurements in this paper. Index Terms—Channel response, differential -parameters, eye pattern, high-speed differential links, impedance mismatch, passive equalization, signal integrity, termination.
I. INTRODUCTION
D
IGITAL interface channels implemented in commercial electronics need to conform to the current trend of higher data rates. A maximum data rate from several hundred Mbps to a few Gbps is supported in advanced technologies such as USB, IEEE1394, HDMI, DVI, PCI-Express, and S-ATA [1]. Since channels should maintain a frequency band covering maximum bit rates, the frequency range of interface channels should be extended to the microwave frequency range. At high frequencies, the digital interface channels are faced with nonideal behaviors of passive components and transmission lines. High-frequency loss and impedance mismatch cause delays due to increase of rise/fall time and undesirable signal distortion, respectively. As a solution for these high-frequency problems, various active and passive equalization techniques have been developed [2] based on the strategy of emphasizing high-frequency components or de-emphasizing low-frequency components of digital signals. In the existing equalization techniques, recovering retarded rise/fall time is the main focus, but improving signal distortion due to mismatch has not been considered. This paper proposes a systematic approach for suppressing signal distortions and improving eye patterns in high-speed links. This new approach differs from the conventional concept in the fact that the purpose of equalization is not simply to preserve the rise/fall time but to improve the general quality of eye patterns. Therefore, this is an extended concept of
Manuscript received August 3, 2007; revised October 2, 2007. This work was recommended for publication by Associate Editor M. Cases upon evaluation of the reviewers comments. K. J. Han and M. Swaminathan are with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, GA 30332 USA. H. Takeuchi is with the Advanced EMC Design Technology Section, MONOZUKURI Technology Center, Sony Corporation, Tokyo 141-0001, Japan. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TADVP.2008.915849
Fig. 1. High-speed differential link model. (a) Hardware configuration. (b) Schematic model.
equalization, where the design objective is shaping an original transfer function into a number of desirable responses such as linear or constant attenuation. Since distortions due to driver mismatch are controlled by the use of purely passive components, the extended passive equalization approach is simple and cost-effective for practical high-speed design purposes. This paper is organized as follows. In Section II, technical issues concerning high-speed differential links, the main focus of this paper, are discussed. An extended concept of passive equalization and a design methodology with a simple passive network are presented in Section III. The modeling and simulation procedures, which are critical parts of the equalizer design and evaluation, are explained in Section IV. In Section V, the design of test vehicles and measurement results for verifying the equalization approach are presented. Finally, conclusions are given in Section VI. II. HIGH-SPEED DIFFERENTIAL LINK Fig. 1 shows a high-speed interface applied in consumer electronics, and its differential channel model. Driver LSI is a current source based on the open collector configuration, and the package of the driver chip is 14 mm 14 mm thin-profile quad flat pack (TQFP). An edge-coupled coplanar waveguide with ground (CPWG) [3] is used for the transmission line structure on the PCB. A common-mode filter (CMF) and a voltage dependent resistor (VDR) protect the signal against common-mode noise and electrostatic discharge (ESD), respectively. Traces on flexible cables, fabricated on the organic substrate, are similar to the edge-coupled CPWG type. Because the selection of the
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Fig. 4. General 2-port channel model with a current source. Fig. 2. Typical mismatch distortions in eye-patterns: (a) “ringing” and (b) Shoulder.
Fig. 3. Extended concept of passive equalization.
driver package, connectors, and other components are based on cost considerations, their performance at high frequencies cannot be guaranteed. A main issue in the differential link is signal distortions that originated from the impedance mismatch between various components and lines. Even though the characteristic impedances of transmission lines are well designed to meet standards, additional components used for satisfying electromagnetic compatibility (EMC) and ESD regulations can generate undesired reflections. The effect of the mismatch appears as multiple distortions in time-domain measurements. One form of distortion is “ringing” [Fig. 2(a)], or oscillation, which causes the closure of an eye pattern and generates an excessive level of overshoot in the waveform. Another form of distortion is “shoulder” distortion [Fig. 2(b)], which degrades transient characteristics by retarding rise/fall times. Current mode drivers based on the open collector configuration bear these signal distortion or reflection problems. A serious mismatch problem often arises from components such as unterminated drivers when they are combined with a number of other parasitic elements. A common solution to the distortion problem is terminating the source as well as the load of the channel [4]. However, commercially available drivers for interfaces such as HDMI and DVI do not use the source termination to take advantage of low-power consumption. Adding source termination, with the same input power, reduces the voltage swing by half, so the input power shouldbe doubled to maintain the voltage swing. This power-saving versus reflection issue is not unusual especially in low-cost, low-power designs for consumer electronics.
low-pass filters, so that it is flat and lossless over the desired frequency band. The modified channel provides the opening of the eye pattern by recovering degraded rise/fall time. Since equalizers have to provide frequency-dependent amplification in this design concept, even passive techniques need to accompany active components [5], [6]. However, inclusion of additional active schemes complicates the hardware, which is not favorable for cost-effective design. In addition, distortion due to impedance mismatch is not a major issue in conventional techniques. For channels with considerable mismatch, however, eliminating distortions in eye patterns is more important than recovering the degraded rise/fall time, which is caused by channel dielectric and conductor loss. High-frequency losses such as skin effect and dielectric loss are emphasized only when the length of the channel is electrically long, whereas the mismatch problems are always dangerous for high-speed channels regardless of their electrical length. Thus, the main focus of the extended passive equalization in this paper is to minimize the mismatch distortion by using purely passive components. In the extended equalization, a desired frequency response is defined according to specific design objectives, and then the optimal equalizer is found to shape the original channel transfer function into the desired response. This new approach mainly differs from the existing equalization methods in that the desired responses can be something other than lossless transmission. Monotonic attenuation and constant attenuation over a design frequency band are other possible examples of the desired transfer function. A benefit of using these lossy responses is that no active component is required for realizing them. Fig. 3 illustrates the extended concept, where an equalizer reduced distortions in eye patterns by adjusting the original resonant response to a lossy objective response. B. Design Method for the Passive Equalizer Network When any channel under design is driven by a current source with imperfect matching, as shown in Fig. 4, the following general voltage transfer function with arbitrary source and load impedance should be considered [8]
III. EXTENDED PASSIVE EQUALIZATION (1)
A. Concept of the Extended Passive Equalization The conventional definition of equalization is to modify the frequency response, which is usually similar to the response of
where and are the reflection coefficients for the source and load impedance, respectively. When both the source and
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Fig. 5. Channel model with a passive equalizer included.
load terminations are applied , the gen. Thus, eral form of the transfer function is reduced to , and resonant the channel design is simplified to design of response does not occur unless the channel contains any active component. However, many practical designs including the differential link in Fig. 1 employ only load terminations without source terminations since source terminations require excessive power consumption. The transfer function in case of the un-terminated is as follows: source
Fig. 6. Line-to-line equalizer and its trace of the impedance values on the complex plane. (Direction of arrows indicates an increase in frequency from zero to infinity.).
(2) The transfer function in (2) indicates that the response is affected not only by the transmission coefficient of the channel but also by the reflection coefficient at the input end. A possible undesirable case is that the transfer function becomes unstable is close to one. In the channel model of Fig. 1 for when example, package parasitic elements and other discrete components (VDR, CMF) degrade the reflection coefficient. Although it is known that using only the end termination is sufficient for the design of signal transmission without reflections [9], this is negligibly remedy is valid under the assumption that small over the desired frequency band. In physical terms, the poor input matching condition generates reflections that would make the entire system unstable unless the source is terminated. A basic channel configuration for the proposed design method is shown in Fig. 5. The channel is divided into a driver network and a transmission network, with the passive equalization network inserted between the two. The driver network includes parasitic elements in the driver package and a short section of differential traces connected to the driver package. The signal transmission network is the remaining part of the channel. The main reason why the entire channel is separated into these two parts is that the channel without the driver termination can be improved most effectively by adding the passive network at the end of the driver network. Moreover, it is difficult to control or modify the driver network in practical channel designs. The scattering parameters for each of the block are shown in Fig. 5.
Fig. 7. Procedure of eye-pattern simulation.
The channel transfer function in (2) can be expressed by using the parameters in Fig. 5, as shown in (3) at the bottom of the page , and are transmission where parameters obtained from the conversion from [10]. By over the frequency band, selecting the proper values of can be adjusted to obtain the desirable transfer function. For the purpose of eliminating ringing distortions, a desirable channel should not have resonant peaks at the design frequencies. Since the passive equalizer does not generate any amplification, a monotonically or linearly attenuating function over the frequency range is a possible solution. If a slight reduction in the voltage amplitude is allowable, a constant lossy response can be another solution.
(3)
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Fig. 8. Simulation and measurement data for differential channels. (a) Differential S -parameters without the package model. (b) Differential S -parameters with the package model. (c) Transfer function (with the package model).
For a selected transfer function be expressed as follows:
, the design objective can
(4) For most practical applications, it is not always possible to find the optimal network that satisfies the above condition over
the desired frequency band. The general solution in (3) cannot be uniquely determined. Furthermore, it is not always guaranteed that the solution is realizable using a simple passive circuit. A possible approach is to find the best component values from a predetermined circuit topology. For some special cases, we can convert the above objective in (4) to a more explicit expression with which impedance values in the equalizer can be easily controlled.
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Fig. 9. Measurement scheme for the design of the passive equalizer network.
Fig. 10. Topology of passive equalizers on the differential link. (a) Parallel type equalizer. (b) Series type equalizer.
The line-to-line equalizer, a combination of passive elements connected along the channel in series, is a simple equalizer topology used in this paper (Fig. 6). The employment of the line-to-line configuration originates from the design practice of including a damping resistor for reducing ringing distortions in eye patterns. Together with a single damping resistor, the line-to-line equalizer includes any other reactive components. If the impedance of the equalizer is , the transmission matrix of any line-to-line network can be expressed as in (5), and the transfer function (3) can be simplified as follows:
(5)
(6)
Fig. 11. Pad layouts for equalizers and probes on test vehicles. (Dashed rectangles are probing pads and dotted rectangles are component pads.) (a) Pads for parallel type equalizers. (b) Pads for series type equalizers.
on the original concept of the design practice, a damping resistance is always included. Thus, the possible types of circuit in series, in parconfigurations are categorized into allel, in series, and in parallel. Typical impedance traces of the four networks are plotted in Fig. 6. is the desired transfer function, the above Assuming that formulas, combined with (4), provide the following equation of circle for the desired impedance:
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(7)
HAN et al.: EYE-PATTERN DESIGN FOR HIGH-SPEED DIFFERENTIAL LINKS USING EXTENDED PASSIVE EQUALIZATION
Fig. 12. Design and optimization examples with a passive equalization chart. (a) R-L in the series equalizer, R = 14:9 ohm, L = 2:7 nH, = 3 R-L in the parallel equalizer, R = 27:7 ohm, L = 2:78 nH, = 3:5 10 .
2
where
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2 10
. (b)
response by using the line-to-line equalizer. Similar to the response of lossy transmission lines, the attenuating response can be expressed as follows: (8)
and
The equalizer design is a procedure of locating the equalizer impedance traces (Fig. 6) so that it is as close to any point on the circle of (7) as possible. This procedure is similar to the design of microwave matching circuits on the Smith chart [11]. Since both the radius and the center of the circle vary with frequency, difficulties in the design arise. As the broadband design is essential for the equalizer to be close to the desired impedance circles for all frequencies, extending the design frequency range will cause even more difficulty. From some investigations of -parameter data obtained from the interface channels considered in this paper, a linearly attenuating frequency characteristic is found as a desirable objective
The attenuation coefficient in (8) is an important design parameter. A larger value of the attenuation coefficient will sufficiently reduce the ringing distortion, but it may cause significant closure of the eye pattern. On the other hand, a smaller attenuation coefficient will preserve the rise/fall time, but it may not sufficiently suppress distortion. In the equalization design formula, the attenuation coefficient controls the radius of the desired impedance circle in (7); however, the center of the circle is independent of the attenuation coefficient. IV. MODELING OF DIFFERENTIAL CHANNEL RESPONSE As summarized in Fig. 7, the output eye-pattern simulation requires the spectrum of the pseudorandom bit sequence (PRBS) and differential -parameters. PRBS, which is generally expressed as the Fourier series in the frequency domain, has spectral contents similar to those of periodic trapezoidal
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Fig. 13. Average deviations from the desired impedances for different attenuation coefficients. (a) R-L in parallel. (b) R-L in series.
waves [12]. The data of -parameters can be obtained from measurement or simulation, and then their accuracy can be validated by cross-checking. The output signal spectrum results from multiplying the transfer function from the differential -parameters by the PRBS spectrum in the frequency domain. A time-domain eye pattern is finally obtained by using inverse Fourier transform of the output spectrum. Fig. 17 of the next section will show some eye-pattern simulation results for the test vehicles. -parameter data of edge-coupled CPWG sections, flexible cables, and component pads are obtained from Sonnet, a 3-D electromagnetic field simulator [13]. In the case of the package parasitic elements such as pins and wires in the driver network, direct measurement is difficult since it is virtually impossible to apply probing ports inside of the driver package. Thus, the generation of an accurate package model is important in any 3-D electromagnetic simulation set up to reduce the design effort. Differential -parameters of channel components or the whole system are calculated by using the extended conversion formula, which is similar to [14], [15], for each channel
(9) where is a single-ended -parameter, and
-parameter,
is a differential
Simulation and measurement data of differential -parameters are plotted in Fig. 8. For checking the validity of the simulation, simulated -parameters are compared with measured -parameters in Fig. 8(a), when the package parasitic model is not included. Design of the passive equalization requires the channel data including the package model, shown in Fig. 8(b). Transfer functions in Fig. 8(c) are also obtained from the differential -parameters of the entire channel model
with the package. For all cases, the simulation data and the measured data are well matched. In both the simulation and measurement, predicting the resonance peak, shown at close to 4 GHz, is particularly important. The resonance peak can be estimated by considering that resois close to zero, as shown in (2). nance occurs when If we consider the transfer function as a feedback system rein the numerator and in the denominator can sponse, be regarded as the open loop gain and the feedback gain, respectively. At a frequency in which resonance peaks occur, has zero in phase with a magnitude of close to one. This finding supports the fact that system instability occurs at frequencies at which the magnitude and the phase of the feedback gain are one and 180 , respectively. V. DESIGN OF TEST VEHICLES AND MEASUREMENT RESULTS The first step of verifying the proposed method is the design of test vehicles for measuring -parameters. The test vehicles were composed of differential transmission lines on PCB, flexible cables, and other discrete components, as described in Fig. 1. To validate all possible combinations of the line-to-line equalizer, we designed two different types of layout for both series and parallel combinations shown in Fig. 10. In addition, the data of the channel -parameters had to be obtained separately for lines in the driver network and the transmission network, as illustrated in Fig. 9. In total, four types of test vehicles were fabricated with probing pads added for the ports of the equalizer inputs and outputs. Fig. 11 shows the layout parts for equalizers in the test vehicles for measuring the driver network and the transmission network. For accurate measurement and design, probing pads on these test vehicles were designed to ensure the stability of the attachment of the probes onto the pads. We also designed an additional adaptor between the interface connector and the probe pads for applying probes of a network analyzer on the terminal of the differential link. Four-port -parameters were measured by using the Agilent E8363B network analyzer with Cascade GSSG-500 probes, and then converted into two-port differential -parameters. The first two plots in Fig. 14 show the differential -parameters and the transfer functions of channels when passive equalizers are not
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Fig. 14. Differential S -parameters for channels with and without passive equalizers. (Left: R-L in series equalizer. Right: R-L in parallel equalizer).
mounted. Resonance peaks that occur at near 4 GHz in these combined transfer functions indicate possible ringing distortions in time-domain signal measurements. These resonance effects should be removed by applying proper passive equalizers. Byusingthemeasuredorsimulateddata,optimalcombinations of passive equalizer elements were found to shape the resonant response into a linearly attenuating response. The passive equalization chart in Fig. 12, drawn from the -parameter
data of the test vehicles, enables the proper type of passive equalizer networks to be chosen. Any combination of passive components can be selected if its impedance value is sufficiently close to points of the circles that are drawn on the complex impedance plane for each design frequency from (7). Up to 3 GHz, traces of the - series (straight arrow) as well as the - parallel (curved arrow) network catch up with the circles in Fig. 12 more closely than those of the - series
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Fig. 15. Voltage transfer functions for channels with and without passive equalizers. (Left: Straight lines are objective functions).
or the - parallel network. The optimal combination of and values in Fig. 12 is found by minimizing errors based on the Nelder-Mead algorithm [16] provided in the MATLAB function library. Here, the meaning of an “optimal combination” is that the selected network is closest to the given attenuation profile. Design specifications determine available values of attenuation constants in (8). A smaller attenuation
R -L
in series equalizer. Right:
R -L
in parallel equalizer.
constant is a better choice in practice, but a good design with such a constant is more difficult to find. In summary, Fig. 13 shows the average deviations of the optimal network impedance from the desired impedance for both - parallel and - series networks, respectively. In Table I, optimal component values for the series and parallel equalizers are listed for different attenuation coefficients. The component
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Fig. 16. Effect of the components on the passive equalizer (R = 56 ohm, L = 1:8 nH in parallel). (a) Transfer functions and (b) eye-patterns for a channel with the R-L equalizer and channels with only one component mounted.
TABLE I SELECTED OPTIMAL EQUALIZER COMPONENT ACCORDING TO DIFFERENT ATTENUATION COEFFICIENTS (COMPONENT VALUES IN PARENTHESES ARE AVAILABLE)
values are not actually the same as the optimal values, but they are adjusted so that they meet the available standard values for surface-mount components. Combined -parameters and transfer functions with a passive equalizer mounted are plotted in Figs. 14 and 15, respectively. In the frequency range from zero to 3 GHz, - series and parallel equalizers make the overall channel responses close to the desired attenuation profile. Therefore, resonant peaks are removed. A thorough understanding of the roles of the components in the passive equalizer is important. For one, resistors in the designed equalizers suppress ringing distortions by applying a damping effect to the channel. If eye opening is not a significant design issue, the inclusion of only resistances will solve the ringing distortion problem. However, Fig. 16 shows that this simple approach may result in an over-elongated rise/fall time. In the design of the equalizer, inductors compensate for the high attenuation effects due to resistance, particularly at low frequencies. In other words, as the frequency increases, the contribution of inductors to impedance of the equalizer also increases. Thus, by including inductances, shoulder distortions can be reduced in eye patterns. In summary, the component values of the passive equalizer are optimized by balancing the ringing distortion due to high-frequency resonance with the shoulder distortion due to low-frequency attenuation.
Finally, eye patterns were estimated from the above design results and measured in the time domain, shown in Fig. 17. The input digital signal used in the simulation was 451-bit PRBS, composed of trapezoidal functions with a frequency of 750 Mbps and a rise/fall time of 0.13 ns. Ringing distortions in the original channel were almost eliminated for all measured eye patterns. As expected, rise/fall time increases for larger values of the attenuation constant, but the measured eye patterns sometimes show more attenuations than estimated eye patterns. Since the reason for the difference comes from variations between the package simulation model and the actual package parasitic elements, the accuracy of the design can be improved by adjusting the package model data.
VI. CONCLUSION This research proposed a design method of suppressing distortions in the eye patterns of high-speed differential links and focused mainly on mismatched differential channels that do not have source termination. An analysis of the general channel model found that the main reason for distortions in eye patterns was resonances due to the reflection between the channel input part and the driver part without termination. With the addition of simple passive components, extended passive equalization modified the channel transfer function and reduced distortion. The proposed method was verified with test vehicles designed by using differential -parameter data based on the simulation or measurement. The measurements by the test vehicles validated the assumption that eye patterns can be improved by applying the designed equalizers. Since the proposed method is cost-effective and systematic, it can be applied to the design of high-speed interfaces for low-cost consumer electronics. Furthermore, the design approach is generally applicable to any higher bit-rate system if the approach is accompanied with other proper practices of high-speed design.
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Fig. 17. Estimated and measured output eye patterns for channels with and without passive equalizers. (Left: R-L in a series equalizer. Right: R-L in a parallel equalizer.)
ACKNOWLEDGMENT The authors would like to thank E. Engin of the Georgia Institute of Technology for valuable theoretical advice and K. Yasuda of the Sony Corporation for technical advice about design and measurements. REFERENCES [1] E. Bogatin, Signal Integrity—Simplified. Upper Saddle River, NJ: Prentice Hall PTR, 2004, p. 21. [2] J. Liu and X. Lin, “Equalization in high-speed communication systems,” IEEE Circuits Syst Mag, vol. 4, no. 2, pp. 4–17, 2004. [3] B. Wadell, Transmission Line Design Handbook. Norwell, MA: Artech House, 1991.
[4] A. Singh, W. Massoth, U. Fields, S. Y. Youn, and M. Parten, “Signal integrity improvement in the TMDS link at UXGA,” in 45th Midwest Symp. Circuits Syst., MWSCAS 2002, Aug. 4–7, 2002, vol. 2, pp. II-429–II-432. [5] R. Sun, J. Park, F. O’Mahony, and C. P. Yue, “A low-power, 20-gb/s continuous-time adaptive passive equalizer,” in IEEE Int. Symp. Circuits Systems, ISCAS 2005, Kobe, Japan, May 23–26, 2005, vol. 2, pp. 920–923. [6] E. Sayre, M. Baxter, and E. Sayre III, “Optimized passive infiniband cable equalizer designs,” presented at the DesignCon, Santa Clara, CA, Jan. 27–30, 2003. [7] S. Ahn, S. Baek, J. Lee, and J. Kim, “Compensation of ESD and device input capacitance by using embedded inductor on PCB substrate for 3 gbps serdes applications,” in IEEE Int. Symp. Electromagn. Compatibility, EMC 2004, Silicon Valley, CA, Aug. 2004, vol. 2, pp. 499–504.
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[8] K. J. Han, H. Takeuchi, E. Engin, and M. Swaminathan, “Eye-pattern improvement for design of high-speed differential links using passive equalization,” in IEEE 15th Topical Meeting Electr. Performance Electron. Packag., Scottsdale, AZ, Oct. 23–25, 2006, pp. 241–244. [9] H. Johnson and M. Graham, High Speed Signal Propagation: Advanced Black Magic. Englewood Cliffs, NJ: Prentice-Hall, 2003. [10] D. M. Pozar, Microwave Engineering. New York: Wiley, 2005. [11] G. Gonzalez, Microwave Transistor Amplifiers: Analysis and Design. Englewood Cliffs, NJ: Prentice Hall, 1997. [12] H. W. Ott, Noise Reduction Techniques in Electronic Systems. New York: Wiley, 1988, pp. 302–304. [13] Sonnet 8.53. Sonnet Software, Inc., North Syracuse, NY. [14] D. E. Bockelman and W. R. Eisenstadt, “Combined differential-mode and common-mode scattering parameters: Theory and simulation,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 7, pp. 1530–1539, Jul. 1995. [15] S. Baek, S. Ahn, J. Park, J. Kim, J. Kim, and J.-H. Cho, “Accurate high frequency lossy model of differential signal line including mode-conversion and common-mode propagation effect,” in Int. Electromagn. Compat. Symp., Aug. 2004, vol. 2, pp. 562–566. [16] W. H. Press, S. A. Teukolsky, W. T. Vetterling, and B. P. Flannery, Numerical Recipes in C—The Art of Scientific Computing, 2nd ed. Cambridge, U.K.: Cambridge Univ. Press, 1995, pp. 408–410.
Ki Jin Han (S’06) received the B.S. degree (summa cum laude) and the M.S. degree, both in electrical engineering, from Seoul National University, Seoul, Korea, in 1998 and 2000, respectively. He is currently working toward the Ph.D. degree in the School of Electrical and Computer Engineering (ECE) at the Georgia Institute of Technology, Atlanta. From 2000 to 2005, he was with the System Research and Development Laboratory of LG Precision (currently named LIG Nex1), where he was involved in the development of RF/microwave transceiver and antenna system for airborne and naval radars. He is currently a graduate research assistant in the School of Electrical and Computer Engineering at the Georgia Institute of Technology, Atlanta. His current research interests include computational electromagnetics, signal/power integrity for high-speed digital design, and the electromagnetic modeling of electronic packaging and interconnects.
Hayato Takeuchi received the B.S. and M.S. degrees in electrical engineering from Hokkaido University, Japan, in 1993 and 1995, respectively. He joined the Assembly Technology Development Division of Sony Corporation, in 1995, as a High Performance LSI Package Characterization Engineer. Currently, he also serves as a Chief EMC and Signal Integrity Researcher for LSIs. In 2006, he was a Visiting Industrial Engineer of Packaging Research Center (PRC) at the Georgia Institute of Technology, Atlanta, for one year. His research interest is development of methods for improved high-speed differential transmission lines in packages and boards.
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Madhavan Swaminathan (M’95–SM’98–F’06) received the B.E. degree in electronics and communication from the University of Madras, Chennai, India, and the M.S. and Ph.D. degrees in electrical engineering from Syracuse University, Syracuse, NY. He is currently the Joseph M. Pettit Professor in Electronics in the School of Electrical and Computer Engineering, Georgia Institute of Technology (Georgia Tech), Atlanta, and the Deputy Director of the Packaging Research Center, Georgia Tech. He is the co-founder of Jacket Micro Devices, a company specializing in integrated devices and modules for wireless applications where he serves as the Chief Scientist. Prior to joining Georgia Tech, he was with the Advanced Packaging Laboratory at IBM working on packaging for super computers. He has over 300 publications in refereed journals and conferences, has coauthored three book chapters, has 15 issued patents and has several patents pending. While at IBM, he reached the second invention plateau. He is also the author of the book on Power Integrity Modeling and Design for Semiconductors and Systems, (Prentice Hall, 2007) and coeditor of the book on Introduction to SOC, SIP and SOP, (McGraw Hill, to appear in 2008). His research interests are in mixed signal microsystem and nano-system integration. Dr. Swaminathan is the recipient of the 2002 Outstanding Graduate Research Advisor Award from the School of Electrical and Computer Engineering, Georgia Tech and the 2003 Outstanding Faculty Leadership Award for the mentoring of graduate research assistants from Georgia Tech. He is also the recipient of the 2003 Presidential Special Recognition Award from IEEE CPMT Society for his leadership of TC-12 and the IBM Faculty Award in 2004 and 2005. He has also served as the coauthor and advisor for a number of outstanding student paper awards at EPEP’00, EPEP’02, EPEP’03, EPEP’04, ECTC’98, APMC’05, and the 1997 IMAPS Education Award. He is the recipient of the Shri. Mukhopadyay best paper award at the International Conference on Electromagnetic Interference and Compatibility (INCEMIC), Chennai, India, 2003, the 2004 best paper award in the IEEE TRANSACTIONS ON ADVANCED PACKAGING, the 2004 commendable paper award in the IEEE TRANSACTIONS ON ADVANCED PACKAGING and the best poster paper award at ECTC’04 and ’06. In 2007, he and his students were recognized for their research by the Technical Excellence Award given by Semiconductor Research Corporation (SRC) and Global Research Corporation (GRC). He served as the Co-Chair for the 1998 and 1999 IEEE Topical Meeting on Electrical Performance of Electronic Packaging (EPEP), served as the Technical and General Chair for the IMAPS Next Generation IC & Package Design Workshop, serves as the Chair of TC-12, the Technical Committee on Electrical Design, Modeling and Simulation within the IEEE CPMT society and was the Co-Chair for the 2001 IEEE Future Directions in IC and Package Design Workshop. He is the cofounder of the IMAPS Next Generation IC & Package Design Workshop and the IEEE Future Directions in IC and Package Design Workshop. He also serves on the technical program committees of DAC, EPEP, Signal Propagation on Interconnects (SPI) workshop, Solid State Devices and Materials Conference (SSDM), Electronic Components and Technology Conference (ECTC), and International Symposium on Quality Electronic Design (ISQED). He has been a guest editor for the IEEE TRANSACTIONS ON ADVANCED PACKAGING and IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. He was the Associate Editor of the IEEE TRANSACTIONS ON COMPONENTS AND PACKAGING TECHNOLOGIES.
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