New Microstrip Bandpass Filter Configuration with Improved Attenuation of One Adjacent Channel Nicolae Militaru, Teodor Petrescu, George Lojewski
Marian Gabriel Banciu
Department of Telecommunications University POLITEHNICA of Bucharest Bucharest, Romania
[email protected]
Group of Physics of High Frequency Materials and Devices National Institute of Materials Physics Magurele – Ilfov, Romania
[email protected]
Abstract— A new topology for fourth-order microstrip bandpass filters is investigated in this paper. Compared to the classical quadruplet, the proposed new topology offers an extra coupling between two resonators, allowing this way the design of fourth order filters with two attenuation poles situated on the same side of the passband. A strongly asymmetric transfer characteristic can be obtained by a proper choice of the positions of the two attenuation poles, providing a high attenuation not only at a single frequency, but in a large adjacent frequency band, as needed in some diplexer applications. Keywords: microstrip filters; diplexers; coupling matrix;
I.
INTRODUCTION
Recent advances in the theory of filter synthesis and in the design techniques for microwave filters based on electromagnetic field simulation and optimization have opened new interesting ways for improving the performances of these devices [1], [2]. It is possible now to investigate filters with new topologies, with special characteristics, impossible to be designed in classical structures. A fourth-order bandpass filter having the classical symmetric quadruplet structure [3] generates two symmetric transmission zeros on the real-frequency axis, one on each side of the passband. However, the rejection of one of the two adjacent channels can be considerably improved if both available zeros of this fourth-order filter are placed on the same side of the passband. In order to obtain such a strongly asymmetric response, an extra-coupling is needed (3 - 4 coupling, in Fig. 1.a). 1
0
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5 3
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Figure 1. The coupling topology of a fourth-order band-pass filter with a strongly asymmetric response (a) and its basic planar geometry (b).
A filter with these characteristics cannot be realized in the usual quadruplet configuration, but this extra-coupling can be obtained, however, by considering a novel quadruplet topology
proposed in Fig. 1.b. In order to accommodate the necessary couplings, this new microstrip structure is composed of resonators with different shapes. The elements no. 1 and no. 2 are classical open-ended half-wavelength resonators, while the elements no. 3 and no. 4 are resonators of hairpin type. The 50 Ω feed-lines (0 and 5) are coupled to the corresponding resonators (1 and 4, respectively). II.
DESIGN PROCEDURE AND SIMULATION RESULTS
The design methods presented in the literature [4], [5] offer the possibility to synthesize normalized coupling matrices M for quasi-elliptic filters of different orders, starting from an imposed constant ripple in the passband and from specified positions of all transmission zeros. The individual resonance frequencies of the resonators in the designed filter and the necessary couplings between its elements can then be found from M through some simple de-normalization procedures. Based on the novel topology in Fig. 1, two microstrip bandpass filters were designed, simulated, fabricated and tested. Both filters have the next common requirements: center frequency f0 = 3.5 GHz, bandwidth B = 175 MHz (fractional bandwidth w = 5 %, with cut-off frequencies f1 = 3.4125 GHz and f2 = 3.5875 GHz), a Chebyshev response with a constant ripple R = 0.46 dB (corresponding to an in-band reflection loss LR = 10 dB), and 50 Ω terminations. First filter has two attenuation poles in the upper adjacent stopband at normalized values fz1 = 2 (3.675 GHz) and fz2 = 3 (3.7625 GHz), while the second has two attenuation poles in its lower adjacent stopband at normalized values fz1 = –2 (3.325 GHz) and fz2 = –3 (3.2375 GHz). With a home-made application based on the methods presented in [4], [5], a synthesis procedure starting from these requirements leads, for the first filter, to the following normalized coupling matrix M1: 0.78529 0 0 0 0 ⎤ ⎡ 0 ⎢0.78529 0.03377 0 0.71770 0.05591 0 ⎥⎥ − ⎢ ⎢ 0 −0.52370 0.48362 −0.62656 0 0 ⎥ M1 = ⎢ ⎥ 0.71770 0.49362 0.01663 −0.35002 0 ⎥ ⎢ 0 ⎢ 0 −0.05591 −0.62656 −0.35002 0.03377 0.78529⎥ ⎢ ⎥ 0 0 0 0 0.78529 0 ⎦ ⎣
The second filter has a similar coupling matrix M2, with opposite signs of all terms located on the main diagonal and with an opposite sign of M34:
modified only in areas which do not contribute significantly to the coupling. These considerations are essential for the convergence of the correction algorithm.
0.78529 0 0 0 0 ⎤ ⎡ 0 ⎢0.78529 −0.03377 ⎥ 0 0.71770 0.05591 0 − ⎢ ⎥ ⎢ 0 0 0.52370 0.48362 −0.62656 0 ⎥ M2 = ⎢ ⎥ 0.71770 0.49362 −0.01663 0.35002 0 ⎥ ⎢ 0 ⎢ 0 −0.05591 −0.62656 0.35002 −0.03377 0.78529⎥ ⎢ ⎥ 0 0 0 0.78529 0 ⎦ ⎣ 0
Finally, the remaining errors, caused by all influences that were not completely removed in the previous steps, can be even better corrected by using a new and ingenious computerbased optimization procedure that combines the accuracy of electromagnetic-field simulation with the speed and convenience of linear circuit optimization [1], [2]. In this sense a number of extra-ports are attached to the layout resonators, offering the possibility to capture, through electromagnetic field simulation, the actual couplings in the microwave layout, with their actual frequency dependence. Then some external reactances can be attached to these extra-ports and used as variables in an automatic optimization procedure, within a linear circuit simulation, the microwave layout being treated now as a constant building block. The values of the external elements found through optimization represent, in fact, the errors in the resonance frequencies and in the couplings. Once these errors identified and quantified, they can be corrected going back to the EM-field simulator and resuming the analysis on individual resonances and on couplings.
The synthesis procedure was tested and validated by considering lumped-elements circuit models (LC resonators and ideal inverters, as coupling elements). As expected, the responses of these models meet near perfectly all the design requirements, validating this way the synthesis procedure. Their simulated responses can then be used as terms of comparison (Fig. 4 and Fig. 5). Based on the parameters of the filters resulted from M1 and M2 through de-normalization, the two microstrip bandpass filters were designed on a Rogers3003 dielectric laminate. This substrate has a thickness of 0.508 mm, a dielectric constant εr = 3, a dielectric loss tangent tanδ = 0.0013 and copper metallic sheets with thicknesses of 0.035 mm. The design of the first filter starts by considering a layout with a configuration similar to that shown in Fig. 1.b. The dimensions of the two types of resonators, the straight one and the folded one, were accurately found by electromagnetic field simulation [6]. A similar procedure was applied to find the position of the couplings of the two feed-lines with the input and output resonators, 1 and 4, leading to the required external Q, as well as for the relative position of each pair of coupled resonators, leading to the required coupling coefficient, k. Although practically it is hardly possible to get a planar layout corresponding exactly, simultaneously, to all these requirements, an approximate solution can however be found. In this sense, this first layout can be considered as a first iteration in the design procedure. In order to improve the design, it is then necessary to identify and to correct the errors in the individual resonance frequencies and in the couplings. Indeed, each element and each pair of elements in the filter's structure – designed independently – is, in fact, also slightly influenced by the presence of the other neighboring elements. When the whole structure of the filter is assembled, its parameters do not correspond any more exactly to the design intentions, so that some adjustments are needed. In the attempt to correct the parameters, a special care must be taken to the sequence of the adjustments. The adjustments should begin with a fine correction of the couplings of the feed-lines 0 and 5 with the end resonators 1 and 4, because these couplings are not significantly affected by the presence of neighboring elements. Next, the individual resonances of the end resonators, usually influenced by the couplings with 50 Ω feed-lines, can be corrected. The couplings between different pairs of resonators should be adjusted before the fine-tuning of their individual resonance frequencies, because the couplings affect the individual resonances, while the resonances can be adjusted without any noticeably effect on the couplings, if the layout is
The layout of the first filter, after the adjustments on its elements, is shown in Fig. 2. It can be noticed that simple and regular geometries of the resonators from Fig. 1.b were slightly modified, during the fine tuning procedure of resonance frequencies and/or of couplings.
Figure 2. Layout of the first filter, after optimization.
The second filter, with attenuation poles located below the passband, was designed in a similar way. However, the coupling between resonators 3 and 4 must be now of magnetic type, to accommodate the changed sign of the corresponding coupling coefficient M34 in M2. To avoid the type-II mixed coupling [3] between resonators 1 and 2 with 3 and 4, the shapes of the resonators 1 and 2 in this filter have been changed. The layout of the second filter, after the adjustment of all its elements, is presented in Fig. 3.
Figure 3. Layout of the second filter, after optimization.
Once again, it can be noticed that simple and regular geometries of the resonators from Fig. 1.b were slightly
modified during the fine tunings of resonance frequencies and/or of couplings. The optimized responses of the designed filters, obtained by electromagnetic field simulation of their layouts in lossless conditions, in comparison to the responses of their lumpedelement models, are shown in Fig. 4 and in Fig. 5.
explained as being a consequence of the finite layout resolution accepted in the electromagnetic-field simulation (0.1 mm, in these designs). However, there are also some extra difficulties represented by the important difference between the numerical values of the coupling coefficients, especially between k13 and k14. Such severe discrepancies make very difficult a precise adjustment of the weaker coupling. Fortunately, the obtained overall attenuation of the adjacent channel remains very high even if the transmission zeros are not very precisely positioned, so that the main goal of the design is practically achieved. The simulated characteristics of the filters from Fig. 2 and from Fig. 3, in the presence of losses in the dielectric substrate and in the metallic sheets, are shown in Fig. 7 and in Fig. 8. The insertion loss of these filters, anticipated by simulation, is around 2.5 dB. It can be noticed the properties of the filters are in good agreement with the corresponding initial design specification. EXPERIMENTAL RESULTS
III.
The filters from Fig. 2 and Fig. 3 were fabricated in a classical microstrip technology and encapsulated in an appropriate metallic housing (Fig. 6). Figure 4. The response of the (lossless) filter from Fig. 2 and of its lumped-elements model.
a)
b)
Figure 6. Experimental models of the designed filters.
Both filters were tested using an Agilent E5071C vector network analyzer. Their measured characteristics are shown in Fig. 7 and in Fig. 8, compared to their EM-field simulated responses in the presence of losses. 0 -10 -20
The simulated responses of the designed filters show strongly asymmetric characteristics, allowing an excellent attenuation of the upper, respectively lower adjacent channels. The differences between the characteristics of the microstrip filters and their lumped-element models can be explained, at least partially, by the fact that a good equivalence of the two circuits is possible only in a narrow bandwidth, largely exceeded in this design example. A specific issue, at least for the considered filter specification, is the difficulty of obtaining the accurate positions of the prescribed rejection frequencies. In the actual designs, the filter's zero transmission frequencies are not exactly located at the prescribed values. This can be partially
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Figure 5. The response of the (lossless) filter from Fig. 3 and of its lumped-elements model.
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Figure 7. Measured and simulated (including losses) characteristics of the filter from Fig. 6.a.
The experimental models exhibit a center frequency of around 3.52 GHz, an in-band insertion loss of about 3 dB (filter from Fig. 6.a) and of about 2.5dB (filter from Fig. 6.b). The
return loss of both filters is better than 10 dB, as expected. The measured 3 dB-bandwidth is of about 195 MHz for the first filter and about 209 MHz for the second one, slightly larger than the equi-ripple band specification. In both cases the attenuation of the corresponding adjacent channel can be estimated as better than 48 dB, in a frequency band larger than this channel. 0
The new structures are well-suited for miniaturization. Microwave filters of the type investigated in this paper seem to be very appropriate for the design of planar low-cost miniature diplexers.
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In comparison to other (higher-order) filters with similar adjacent channel attenuations, the filters with this topology exhibits responses with lower insertion loss.
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A minor drawback of the new configuration stays in the difficulty of obtaining exactly the prescribed positions for the rejection frequencies. However, this lack of precision has only a very limited effect on the characteristics in real (lossy) filters, usually not affecting significantly the high attenuation of the adjacent channel.
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ACKNOWLEDGMENT
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The work has been co-funded by the Sector Operational Program Human Resources Development 2007-2013 of the Romanian Ministry of Labor, Family and Social Protection through the Financial Agreement POSDRU/89/1.5/S/62557.
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Figure 8. Measured and simulated (including losses) characteristics of the filter from Fig. 6.b.
IV.
CONCLUSIONS
The paper proposes a novel configuration of fourth-order microwave planar bandpass filters with strongly asymmetric responses. The new topology offers two transmission zeros on the same side of the filter's passband, leading to improved performances in term of one adjacent channel attenuation. The paper shows also the possibility of an accurate design of such structures. The performances of the experimental models are relatively close to the design values, despite of the available low-cost technology, not very accurate.
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D. Swanson and G. Macchiarella, “Microwave filter design by synthesis and optimization,” IEEE Microw. Magazine, vol. 8, pp. 55–69, April 2007. D. G. Swanson, “Narrow-band microwave filter design”, IEEE Microw. Magazine, vol. 8, pp. 105–114, October 2007. J. S. Hong and M. J. Lancaster, “Couplings of Microstrip Square OpenLoop Resonators for Cross-Coupled Planar Microwave Filters”, IEEE Trans. Microw. Theory Tech., vol. 44, pp. 2099–2109, November 1996. R. J. Cameron, “General Coupling Matrix Synthesis Methods for Chebyshev Filtering Functions”, IEEE Trans. Microw. Theory Tech., vol. 47, pp. 433–442, April 1999. R. J. Cameron, “Advanced Coupling Matrix Synthesis Techniques for Microwave Filters”, IEEE Trans. Microw. Theory Tech., vol. 41, pp. 1– 10, January 2003. ***, “em User’s Manual”, Sonnet Software, Inc., New York – Sonnet Professional ver. 12.56, Available at : http://www.sonnetsoftware.com.
