IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 4, APRIL 2008
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On the Efficiency of Voltage Source and Current Source Inverters for High-Power Drives Eduardo P. Wiechmann, Senior Member, IEEE, Pablo Aqueveque, Student Member, IEEE, Rolando Burgos, Member, IEEE, and José Rodríguez, Senior Member, IEEE
Abstract—The energy performance of various types of voltagesource and current-source converters is examined. For fairness and completeness, efficiency is calculated for three major battleground scenarios. The first is a low dynamic nonregenerative group of applications such as pumps, fans, and compressors. This group represents 85% of high power (∼2 MW) industrial applications where energy savings are usually a primary consideration justifying investment. The second scenario considers applications requiring good dynamic response and regenerative braking. Finally, the third group considers very high power applications (over 20 MW). The evaluation presented takes into account semiconductor switching and conduction losses, losses in the medium voltage feeding transformer (determined per IEEE Standard C57.18.10-1998), and the losses in ac and dc filters. For purposes of analysis, computer simulations validated against measurements taken on a 1-MW voltage source inverter (VSI) and a 1.4-MW current source inverter (CSI) were used. The results of the first scenario show competitive efficiencies for VSI and CSI drives, whereas voltage source-based solutions are more energy efficient in the second scenario considered. For the last group, the current source load-commutated inverter exhibits the best performance. Index Terms—AC motor drives, efficiency, electric machines, motor drives, multilevel systems, power electronics, pulsewidth modulation (PWM).
I. I NTRODUCTION
R
ECENT advancement in power semiconductors technology has seen insulated gate bipolar transistors (IGBT) and integrated gate commutated thyristors (IGCT) rapidly replace gate turn-off thyristors (GTO) in high-power applications [1]–[3]. IGBTs and GCTs presently have a voltage blocking capability of up to 6 kV, with saturation voltages ranging between 3 and 4.5 V and 2 and 3 V, respectively. Thus, they can switch at frequencies up to 1 kHz, enabling their usage in high power medium-voltage adjustable speed drives (ASD) in numerous industrial processes [4]. Manuscript received March 21, 2007; revised January 3, 2008. This work was supported by Grant 1060902 from the Chilean Fund for Science and Technology Development (CONICYT) and by ENAP Chile for facilitating measurement of efficiency at its oil refinery installations in Concepción, Chile. This work used the Engineering Research Center Shared Facilities supported by the National Science Foundation under NSF Award EEC-9731677 and the CPES Industry Partnership Program. E. P. Wiechmann and P. Aqueveque are with the Department of Electrical Engineering, Concepción University, Concepción 160-C, Chile (e-mail:
[email protected]). R. Burgos is with the Center for Power Electronics Systems, Virginia Polytechnic Institute and State University, Blacksburg, VA 24060 USA. J. Rodríguez is with the Electronics Engineering Department, Universidad Técnica Federico Santa María, Valparaíso 110-V, Chile. Digital Object Identifier 10.1109/TIE.2008.918625
Nearly 85% of motors in industrial processes are used to drive pumps, fans, and compressors. These are relatively simple applications that do not require any advanced controls due to their relatively benign dynamics—slow torque and speed transitions, which can nonetheless achieve significant energy and cost savings, has been well proven in the past [1], [4]. Processes like mining, metallurgy, and traction have therefore readily incorporated ASD technology, benefiting from its advantages, pursuing further increases in production, product quality, and revenues [4] as well. The remaining applications require more advanced ASDs, which usually employ regenerative braking and demand a high dynamic performance [8], [9]. Finally, an important family of applications not covered by this paper demands low motor speeds usually less than 1200 rpm. The preferred solution is to use custom-made cycloconverter and slip-power recovery drives [1], [2]. ASD manufacturers have developed multiple types of drives to answer the needs of multiple industrial applications; however, the choice of standard equipments (not custom made) for high-power processes is not as broad and is actually quite limited, as shown in Table I. These drives can be classified into two main categories: voltage and current source converters. Past research has studied and analyzed control strategies as well as pulsewidth modulation (PWM) schemes for voltage- and current-source inverters and rectifiers (VSIs and CSIs) where significant accomplishments have been achieved [10], [16], [21], [22], e.g., harmonic distortion minimization, high-input power factor, and reduced switching frequencies, among others. Power conversion efficiency has not been as thoroughly studied, however, or even compared with these two alternative VSI and CSI families of converters [10]–[12], [18], [20]. Naturally, comparison is of utmost importance at the high-power level and is often the tipping point for selecting one or the other. VSIs have the advantage of requiring only a simple diode bridge as a front-end converter, featuring minimum costs, high efficiency, and high reliability for the rectifying stage. The main disadvantage of VSI converters is the generation of high dv/dt transients of harmful effects on the motor insulation—depending on the feeder impedance, and thus distance between converter and motor [12], which, if not filtered, can reduce the motor lifetime significantly. Another drawback of this topology is the generation of common-mode voltages, also affecting motor insulation and motor bearings. On the other hand, high-power CSIs have the advantage of requiring a reduced number of semiconductors since symmetric IGCTs can block reverse voltage. Also, since output capacitors are required for CSI operation, voltages and currents applied
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TABLE I HIGH-POWER DRIVES OFFERED BY MAJOR MANUFACTURERS
to the motor are quasi-sinusoidal because this capacitive filter eliminates the high di/dt of the inverter PWM action [17]. This type of drive also requires a controlled current source in its dc link, for which an active front-end converter is used to regulate this current [19], [30]. Although this is a disadvantage for nonregenerative applications, since it increases drive complexity, the PWM CSI converter intrinsically offers sinusoidal currents and a high-power factor at its input terminals [21]. Also, the converter offers an additional short-circuit protection provided by the dc-link choke (current source). The latter, however, represents a drawback regarding size, volume, and weight because this dc-inductor is usually quite heavy and bulky. Another disadvantage of this converter type is the possible occurrence of resonances between the output capacitive filter and the motor inductances [19]. As we can see, there are many trade-offs involved in choosing between the different ASD technologies. The final decision is usually made based on cost and economical factors, or simply by following desired industrial practices. The objective of this paper is to provide a fair analysis and comparison tool for highpower VSI and CSI converters, based on the power conversion efficiency analysis of the main set of industrial high-power motor drives, including pumps, fans, and compressor applications, motor drives requiring more advanced features such as regeneration or high dynamic performance, and converters used in very high power applications (> 20 MW). The efficiency comparison performed takes the semiconductor conduction and switching losses [13] into account as well as the losses on the feeding transformer [23], which is required for isolation purposes and for harmonic distortion minimization in medium voltage converters. This analysis methodology is easily applied to alternative high-power medium-voltage converters topologies. II. H IGH -P OWER D RIVE T OPOLOGIES Different manufacturers currently participate in the mediumvoltage market offering a variety of drives [4], [14]. Table I summarizes the main series of products available, indicating the type of topology, semiconductor devices, power and voltage
range used as well as manufacturer and model name. The mostused inverter topologies are the two-level, three-level neutral point clamped (NPC) [5]–[7], and cascaded multilevel H-bridge VSIs. All of these use diode front-end rectifiers, either 12 or 18 pulses, depending on the specific harmonic requirements [15]. If the application at hand requires regeneration, then PWM rectifiers are used to replace diode front-end converters. Because this paper focuses on comparing voltage- and currentsource topologies, only two- and three-level topologies are analyzed in detail. Cascaded multilevel H-bridge inverters were not evaluated because there is currently no direct current-source equivalent, and the feasibility of such industrial development is very low. As for CSI converters, the most used topology is the threephase bridge using series semiconductors to support higher operating voltages. The rectifying stage for these converters employs PWM current-source rectifiers (CSRs), which have replaced thyristor-phase-controlled rectifiers offering sinusoidal input currents with a near-unity power factor. So far, Rockwell Automation is the only manufacturer producing current-source converters. Very high power applications, on the other hand, use load-commutated inverters, which are implemented as CSIs driving synchronous machines up to 72 MW. These converters are well known for their high efficiency due to the reduced conduction voltage drop of their thyristors, which usually present saturation voltages on the order of 1.4 V. III. R ELIABILITY AND F AULT T OLERANCE The reliability and fault tolerance of medium-voltage converters is of interest, given the critical nature of their functions within industrial sites. In VSI topologies, the most common and destructive fault by far is the shoot-through [31], for which different protection schemes have been developed based on the desaturation protection that effectively opens the switches in the phase leg before their destruction occurs [31], [32]. CSI topologies, on the other hand, are prone to destructive failures under open circuit faults—due to faulty semiconductor gate driving—for which no solution exists. However, the use of light-triggered thyristors has virtually eliminated this fault [33].
WIECHMANN et al.: ON THE EFFICIENCY OF VOLTAGE SOURCE AND CURRENT SOURCE INVERTERS FOR HIGH-POWER DRIVES
For very high power drives, the use of redundant semiconductors is a standard practice to increase the converter reliability. These can be implemented as paralleled devices or paralleled power modules, for instance redundant phase legs. Welchko et al. [31] show some usual practices to improve converter availability.
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TABLE II SEMICONDUCTORS PARAMETERS
IV. L OSSES IN H IGH -P OWER D RIVES A. Conduction Losses Conduction losses of high-power semiconductors are a function of the device saturation voltage (VSat ) and the instantaneous current passing through it, i(t). The device saturation voltage can be modeled using a first-order linear approximation comprised of a threshold voltage (VT O ) and a series resistance (rT ) as follows: VSat (t) = VT 0 + rT · i(t).
(1)
The device average conduction losses (Pcond ) over a fundamental frequency period may then be obtained multiplying (1) by i(t) and integrating accordingly. These are given as follows: 2 Pcond = VT 0 · IAvg + rT · IRM S
(2)
where IAvg corresponds to the average current flowing through the device, and Irms to the rms value. In this paper, the conduction power losses were calculated numerically using computer simulations; specifically, using (2) to continuously calculate the total conduction losses of each converter considered. B. Switching Losses Power semiconductor switching losses are determined by the total commutation time when the device is turned on or off, and by the voltage (V ) and current (I) across the device during the process. The energy dissipated during commutation is Eon and Eoff for turn-on and turn-off, respectively, and is provided by the device manufacturers on their datasheet; it corresponds to the integral of the instantaneous power during the commutation process. This energy value is given by manufacturers at a specified voltage and current. The following: linear interpolation is used at every switching instant to determine the energy dissipated at the operating points throughout a line cycle Eon/off (t) = Eon/off−datasheet ·
V (t) · I(t) . (3) Vdatasheet · Idatasheet
In (3), the average switching power losses (Psw ) over a complete fundamental-frequency cycle may be determined using the following summation, where fsw corresponds to the switching frequency used, T to the fundamental frequency period, and k to each of the discrete switching instants (T fsw/cycle ) Psw =
T ·fsw 1 (Eon (k) + Eoff (k)) . T k=1
(4)
The switching losses in this paper are calculated numerically using computer simulations just as for conduction losses, which in this case evaluate (4) to calculate the total switching losses of each converter considered (per switch). The semiconductor parameters used for the different topologies and devices are given in Table II. For high-power VSI and CSI, the use of semiconductors in series requires snubbers to balance voltage. Their losses have been computed and added as switching losses [1]. C. DC-Link Losses The losses on the dc-link choke for CSI converters are produced by its equivalent series resistance; these are usually 2%–4% of the rated power of the inductor. A disadvantage of this type of converter is that when the inverter operates with nominal dc current, losses on the inductor will be the maximum regardless of the load demand. However, control schemes regulating the dc-link current, such as those for drives with low dynamic requirements, will present fewer losses, being more favorable from an efficiency point of view. On the other hand, VSIs present lower losses when compared to CSI converters, since capacitors are very efficient devices in general. In fact, the total loss of a dc-link capacitor bank rarely exceeds 0.5% of its rated value. D. Transformer Losses High-power transformers feature very high efficiency (premium manufacturing category) due to the considerable economic impact that their operation has on industrial processes [25]. Thus, an efficiency of 98.5% is common for mediumvoltage transformers. Still, the different transformer configurations used by medium-voltage drives could easily tilt the efficiency performance from one to another, a reason why the losses in the feeding transformers have also been taken into account in this study. Specifically, IEEE Standard 57.18.10–1998 presents a method for calculating all factors affecting the determination of transformer losses under the presence of harmonic currents [24], which are the main contributors to transformer power losses in motor drives. This method has been chosen in this paper, since the currents circulating through the transformer windings may be easily determined from the detailed switching models used.
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Fig. 1. ASD with diodes rectifier and PWM VSI.
For example, this section discusses the loss calculation for the delta-wye isolation transformer of the diode front-end PWM VSI motor drive, although the method has been applied to all the topologies under consideration. Transformer losses are proportional to the square of the voltage (iron losses) and proportional to the square of the load current (copper losses); therefore, the power factor seen at the input of the drive will have a direct impact on the total losses. Additionally, the harmonic content of the converter currents— measured by the k factor—will further increase losses on the transformer. It is clear, then, that the better the power factor is and the more sinusoidal the currents flowing through the transformer are, the lower the total losses will be in the transformer. Further loss reduction requires optimizing transformer mechanical design and construction. For instance, increasing the conductor section reduces losses by reducing the conduction losses and allowing the transformer to operate at a lower temperature, since it is known that operating at lower temperatures improves the transformer efficiency. This type of solution, however, should be weighed carefully to determine the optimum balance between increased efficiency and the additional initial investment required in the modified transformer. As an example, losses of the six-pulse motor-drive transformer are shown in Appendix A. This procedure is in accordance with IEEE Standard 57.18.10–1998. E. Motor Losses The use of PWM inverters for the speed control of induction machines generates additional losses due to the harmonics currents that circulate through their windings. These losses depend on the types of modulation schemes and machines that are used [27]. Specifically, voltage source inverters increase iron losses, which has been thoroughly studied, as seen in [28], [29]. It is found that the use of sinusoidal PWM and space vector modulation increase losses in approximately 1.5% of the cases. CSIs, on the other hand, offer a much better motor loss performance due to the mandatory capacitive filter at its output terminals. Losses induced on induction machines by these converters are then perfectly comparable to motors fed from sinusoidal three-phase voltages [30] and are significantly lower than those generated by PWM VSI converters. A comparison of total losses is presented in Section IX.
V. H IGH -P OWER D RIVE E FFICIENCY FOR C ONVENTIONAL A PPLICATIONS As mentioned previously, the vast majority of industrial applications employing medium-voltage drives are fans (30%), pumps (40%), and compressors, extruders, and conveyors (15%). The remaining 15% corresponds to applications requiring more advanced and specific drive types. These are analyzed in Section VI. Conventional applications normally operate with low speed dynamics and do not regenerate energy into the grid. The most common converters currently available in the market are 12-pulse rectifiers with PWM-VSI, 12-pulse rectifiers with three-level NPC VSI, and PWM current-source converters (CSR + CSI). The following sections show the analysis and loss calculation for each of these three converters driving a 1-MW motor. The losses considered are power transformers, semiconductors, and filters.
A. Diode Rectifier and PWM VSI The most-used converter for low-voltage (< 1 kV) industrial applications consists of a six-pulse diode rectifier, an LC dc-bus filter, and a PWM VSI. This topology has been extended to include medium-voltage applications, thanks to the development of high-voltage IGBTs, which, when used by series-connected devices, are capable of feeding motors of up to 4160 V as shown in Fig. 1. The purpose of the dc inductor is to smooth out the dc-link current and improve the input current distortion, reducing the fifth and seventh current harmonics at the input. Diode commutation losses vary with the actual current and the grid impedance. Because the reverse recovery of fast power diodes is very small, switch losses are neglected in this study. The parameters VT 0 and rT for the diodes used are 0.94 V and 0.147 mΩ as per Table II. The VSI, on the other hand, has eight possible active states. Six of them transfer energy to the load, and two are zero voltage or freewheeling states where the motor phases are shorted. Regardless of the state, only the top or bottom IGBTs (highlighted in Fig. 1) are turned on per phase leg at any given time to avoid shorting the dc bus—a fatal failure. Semiconductor thermal characteristics restrain the highpower VSIs (> 1 MW) to use switching frequencies of up to 1 kHz, which also limits the switching losses of the motor drive. To match experimental data in this paper, a switching frequency
WIECHMANN et al.: ON THE EFFICIENCY OF VOLTAGE SOURCE AND CURRENT SOURCE INVERTERS FOR HIGH-POWER DRIVES
Fig. 2.
ASD with 12 pulses diodes rectifier and NPC multilevel VSI.
Fig. 3.
CSI converter with PWM rectifier and inverter.
of 1 KHz is used at a nominal load to calculate the power losses of the VSI for a 1-MW motor drive fed by a six-pulse diode rectifier. The losses of the delta-wye isolation transformer are also calculated. B. 12-Pulse Rectifier and Three-Level NPC VSI The higher power demand of industrial applications using ac motors has driven the increase of dc-bus voltages to limit current amplitude and achieve high-power conversion efficiencies. To compensate for the lack of high-voltage semiconductor devices, multilevel converters have been developed; NPC topology is by far the most utilized for medium-voltage VSIs as shown in Fig. 2. The main effect of multilevel voltage waveforms is to reduce the dv/dt voltage stress on the motor insulation, thus extending its lifetime. Also, the harmonic spectrum is improved for the same switching frequency when compared to a two-level VSI, reducing the losses and torque ripple in the motor. Use of a 12-pulse diode rectifier as a frontend converter also improves the harmonic distortion at the input lines, increasing the power factor and reducing the losses on the 12-pulse isolation transformer. Its main drawback is that the dc-link current is now conducted by four diodes, although this has a small overall effect because the voltage level for this type of converter is usually higher. The dc-link current magnitude in consequence is lower for the same power level when compared to a 6-pulse rectifier. Overall, the reduced input and output harmonic distortion will contribute to a higher efficiency of this medium-voltage drive topology.
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The NPC-VSI has two IGBTs in the turned-on state at any given time per converter phase-leg. Consequently, the output phase-voltage of the converter can take any of three possible levels: −Vdc /2, 0 or +Vdc /2. Energy is drawn or injected into the dc-link clamped capacitors depending on the converter state applied. As a result, the capacitor voltages need to be controlled to avoid any unbalance between them. Multiple control schemes have been developed effectively for this purpose, thus solving this problem [6], [7]. As we can see, the two center IGBTs in the phase leg (Sw1b and Sw4a for phase leg a, Fig. 2) conduct continuously for nearly half the fundamental frequency period, whereas the top and bottom phase leg IGBTs conduct in a pulse fashion during the same period. This implies that the predominant losses for the center and outer IGBTs are conduction and switching losses, respectively. C. PWM CSR-CSI Converters Fig. 3 shows the circuit schematic of a current-source converter comprised of a PWM CSR and CSI using symmetric GCT semiconductors. These GCTs have high saturation voltages peaking at 8 V for currents of 1 kA (ABB 5SHZ 08F6000, Mitsubishi GCU04AA-130), which correspond to twice the saturation voltage of asymmetric IGCTs used for voltage-source converters. The advantage of the symmetric device is its higher voltage blocking capacity (up to 6 kV), which, if combined with the use of series-connected arrangements of GCTs, allows this topology to drive 6.6 kV motors at high-power levels. As seen in this figure, both rectifier and inverter use the same bridge
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Fig. 4. PWM AFE and VSI.
topology. Both converters are also controlled using space vector modulation, a natural fit for current-source topologies [19]. An advantage of the current-source converter over voltagesource types is that the dc-link current may be controlled at will if the load dynamic allows it. Thus in fan, pump, and compressor applications with low dynamic requirements, this current may be reduced according to the operating point to maintain a high modulation index of the CSI and reduce the switching and conduction losses of the semiconductor devices. CSRs or CSIs have nine electrical states, six active ones where power is transferred to the load and three zero states where the dc-link current is freewheeled through the inverter back into the dc-link. The switching losses for these two converters are naturally different since the CSR blocks the line voltages while the CSI blocks the operating point motor voltages.
thus its modulation and operation are the same as that of a VSI. The losses of this topology are slightly higher than its passive front-end counterpart due to higher losses of IGBTs when compared to power diodes. The AFE and VSI also present similar losses, although these losses are distributed differently between IGBTs and diodes, because the diodes conduct more than the diodes of the VSI for the front-end converter due to the near-unity displacement power factor at the input terminals. This converter topology, with series-connected IGBTs, can operate up to voltage levels of 4 kV; however, its usage in highpower applications is still limited due to the reduced reliability of the series connection of devices. For the same reason, manufacturers normally recommend the NPC back-to-back voltagesource converter for high-performance applications. B. Back-to-Back NPC Voltage-Source Converter
VI. E FFICIENCY OF H IGH -P ERFORMANCE C ONVERTERS High-performance drives require front-end converters with regenerative capabilities as well as inverters with fast dynamic responses to accomplish precise motion controls and compensate for sudden load changes. The previously presented medium-voltage drives do not meet these requirements, and thus modified topologies have been developed. Specifically, VSI-based topologies replace the diode front end with an active front-end voltage-source converter with which they can readily regenerate energy into the grid. Current-source converters, on the contrary, are intrinsically regenerative (by dc-link voltage inversion) and do not require any hardware modification. However, to achieve a high dynamic response, the dc-link current must be kept at its nominal value to provide an instantaneous full-torque capability. The following: sections present highperformance medium-voltage converters currently available. A. PWM VSI Converters With Active Front Ends (AFE) The main features of the AFE-VSI converter are its regenerative capability and high-input power factor attained by a zero displacement angle and low harmonic distortion of its input currents. Fig. 4 shows the simplest topology embodying this drive. The AFE is simply a VSI connected to the grid,
The three-level NPC-VSI has been successfully used in the industry in past years. The same multilevel topology has been used to realize the AFE converter for this inverter, providing the required regeneration capability for the drive. The use of a three-level NPC AFE (Fig. 5) has the additional advantage of reducing the input current harmonic distortion. The operation and control of this converter is the same as that of its inverter counterpart—just as for the two-level voltage-source converter. Therefore, the loss distribution is similar among each phase-leg IGBTs, with the outer IGBTs having higher switching losses and the center IGBTs higher conduction losses. Also, the loss distribution between IGBTs and diodes is different for the AFE and VSI due to the near unity input displacement power factor of the AFE. C. CSI Converter With PWM Rectifier For high-performance applications requiring a fast dynamic response, the CSR-CSI is another topology available in the market. This is exactly the same topology used for conventional medium-voltage drives; therefore, the dc-link current is always conducted by two switches per converter (Fig. 3). The difference lies in the fact that to achieve a high dynamic response, the dc-link current is always kept at its nominal value. By doing so,
WIECHMANN et al.: ON THE EFFICIENCY OF VOLTAGE SOURCE AND CURRENT SOURCE INVERTERS FOR HIGH-POWER DRIVES
Fig. 5.
Multilevel NPC AFE and VSI.
Fig. 6.
Five-level VSI converter for high-power applications.
the load current may be changed by the PWM modulator from zero to its nominal value almost instantaneously if required. The downside of this scheme is a significant increase in both conduction and switching losses of the rectifier and inverter, since the dc-link current is now at maximum regardless of the operating point. This affects the efficiency of this converter, which when operating at 10% nominal power, presents efficiencies lower than 90%, for example. The results presented in Section VII verify this. VII. D RIVES FOR V ERY H IGH -P OWER A PPLICATIONS For very high power applications, the previously described topologies cannot be used, which is why manufacturers have developed numerous topologies appropriate for motor drives operating at high-power ratings, such as in excess of 20 MW. In the case of voltage-source inverters, five-level inverters fed from 24- or 36-pulse diode rectifiers are a standard solution. Current source topologies, on the other hand, use thyristorbased rectifiers and inverters, such as phase-controlled rectifiers and load commutated inverters (LCI). A. 24-Pulse Rectifier and Five-Level VSI The five-level diode clamped VSI uses eight series-connected semiconductors per converter phase leg, enabling operation with voltages of up to 8 kV, supported by four series capacitors in the dc-bus. This high voltage translates into relatively low current levels for the motor drive so that the semiconductor
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losses remain low. Fig. 6 shows a circuit diagram of this converter consisting of 24 high-voltage IGBTs. Different vendors indistinctively offer IGBT- or IGCT-based configurations. This converter can operate motors up to 20 MW, usually encountered in grinding mills or low-speed applications. As the figure shows, the front-end rectifier is a 24-pulse rectifier with four series-connected three-phase bridges [15]. The input current THD of this rectifier is less than 3%; however, depending on the harmonic requirements, this configuration is increased to 36 pulses. The input power factor is also very high, being close to unity throughout the complete operating range. Although this converter is nonregenerative, few high-power applications have these requirements, so the topology is still significantly used. B. Phase-Controlled Rectifier and Load-Commutated Inverter This motor drive is only used for very high-power applications driving synchronous machines. Fig. 7 shows a circuit diagram of the converter depicting its full-thyristor realization. The use of thyristors enables this converter to reach voltages as high as 10 kV, featuring saturation voltages in the order of 1.3 and 1.5 V, respectively, for currents up to 4 kA. Consequently, this converter is highly efficient and presents losses significantly lower than any other converter using IGBTs, IGCTs, or symmetric GCTs. The major drawback of this converter is its relatively high harmonic distortion due to the intrinsic six-pulse operation as well as its poor input power factor due to the phase control of the rectifying stage.
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Fig. 7. Load-commutated current-source motor drive.
The inverter works similarly to the rectifier under natural load voltage commutation—instead of line voltage commutation, only synchronous motors can be driven because these machines can have a displacement angle between their terminal voltage and current adjusted to ensure the safe commutation of thyristors. Both the inverter and rectifier have six active states, just as a diode bridge does; therefore, two thyristors conduct in each converter at a time. The conduction losses can then be determined by calculating the losses on four thyristors conducting the dc-link current. For higher power applications, multiple three-phase bridges can be connected in series-driving motors up to 72 MW. VIII. E XPERIMENTAL V ERIFICATION To validate the results obtained through simulation, several measurements to calculate the efficiency of medium-voltage VSI and CSI were conducted. For VSI a 1 MVA, 2.4 kV pump operating in an oil refinery was considered. Efficiency measurements include a transformer, a six-pulse diode front end, a dc-link filter, and an IGBT-based two-level VSI. The efficiency measured in this converter operating at 56% nominal power was 96.0%, and was increased to 96.2% at nominal power. The second converter was a CSR-CSI driving a blower in the grinding plant of a mining site. The converter was fed from 3300 V 50 Hz and drove a load of 1250 HP. This measurement included transformer losses, rectifier and inverter semiconductor losses (SGCT), a dc-link inductor, and output filter. The efficiency measured at nominal power was 94.9%. The measurements were taken simultaneously at the input of the transformer and the output terminals of the converter, capturing the input and output three-phase voltages and currents. For purposes of accuracy, two sets of measurements were conducted on each converter, one using oscilloscopes (Fluke 199C and Tektronix TPS2014), and one using a power quality analyzer (Dranetz-BMI Power Guide 4400). These results are in good agreement with the efficiency predicted through simulation in the proposed method as shown in Tables IV and V and Figs. 8 and 9.
Fig. 8. Efficiency of VSI and CSI converters for full range of operation (Case 1).
Fig. 9. Efficiency of VSI and CSI converters for full range of operation (Case 2). TABLE III SIMULATION PARAMETERS
IX. G ENERAL E VALUATION The topologies presented in this paper for medium-voltage high-power converters have been simulated to calculate semiconductor, transformer, and filter losses and establish the advantages and disadvantages that each of them presents. To this end, all motor drives were run from 10% to a full load,
calculating the losses throughout the operating range using the parameters in Table III. The evaluation has been split three ways according to the paper’s structure—conventional, highperformance, and very high power applications. The results are presented as follows.
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TABLE IV LOSSES AND EFFICIENCY FOR VS- AND CSI-BASED DRIVES IN FULL RANGE OF OPERATION (CASE 1)
TABLE V LOSSES AND EFFICIENCY FOR VSI- AND CSI-BASED DRIVES IN FULL RANGE OF OPERATION (CASE 2)
A. Case 1 Table IV shows the results obtained for conventional medium-voltage converters. All converters drove the same 1 MW motor rated at 2.4 kV, shown in Table IV. As we can see, voltage-source topologies present a slightly higher efficiency due to the higher saturation voltage of the symmetric GCTs and the dc link inductor used by the PWM CSR-CSI converter. However, this 2% higher efficiency exhibited by VSIs becomes negligible when motor losses are included. As mentioned previously, CSI converters are 1.5% more efficient in the motor section of the drive. Table IV also shows that the multilevel NPC-VSI has the highest efficiency, achieved by the better harmonic spectrum of its currents, which reduces harmonic losses in the system. Due to lower conduction losses, the NPC with IGCT exhibits a 0.2% higher efficiency than the NPC with IGBT. However, for higher commutation frequencies, the efficiencies are similar due to the extra commutation losses of the IGCT. B. Case 2 Table V shows the results obtained for the high-performance converters under discussion. Simulation parameters are the same as for Case 1. Fig. 9 shows the efficiency plots for these motor converters. The current-source topology is comparable in efficiency with the voltage-source topologies at nominal load; however, the voltage-source topology presents a better performance even when considering 1.5% motor losses at a lower load level. This performance is due to the control scheme used for the current-source topology that keeps the dc-link current at a nominal value to maintain its instantaneous nominal torque
Fig. 10. Efficiency of VSI and LCI converters. 20 MW (Case 3).
capability. Consequently, the efficiency of this converter is reduced to 60% at light load conditions—a significant drop compared to the 93% efficiency of the voltage-source counterparts. Also, the multilevel back-to-back topology presents a slightly higher efficiency than the two-level back-to-back arrangement. Specifically, the NPC IGCT shows a slightly better efficiency thanks to the lower conduction losses of this type of device. C. Case 3 This evaluation used a 20-MW motor as the load, rated at 6.6 kV and 2.2 kA. As shown in Fig. 10, the losses of the voltage-source topology are nearly 50% higher than those of the current-source thyristor-based topology, which represents a difference of 250 kW at full load. Both converters use output filters, and therefore motor losses are quite similar. Also, the
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TABLE VI LOSSES AND EFFICIENCY FOR VSI- AND LCI-BASED DRIVES IN FULL RANGE OF OPERATION FOR 20 MW NOMINAL (CASE 3)
TABLE VII TRANSFORMER PARAMETERS
TABLE VIII HARMONIC LOSS FACTORS FOR DELTA-WYE TRANSFORMER
current-source topology in this case operates with minimum dc-link current, given the low dynamics of the load. Results are summarized in Table VI. For higher power converters (above 20 MW), LCI thyristor drives are the only choices for medium- to high-frequency applications because of their high efficiency and reliability. X. C ONCLUSION This paper presented the efficiency evaluation of voltagesource and current-source ac-to-ac medium-voltage high-power converter topologies, taking into consideration conventional nonregenerative low-dynamic applications, regenerative highperformance applications, and very high power applications. The analysis took into account semiconductor conduction and switching losses as well as the power losses of medium-voltage isolation transformers and filters. Results show that VSIs and CSIs are competitive for nonregenerative low dynamic requirement drives. For regenerative applications, the three-level NPC VSI achieves a higher efficiency in comparison to the CSI converter. For very high power applications, the thyristor-based current-source topology presents a significantly better performance due to the low-voltage drop of the semiconductors used. The proposed methodology may also be readily used for the analysis and evaluation of alternative high-power mediumvoltage topologies.
of rectifier being fed. The harmonic spectrum used for this example is given in Table VIII. These data are used to calculate the harmonic loss factors defined in IEEE Standard C57.18.10-1998 and are shown in Table VIII, where FHL−WE is defined by (5) and corresponds to the harmonic factor that increases the eddy current losses PEC , and FHL−OSL is defined by (6) and corresponds to the harmonic factor multiplying the stray inductance losses POSL FHL−WE =
n
Ih (pu)2 h2
(5)
Ih (pu)2 h0.8 .
(6)
1
FHL−OSL =
n 1
The kilovolt-ampere ratings of the transformer are as follows: A PPENDIX E XAMPLE FOR L OSS C ALCULATION IN P OWER T RANSFORMERS The resistance values measured for the transformer’s different windings are as follows: Primary winding 366.8 mΩ Secondary winding 9.39 mΩ Measured impedance 6.18% Tested core loss 5328 W Tested load loss 28 452 W The transformer used in this example has the same harmonic current spectrum in its primary and secondary windings (parameters given in Table VII). This is not valid naturally for multiwinding transformers because the current distribution for this case can vary significantly depending on the number of rectifier pulses, the transformer configuration, and the type
Primary rms kVA = 1.0304 × 2700 = 2782 kVA Then the measured dc resistance losses at nominal load and at the fundamental frequency may be determined as follows, with iprim = 113 A and isec = 650 A: Primary winding I 2 R = 3 × (113)2 × 0.3668 = 14 041 W Secondary winding = 3 × (650)2 × 0.00939 = 11 884 W Total dc resistance load loss = 25 925 W. So far, the losses calculated for this transformer at the fundamental frequency are 25 925 W. Taking the new data into consideration, total stray losses POSL and winding eddy current PEC loss are then equal to 28 452 − 25 925 = 2527 W.
(7)
WIECHMANN et al.: ON THE EFFICIENCY OF VOLTAGE SOURCE AND CURRENT SOURCE INVERTERS FOR HIGH-POWER DRIVES
In this case, PEC data per winding were obtained from the transformer manufacturer as follows: Primary winding = 447 W Secondary winding = 1644 W Total windings PEC = 2091 W. With these values, the total stray losses can be calculated as follows: POSL = 2527 − 2091 = 436 W. The eddy current losses are measured through experimental tests and are normally provided by the manufacturer. If this information is not available, a series of measurements supplying the transformer with varying frequency voltages may be performed as indicated in [26], which presents a detailed description of this procedure. Nonetheless, IEEE Standard 57.18.10–1998 indicates that in case PEC and POSL are not available, a good approximation is to split the losses by 60% and 40%, respectively. The total service I 2 R loss is the fundamental I 2 R loss × (Irms /I1 )2 . The service winding eddy current loss is the fundamental eddy loss × FHL−WE . The additional stray loss is the fundamental additional stray loss × FHL−OSL Primary service I 2 R loss = 14 041 × 1.03042 = 14 908 W Primary service winding eddy current loss : PEC−P
= 447 × 4.2190 = 1886 W Total primary winding service loss : PR−P = 16 794 W Secondary service I 2 R loss = 11 884 × 1.03042 = 12 618 W Secondary service winding eddy current loss : PEC−S
= 1644 × 4.2190 = 6936 W Total secondary winding service loss : PR−S
= 19 554 W Service additional stray loss : POSL
= 436 × 1.2783 = 557 W. Total service load loss for the main transformer with distorted load current = 16 794 + 19 554 + 557 = 36 904 W. PR−m
To obtain the total service loss, the tested core losses are added to the total service load loss calculated above, which yields = 5328 + 36 904 = 42 232 W. PTOTAL
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R EFERENCES [1] B. Wu, High Power Converters and AC Drives. Piscataway, NJ: IEEE Press, 2006. [2] B. Bose, Power Electronics and Motor Drives—Advances and Trends. New York: Academic, 2006. [3] P. K. Steimer and H. E. Grunning, “IGCT—A new emerging technology for high power, low cost inverters,” IEEE Ind. Appl. Mag., vol. 5, no. 4, pp. 12–18, Jul./Aug. 1999. [4] J. A. Oliver, M. J. Samotyj, Y. Shakarian, and Y. Vinitsky, “Adjustablespeed drives for thermal power plant boiler feed pumps. Russian and American experience,” in Proc. IEEE Elect. Mach. Drives Conf., May 18–21, 1997, pp. TC2/7.1–TC2/7.3. [5] R. Sommer and A. Mertens, “New medium voltage drive systems using three-level neutral point clamped inverter with high voltage IGBT,” in Proc. IEEE Ind. Appl. Soc. Annu. Meeting, 1999, pp. 1513–1519. [6] J. E. Espinoza, J. R. Espinoza, and L. A. Moran, “A systematic controllerdesign approach for neutral-point-clamped three-level inverters,” IEEE Trans. Ind. Electron., vol. 52, no. 6, pp. 1589–1599, Dec. 2005. [7] A. Bendre, G. Venkataramenan, D. Rosene, and V. Srinivasan, “Modeling and design of a neutral-point voltage regulator for a three-level diodeclamped inverter using multiple-carrier modulation,” IEEE Trans. Ind. Electron., vol. 53, no. 3, pp. 718–726, Jun. 2006. [8] J. Rodríguez, J. Dixon, J. Espinoza, J. Pontt, and P. Lezana, “PWM regenerative rectifiers: State of the art,” IEEE Trans. Ind. Electron., vol. 52, no. 1, pp. 5–22, Feb. 2005. [9] J. Rodríguez, J. Pontt, N. Becker, and A. Weinstein, “Regenerative drivers in the megawatt range for high-performance downhill conveyors,” IEEE Trans. Ind. Appl., vol. 38, no. 1, pp. 203–210, Jan./Feb. 2002. [10] F. Blaabjerg, U. Jaeger, and S. Munk-Nielsen, “Power losses in PWM-VSI inverter using NPT or PT-IGBT devices,” IEEE Trans. Power Electron., vol. 10, no. 3, pp. 358–367, May 1995. [11] A. Fratta and F. Scapiano, “Modeling inverter losses for circuit simulation,” in Proc. 35th Annu. IEEE Power Electron. Spec. Conf., Aachen, Germany, Jun. 20–25, 2004, vol. 6, pp. 4479–4485. [12] E. Matheson, A. Von Jouanne, and A. Wallace, “Evaluation of inverter and cable losses in adjustable speed drive applications with long motor leads,” in Proc. IEDM, May 9–12, 1999, pp. 159–161. [13] T. Setz and M. Lüscher, Applying IGCTs. ABB Semiconductors, Feb. 2006. [14] “Rockwell Automation,” Power Flex 7000 Medium Voltage Drives, 2002. [Online]. Available: www.ab.com [15] D. Paice, Power Electronic Converter Harmonics—Multipulse Methods for Clean Power, 2nd ed. New York: IEEE Press, 1996. [16] M. H. Rashid, Ed., Handbook of Power Electronics, New York: Academic, 2001, ch. 12, pp. 599–627. [17] J. Rodríguez, L. Morán, J. Pontt, R. Osorio, and S. Kouro, “Modeling and analysis of common-mode voltages generated in medium voltage PWM-CSI drives,” IEEE Trans. Power Electron., vol. 18, no. 3, pp. 873– 879, May 2003. [18] J. Espinoza and G. Joos, “Current source converter on-line pattern generator switching frequency minimization,” IEEE Trans. Ind. Electron., vol. 44, no. 2, pp. 198–206, Apr. 1997. [19] E. P. Wiechmann, R. P. Burgos, and J. R. Rodriguez, “Reduced switching frequency active front end converter for medium voltage current source drive using space vector modulation,” in Proc. IEEE ISIE, 2000, pp. 288–293. [20] J. Espinoza and G. Joos, “State variable decoupling and power flow control in PWM current-source rectifiers,” IEEE Trans. Ind. Electron., vol. 45, no. 1, pp. 78–87, Feb. 1998. [21] S. Kwak and H. Toliyat, “Current-source-rectifier topologies for sinusoidal supply current: Theoretical studies and analyses,” IEEE Trans. Ind. Electron.—Letters to the Editor, vol. 53, no. 3, pp. 78–87, Jun. 2006. [22] J. Espinoza and G. Joós, “DSP based space vector PWM pattern generators for current source converters,” Can. J. Electr. Comput. Eng., vol. 22, no. 4, pp. 155–161, Nov. 1997. [23] L. Pierce, “Transformer design and application considerations for nonsinusoidal load currents,” IEEE Trans. Ind. Appl., vol. 32, no. 3, pp. 633– 645, May/Jun. 1996. [24] IEEE Standard Practices and Requirements for Semiconductor Power Rectifier Transformer, IEEE Std. 57.18.10-1998, 1998. [25] J. A. Philip and P. E. Ling. (2003, Aug.). Overcoming Transformer Losses, Powersmiths Corp. Brampton, ON, [Online]. Available: http:// ecmweb.com [26] E. F. Fuchs, D. Yildirim, and W. M. Grady, “Measurement of eddy-current loss coefficient PEC−R , derating of single-phase transformers, and comparison with K-factor approach,” IEEE Trans. Power Del., vol. 15, no. 1, pp. 148–154, Jan. 2000.
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[27] Y. Wu, R. A. Mcmahon, Y. Zhan, and A. M. Knight, “Impact of PWM schemes on induction motor losses,” in Conf. Rec. 41st IEEE IAS Annu. Meeting, Tampa, FL, Oct. 2006, vol. 2, pp. 813–818. [28] A. Boglietti, P. Ferraris, M. Lazzari, and M. Pastorelli, “Influence of the inverter characteristics on the iron losses in PWM inverter-fed induction motors,” IEEE Trans. Ind. Appl., vol. 32, no. 5, pp. 1190–1194, Sep./Oct. 1996. [29] J. Lee, Y. Kim, H. Nam, K.-H. Ha, J. Hong, and D. Hwang, “Loss distribution of three-phase induction motor fed by pulsewidth-modulated inverter,” IEEE Trans. Magn., vol. 40, no. 2, pp. 762–765, Mar. 2004. [30] J. R. Espinoza and G. Joos, “A current-source-inverter-fed induction motor drive system with reduced losses,” IEEE Trans. Ind. Appl., vol. 34, no. 4, pp. 796–805, Jul./Aug. 1998. [31] B. A. Welchko, T. A. Lipo, T. M. Jahns, and S. E. Schulz, “Fault tolerant three-phase AC motor drive topologies: A comparison of features, cost, and limitations,” IEEE Trans. Power Electron., vol. 19, no. 4, pp. 1108–1116, Jul. 2004. [32] D. Kastha and B. K. Bose, “Investigation of fault modes of voltagefed inverter system for induction motor drive,” IEEE Trans. Ind. Appl., vol. 30, no. 4, pp. 1028–1038, Jul./Aug. 1994. [33] M. S. Towers and P. Mawby, “Self protected light triggered thyristors,” in Proc. IEE Colloq. Recent Advances Power Devices, 1999, pp. 8/1–810.
Eduardo P. Wiechmann (S’81–M’86–SM’94) received the E.E. degree from Santa Maria University, Valparaiso, Chile, in 1975, and the Ph.D. degree in electrical engineering from Concordia University, Montreal, P.Q., Canada in 1985. Since 1976, he has been with the University of Concepción, Chile. He is now a Professor in the Electrical Engineering Department of the University of Concepción. His research interests are power converters, high-current rectifiers, ac drives, UPS systems, harmonics, and power factor control in industrial power distribution systems. His industrial experience includes more than 6000 hours in engineering projects and consulting. Dr. Wiechmann has published numerous technical papers and has coauthored technical books. He received the year 2000 Concepción City Award for Outstanding Achievements in Applied Research.
Pablo Aqueveque (S’05) was born in Santiago, Chile, in 1976. He received the B.S. and E.E. degrees from the University of Concepción, Chile, in 2000 and 2002, respectively. He is currently working toward the Ph.D. degree in power electronics at the University of Concepción. His research interests include high-current rectifiers, power converters, and modern digital devices.
Rolando Burgos (S’96–M’03) received the B.S., E.E., M.S., and Ph.D. degrees from the University of Concepción, Chile, in 1995, 1997, 1999, and 2002, respectively. In 2002, he joined the Center for Power Electronics Systems (CPES) at Virginia Polytechnic Institute and State University, Blacksburg, VA, as a Postdoctoral Fellow. Since 2005 he has been Research Assistant Professor in the Bradley Department of Electrical and Computer Engineering, Center for Power Electronics Systems. His research interests include multiphase power conversion, stability of ac power electronics systems, hierarchical modeling, control theory, and the synthesis of power electronics conversion systems for sea, air, and land vehicular applications. He is Secretary of the Committee on Simulation, Modeling, and Control of the IEEE Power Electronics Society.
José Rodríguez (S’81–M’83–SM’94) received the Eng. and Dr.-Ing degrees from the Universidad Técnica Federico Santa Maria in Valparaíso, Chile, and the University of Erlangen in Germany in 1977 and 1985, respectively, both in electrical engineering. Since 1977, he has been a professor at the University Federico Santa María. From 2001 to 2004 he was Director of the Electronics Engineering Department. From 2004 to 2005 he was Vice-Rector of Academic Affairs, and in 2005 he was elected Rector at the same university, a position he holds to this day. During his sabbatical leave in 1996, he was responsible for the mining division of the Siemens Corporation in Chile. He has a large consulting experience in the mining industry, especially in the application of large drives like cycloconverter-fed synchronous motors for SAG mills, high-power conveyors, controlled ac drives for shovels, and power quality issues. His main research interests include multilevel inverters, new converter topologies, and adjustable speed drives. He has directed over 40 R&D projects in the field of industrial electronics, he has coauthored over 50 journal and 130 conference papers. His research group has been recognized as one of the two centers of excellence in engineering in Chile in 2005 and 2006.
