Passive integration and RF MEMS: a toolkit for adaptive LC circuits Th.G.S.M. Rijksa, J.T.M. van Beeka, M.J.E. Ulenaersa, J. De Costerb, R. Puersb, A. den Dekkerc, and L. van Teeffelenc a
Philips Research, Prof. Holstlaan 4, 5656 AA Eindhoven, The Netherlands Tel: +31-402743748. Fax: +31-402743352. E-mail:
[email protected] b K.U. Leuven, Dept. of Electrical Engineering ESAT-MICAS, Kasteelpark Arenberg 10, B-3001 Leuven, Belgium c Philips Semiconductors, Gerstweg 2, 6534 AE Nijmegen, The Netherlands
Abstract: RF MEMS variable capacitors have been realized in an industrialized thin-film process for manufacturing highquality inductors and capacitors on high-ohmic silicon. The fixed as well as the moveable electrode consists of aluminium, the native aluminium oxide is used as a dielectric. A tuning ratio of 1.35 and a switching ratio up to 29 have been measured. At RF frequencies they show low ohmic losses, which makes them very suitable for use in high-quality adaptive LC networks. The feasibility of the application of switched capacitors for load switching in GSM power amplifiers has been demonstrated through simulations based on experimentally derived components and system parameters. 1.
Introduction
In RF transceiver front ends for wireless mobile communication accurate and high-quality passive circuits are of prime importance, especially in the power amplifier section, where high power levels are being handled. The efficiency of a power amplifier is to a large extent determined by the performance of the impedance matching network used to transform the low-ohmic output impedance of the power amplifier to the highohmic antenna impedance. Overall power-added efficiency is gained when these matching networks can be made adaptive to the various operational conditions such as output power, frequency, and antenna impedance. This can be achieved by incorporating highquality tuning or switching components in the impedance matching circuits. RF MEMS variable capacitors and switches are very promising as tuning or switching components due to their superior characteristics in terms of insertion loss, power consumption, and linearity as compared to their semiconductor counterparts. They offer the opportunity to realize adaptive RF circuits such as tunable band pass filters and VCO tank circuits [1]. In this paper we present RF MEMS variable capacitors as part of a passive integration technology, and demonstrate the feasibility of a MEMS-based adaptive impedance matching network for a GSM power amplifier module.
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Fig. 1: Cross-section of a RF MEMS variable capacitor in PASSITM technology 2.
Fabrication of RF MEMS
As a technological and functional platform for the development of RF MEMS an industrialized low-cost process for passive integration is used: the Philips’ PASSITM process. This thin-film passive integration technology on high-ohmic silicon combines three metal layers and two dielectric layers in five mask steps to form integrated inductors and capacitors [2,3]. For the realization of RF MEMS, the standard process is slightly modified and extended with surface micro-machining as a back-end module. The resulting process turns out to be a very simple approach to manufacture RF MEMS switches and variable capacitors, and high-quality inductors and fixed capacitors on the same die. A cross-section of a RF MEMS variable capacitor in PASSITM is shown in Fig. 1, using two of the available metal layers. The substrate is high-ohmic silicon (ρ > 5 kΩcm) in order to suppress RF losses in the substrate. The bottom electrode consists of 0.5 µm aluminium. The top electrode, which is also used as the structural layer, consists of 5 µm aluminium. Silicon nitride (SixNy) is used as a sacrificial layer in order to create an air gap of 1.5 µm between the top and bottom electrode. In Fig. 2 a scanning electron microscopy (SEM) photograph of a RF MEMS variable capacitor is shown. The top electrode is suspended 1.5 µm over the (fixed) bottom electrode. The air gap is clearly visible in the close-up in Fig. 2. The native aluminium oxide layer that is formed on the surface of both electrodes after removal
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Fig. 2: SEM image of a RF MEMS variable capacitor. The close-up clearly shows the air gap between top and bottom electrode.
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RF MEMS characterization
A DC voltage between top and bottom electrode, superimposed on the RF signal, is used to tune the height of the air gap, therewith tuning the effective capacitance. For this type of electrostatically-driven variable capacitors the pull-in phenomenon is well known [4]. Below the pull-in voltage continuous tuning of the capacitance can be achieved with a tuning factor of 1.5. However, this is only valid for a perfectly flat and rigid suspended top electrode. In practice, lower tuning ratios are measured due to deformation of the suspended electrode. If the actuation voltage exceeds the pull-in voltage, the suspended top electrode collapses on the bottom, i.e., the air gap virtually reduces to zero, resulting in a large increase of the capacitance. In this mode, the MEMS capacitor is used as a switched capacitor (or switch). Experimental results of the capacitance versus the actuation voltage of a MEMS capacitor, designed as in Fig. 2, are shown in Fig. 3. The electrode area was 475×475 µm2. The capacitance is derived from impedance measurements at 4 MHz, using a HP4275A LCR meter. The pull-in voltage is 4.7 V and is determined by the electrode area, the initial air gap, and the spring constant of the suspension springs. In Fig. 3 (top) the arrows indicate the sweep direction of the actuation voltage. The different curves for sweeping the actuation voltage upward and downward are well-known for parallel-plate capacitors and are caused by the fact that the electrostatic force is quadratic in the gap height whereas the spring force scales linearly with the gap height. The capacitance measured at 0 V (OFF state) is 2.3 pF, considerably higher than the calculated value of 1.23 pF. This is due to a parasitic RC path through the substrate, in parallel with the MEMS capacitor, as illustrated in Fig. 4 [5]. This parasitic capacitance only plays a significant role at frequencies below typically 100 MHz, i.e., when the impedance of the substrate path is smaller
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Fig. 4: Equivalent circuit of a MEMS capacitor with capacitance C and series resistance R in PASSITM.Csub and Rsub are the substrate capacitance and resistance, respectively. than that of the MEMS capacitor. This is confirmed by the RF measurements in Fig. 5, performed on a similar capacitor. At 1 GHz an OFF capacitance of 1.33 pF has been measured. In the ON state the capacitance is 27.8 pF and the influence of the substrate capacitance is negligible. The ON capacitance in Fig. 3 is 21.1 pF, resulting in a capacitance density of 107 pF/mm2. It is demonstrated here that the native aluminium oxide can be a stable dielectric for aluminium based MEMS capacitors. It has been found to break down at a DC voltage of 10-12 V. Capacitance densities up to 374 pF/mm2 have been measured in this type of devices. This can be recalculated to a remaining air gap of 24 nm between the electrodes in the ON state. With atomic-force microscopy an average roughness of 8-10 nm has been measured on each of the contacting surfaces, which can account for an effective air gap of 16-20 nm. In addition,
impedance matching circuits. For comparison, the ESR of an integrated fixed capacitor in series with a PIN switch diode is about 2.2 Ω in the ON state.
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Fig.5: Capacitance (top) and equivalent series resistance (bottom) versus frequency in the OFF (0 V) and ON (8 V) state as mentioned before, the suspended top electrode may be not completely flat, resulting locally in a larger air gap. The switching ratio is defined as the quotient of the capacitance in the ON and OFF state, respectively. Switching ratios as high as 29 have been measured on the devices under investigation. Fig. 3 (bottom) shows the capacitance for actuation voltages smaller than the pull-in voltage. Continuous tuning of the capacitance is demonstrated, with a tuning ratio of 1.35 (after correction for the substrate capacitance). In Fig. 5 the RF performance of a similar MEMS capacitor is shown, derived from a measurement of the scattering parameters using a HP8753D network analyzer. Fig. 5 (top) shows the capacitance versus frequency and has been discussed already. The onset of self-resonance of the MEMS capacitor with the inductance of the contact leads is clearly visible. It should be remarked that the present designs have not been optimized for a low series inductance. In Fig. 5 (bottom) the equivalent series resistance (ESR) of the MEMS capacitor is shown. The substrate resistance, due to the finite resistivity of the silicon substrate, plays a significant role in the ESR, especially in the OFF state when the capacitance is small [5]. The ESR at 1 GHz is 1.5 Ω in the OFF state (corresponding to a parallel loss resistance of 9.6 kΩ) but only 0.4 Ω in the ON state. The latter makes MEMS capacitors very suitable for application in adaptive LC networks such as adaptive
Adaptive impedance matching
In a RF power amplifier (PA) module for wireless mobile communication, the impedance matching circuit that transforms the antenna impedance to the optimal load impedance of the final stage transistor is very important for the overall power added efficiency of the PA. To meet the application specifications, the RF losses in the matching circuit should be low and a substantial second and third harmonic suppression of the non-linear power transistor should be realized [6]. The topology of the matching circuit is shown in Fig. 6 (top). Zload is the optimal load impedance of the final stage transistor, the antenna impedance is 50 Ω. Fig. 6 (bottom) shows an implementation of this circuit in passive integration for the GSM band at 900 MHz. The capacitors and the small inductors are integrated on silicon in PASSITM while the larger inductor is integrated in the low-temperature cofired ceramic (LTCC) substrate, that acts as a carrier for the module [3]. The optimal load impedance Zload is dependent on the transmitted power. From load-pull measurements at 900 MHz impedance levels of 2 Ω at 3.7 W (36 dBm) and 4+3j Ω at 1 W (30 dBm) have been derived. Fig. 7 shows the calculated RF transmission through the matching network for the two power levels, based on a measurement of the scattering matrix of the impedance matching network and the load-pull measurements mentioned above. The fixed matching circuit has been optimized for operation at 3.7 W. The return loss at 900 MHz is 21.2 dB, the insertion loss of 0.8 dB is mainly due to ohmic losses. It performs poorly at 1W, showing an insertion loss of 2.8 dB (return loss = 4.4 dB), clearly indicating the mismatch between the offered and desired Zload. It will be shown now that impedance transformation at 1 W can be improved using adaptive impedance matching (also called load switching) with a RF MEMS switched capacitor.
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Fig. 6: (top) Topology of the impedance matching circuit and (bottom) implementation with passive integration applying a partitioning between PASSITM and LTCC
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Frequency (GHz) Fig. 7: RF transmission through the impedance matching circuit of Fig. 6, at two different power levels An equivalent circuit model of the matching network has been built, including component parasitics and bond wires. Agilent’s Advanced Design System (ADS) has been used for circuit simulations. The simulations have shown that the impedance transformation at 1 W output power can be improved considerably when increasing the capacitance of the final matching stage (close to the antenna) by 4 pF. This can be achieved by connecting a RF MEMS capacitor in parallel with the fixed capacitor of the final matching stage, yielding a Zload of 4.6-0.1j Ω when the capacitor is switched ON. The capacitance value of fixed capacitor has to be decreased slightly in order to compensate for the OFF capacitance of the MEMS capacitor. Fig. 8 shows the transmission curves of Fig. 7 together with the results of the optimized matching at 1 W from the simulations. An RF MEMS capacitor with an OFF capacitance of 0.2 pF and a parallel loss resistance of 9.6 kΩ, and an ON capacitance of 4 pF together with an ESR of 0.4 Ω has been used. The lower insertion loss at 900 MHz and a return loss of 8 dB when the MEMS capacitor is switched ON demonstrates that Zload is closer to the desired impedance. With more elaborate adaptive matching circuit
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topologies an even better matching can be achieved. Important is also the very low power consumption of electrostatically driven RF MEMS (only typically 1 nJ per switching cycle). Furthermore, the MEMS-switched matching network in PASSITM is a fully integrated solution. Experimental verification of the adaptive impedance matching concept in terms of amplifier efficiency is underway. For comparison, the same calculation has been performed using a 4 pF fixed capacitor with an ESR of 0.15 Ω together with a PIN switch diode (Philips BAP64-2) with an OFF resistance of about 4 kΩ and an ON resistance of 2 Ω. The high ON resistance results in a higher insertion loss at 900 MHz and less higherharmonics suppression. The PIN diode requires a drive current of 10 mA in the ON state, the power consumption is 7 mW. 5.
Conclusions
We have demonstrated that RF MEMS variable capacitors can be manufactured on silicon using a simple thin-film process designed for fabricating inductors and capacitors (PASSITM). The fixed as well as the moveable electrode consists of aluminium, the native aluminium oxide is used as a dielectric. For these MEMS capacitors a tuning ratio of 1.35 and a switching ratio up to 29 has been measured. At RF frequencies they show low ohmic losses, which makes them very suitable for use in highquality adaptive LC networks. Their application for load switching in adaptive impedance matching circuits for GSM power amplifiers has been demonstrated through simulations based on experimentally-derived component and system parameters. 6.
Acknowledgments
The authors acknowledge the valuable contributions from H. Nulens and J. Dijkhuis. This work has been carried out as part of the IST project ‘MEMS2TUNE’. [1] Harrie A.C. Tilmans, Walter De Raedt, and Eric Beyne, Proceedings MME ’02 Workshop, Sinaia Romania, Oct. 6-8 2002 [2] Nick Pulsford, RF Design Nov. 2002, 40-48. [3] J.T.M van Beek, M.W.H.M. van Delden, A.B.M Jansman, A. Boogaard, and A. Kemmeren, IMAPS 2001, Baltimore USA, Oct. 9-11 2001, 467-470. [4] Harrie A.C. Tilmans, EUROSENSORS XVI, Prague Czech Republic, 15-18 Sept. 2002 (on CD ROM only). [5] A.B.M. Jansman, J.T.M. van Beek, M.H.W.M. van Delden, A.L.A.M. Kemmeren, and A. den Dekker, Proceedings of European Conference on Wireless Technologies 2002, EWCT September 2002, Milan, 189-193 [6] N.J. Pulsford, J.T.M. van Beek, M.W.H.M van Delden, and A. Boogaard, 1999 IEEE MTT-S Digest, Anaheim USA, June 1999, 1897-1900
