Position monitoring of low intensity beams using a ...

Position monitoring of low intensity beams using a digital frequency down converter Hengjie Ma and Craig Drennan Citation: AIP Conf. Proc. 333, 349 (1995); doi: 10.1063/1.48061 View online: http://dx.doi.org/10.1063/1.48061 View Table of Contents: http://proceedings.aip.org/dbt/dbt.jsp?KEY=APCPCS&Volume=333&Issue=1 Published by the American Institute of Physics.

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Position Monitoring of Low Intensity Beams Using A Digital Frequency Down Converter HengjieMa, CraigDrennan Fermi National Accelerator Laboratory, PO Box 500, Batavia, IL Abstract In monitoring the position of very low intensity beams a signal processing scheme similar to that used in an amplitude-comparison monopulse radar may be employed. In this scheme, an I-Q demodulator for both the sum and difference channels and a phase detector are needed to detect the beam position. It is complex and costly to implement the signal processing with discrete analog components. However, a newly available HSP50016 Digital Down Converter (DDC) chip has provided an attractive alternative. This DSP chip processes the digitized output of the IF section by first converting the signal to baseband using an in-phase/quadrature mixer and then filtering the result with a combination of a programmable high decimating filter and a fixed FIR shaping filter. The accuracy of the quadrature demodulation, nearly ideal filter shape factor and filter reject-band attenuation make the DDC a favored choice over a discrete analog design in an application dealing with very weak beam signals.

INTRODUCTION It is desired that Beam Position Monitor (BPM) detectors, such as those used in the Switchyard area at Fermilab, be used in the Fixed Target Areas. Upcoming experiments such as KTEV and E-815 will be seeing higher intensity beam making this type of device feasible. The advantage of using a BPM is that the detector is non-intrusive. This means that material is not placed in the path of the beam causing secondary emissions which increase the uncertainty of the experiments' data. Even though the beam intensities in the Fixed Target area will be higher they are still in or just below the lowest range of the current Switchyard BPM system. There is a need to take position and intensity measurements for three cases; a fast spill of beam, 1012 protons over a 1.5 millisecond interval, a slow spill of 2xl012 protons over a 20 second interval, and a slow spill of beam, 1011 to 1012 protons over a 20 second interval Proposed here is a digital signal processing scheme to directly compute the difference-to-sum ratio (A/E). The digital scheme is aimed at meeting the required sensitivity, noise suppression and flexibility for the lower beam intensity case.

1995 American Institute of Physics

349

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350 Position Monitoring of Low Intensity Beams DESIGN MOTIVATION The functionality of a BPM is similar to that of a simple monopulse tracking radar which detects the azimuth of targets using signal amplitude comparison. Therefore, some of its signal processing schemes and circuit designs can be directly applied to our beam position monitor (1), (2), (3). The current signal processing scheme has evolved from investigation of the following issues.

Measurement Error Due to Channel Mismatch (4) The resonant beam detector currently used in Fermilab Switchyard has a receptive sensitivity of typically -120dBm for a slow spill of 1012 protons over 20 seconds. Therefore, the total system gain will have to be at least 120dB, and the bandwidth less than lOHz in order to get usable measurement data. For the conventional AM/PM BPM, increased gain and reduced bandwidth often result in the instability of instrument zero. To eliminate zero-drift due to channel mismatch the conventional signal processing scheme of one-step AM-to-PM conversion is modified into a two-step conversion of AM-to-A/Z followed by A/Zto-PM with the signal amplification, filtering and frequency down-converting done between the two conversions. Bias of Estimate Due to Noise (5)

In the Fixed Target area applications, the lower beam intensities will not provide good signal-to-noise ratios out of the detector. Noise, assumed to be a zero-mean random process, has a modulation effect on both the amplitude and phase of the BPM signals,(6),(7). Furthermore, noise in the two channels are statistically dependent to some degree due to imperfect channel isolation. With the phase detection method used to perform A-over-Z normalization in the AM/PM BPM, the obtained estimate contains a certain bias. This estimation bias is dependent on both beam position and intensity. Statistical analysis suggests that this dependence increases with the poorer signal-to-noise ratios associated with lower beam intensities. This particular estimation bias problem is avoided by handling the A-andZ directly using a quadrature mixer for down conversion and computing the Aover-Z ratio with a digital processor.

Advantages of Using Digital Signal Processing Methods While the cost of digital signal processors, dedicated function DSP devices, and analog-to-digital converters have dropped, their performance has

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H. Ma and C. Drennan 351

increased to the point where there is a clear advantage in using these digital devices in earlier stages of the electronics. Advantages of the digital I-Q detection method proposed here include the following.

a) The overall performance of the digital detection beyond the ADC is highly repeatable and is not subject to drift over time and temperature. b) Beyond the ADC the signal is not subject to reduction of the signal-to-noise ratio before the final filter due to the low noise floor and high spurious free dynamic range of the demodulator. c) The digital filter bandwidth can be manipulated remotely through a digital communications port. This aids in optimizing the tradeoffs between noise suppression requiring narrow bandwidths and observation of changes in beam position or beams of short duration requiring wide bandwidths. d) Filters can be realized with excellent specifications allowing in many cases relaxation of the specifications on the analog filter stages. e) Converting the signal information to base-band and decimating the stream of signal samples allows relaxation of the data processor speed and memory requirements. CIRCUITRY DESCRIPTION AND IMPLEMENTATION

Tunnel Unit Figure 1 is the block diagram of the analog RF front end for the direct A-Z BPM system currently being developed. The two plates of the resonant beam detector couple the time-harmonic fields generated by the passing beam bunches. The ratio of the two output signal amplitudes, A and B, is related to the displacement of the beam x by approximately 0.67x = 201og(A/B). There is a significant capacitive coupling between the two plates of the beam detector. To increase the detection sensitivity to beam displacement and increase the output, a tuning network is used (8). Cl, LI, and C2, L2 form the two pararell resonant tanks for plate A and plate B respectively. Their center frequency is 53.104 MHz. The unloaded Q is about 200, and the pararell resonant impedance is about 10 K ohm. This yields a bandwidth of 270 KHz. Trimer capacitor C3 decouples plates A and B. The impedance match between the resonant tanks and the 50 ohm input imdedance of the hybrid junction is accomplished through the taps on the coils. A 180° hybrid junction converts the two output signals from the detector into the difference and sum signals. The distance between the beam detector and the instrument rack is several hundred feet. There is certain amount of signal loss on the cables and noise pickup at the interconnections. To avoid the further SNR deterioration due to the long distance signal transmission, it may be necessary to place the two RF preamplifiers near the detector in the tunnel. MITEQ low-noise/60dB RF

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352 Position Monitoring of Low Intensity Beams

amplifier AU-4A-0110 was chosen for the pre amplification. Its survival under the radiation is still under the study. TunnenJnit_ I

Resonant Beam Detector

•(0

Figure 1. Block Diagram of the Tunnel Unit.

RF-to-IF down-converter

The RF-to-IF conversion and IF amplification are done with two NE615 VHP receiver chips. An input tuning network provides a preliminary noise/interference suppression and the impedance match between the 50-ohm cables and the mixer inputs. The conversion gain is about 13dB. The IF amplifier has a gain of 39dB. The insertion loss of the two stage ceramic IF filters is 17dB. Therefore, the net gain of the RF/IF stage is 35dB, and the bandwidth is about 5KHz. U5 and U6 boost the IF signal amplitude to a proper level for the digitization. Analog Front End '

NE615

f,F= 455 KHz

1st LO Input 52.649 MHz '

Figure 2. Analog Front End Block Diagram. PLL LO signal generator

During the flat top period of the beam spill, the Tevatron RF can vary slightly. In order to prevent the IF from drifting out of the IF filter pass band, the LO frequency tracks the Tevatron RF. This is accomplished with a phase-locked loop based LO signal generator. The phase-locked loop based LO signal generator unit provides not only the LO frequency for the analog down converter but also the clock for the digitizer

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H. Ma and C. Drennan 353

and the I-Q demodulator/filter in the DDC's. The IF reference frequency is derived from a crystal oscillator of 1.82 MHz (IF x 4). A crystal-controlled VCO generates the LO frequency, which is 53.104MHz - 455KHz =52.649MHz and varies within 50 ppm around the center frequency. The LO is mixed with Tevatron RF to generate the actual IF. The actual IF is compared with the IF reference frequency. The resultant error provides the base for adjusting the frequency of the VCO. The phase detection in the PLL is implemented with digital circuits. It has a frequency-tracking capability, which ensures an adequate frequency-capture range. To minimize the phase noise in the generated LO, the bandwidth of the loop filter is set to be less than 1KHz. Phase Lock Loop Reference Frequency Generator

i"**^ 4 REF.FREQ.-2ndLOx4 1.62MHz fe

LO^

i •in -il->

1.83 MHz REF.FREQ DISTRIBUTION

OAmp.

52.649MHz IstLO

*Sfc.

***

*

DISTRIBUTION LOOP GAIN

*^

LOOPRLTER

__iii pm

Figure 3. The PLL Reference Frequency Generator Block Diagram.

Digital IF Stage and IQ Detection The sum and difference signals out of the analog IF stages are scaled for input to the +/- 2.5V input of the Analog to Digital Converters (ADC's) using the dual-channel variable-gain amplifier, U7. The gain for each channel is variable from 0 to +41 dB and is controlled by the output of the data processor.. The proposed circuit will use an HI5800 12 bit, 3 MSPS sampling ADC. The device will be operated at 1.82 MSPS,(9). The Digital Down Converter (DDC) is a single chip synthesizer, quadrature mixer and high decimation lowpass filter. The 12 bit sampled data stream from the ADC is set into the 12 most significant bits of the DDC's 16 bit input. The quadrature mixer down converts the amplitude information at the IF frequency to baseband (DC). The complex result is lowpass filtered and decimated with identical real filters in the in-phase (I) and quadrature (Q) processing chains. Lowpass filtering is accomplished via a five stage high decimation filter (HDF) followed by a fixed finite impulse response (FIR) filter. The combined response of the HDF and FIR results in a -3dB to -102dB shape

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354 Position Monitoring of Low Intensity Beams

factor of better than 1.5. The stop band attenuation is greater than 106dB. The composite pass band ripple is less than 0.04dB,(10). The end-to-end noise floor of the HSP50016 is greater than 100 dB below full scale and the spurious free dynamic range (SFDR) of the internal modulation process is greater than 102 dB. The frequency selectivity of the DDC is less than 0.00032 Hz at 2.73 MSPS. The combination of down conversion and decimation is an efficient way to handle a bandpass signal centered at the IF frequency,(ll). Decimation is the process of lowpass filtering the signal that was translated to DC and re-sampling the signal at a lower sampling rate. By using multiple stages of filtering and decimation, filters with sharper frequency cutoffs can be realized with fewer multiplication's. Also, decimation serves to reduce the data rate that a microprocessor or microcontroller must handle. Digitizer jindJDigital I-Q Demodulator^ HI5800

Sample Clock Fs= 1.82MHz

Figure 4. Digital IF and IQ Detection Block Diagram.

There is a direct relationship between the decimation rate for which the HSP50016 is set and the resulting bandwidth of the lowpass filter response. This relationship is -3dB BW - 0.13957 Fs / R and -102dB BW - 0.19903 Fs / R, where Fs is the input sampling rate and R is the HDF stage decimation factor.

The Data Processing Section The I data and Q data are output from the HSP50016 with a synchronous serial connection directly to the data processor or to a FIFO buffer. The processor would also need to setup the Control Words of the DDC. These seven 40 bit control words determine the operation of the down converter, the decimation rate and filter bandwidth, and the input and output formats.

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H. Ma and C. Drennan 355 CONCLUSIONS Experimenters in the Fixed Target Area wish to take advantage of the nonintrusive beam position and intensity measurements provided by the BPM's like those currently used in Switchyard. However, the current AM/PM method in use now is not expected to operate at the lower beam intensities and poorer signal-tonoise ratios in these areas. This is because of the instability of the bias of the signal estimates. The direct A/Z method involves only a linear conversion and does not have this estimate bias problem. Therefore, averaging more signals together or decreasing filter bandwidth should produce better estimates of the signal mean. The availability of less expensive analog-to-digital converters with higher accuracies and higher sample rates allow digitization of signals at the standard IF frequencies. The HSP50016 allows accurate and efficient handling of the stream of digital values representing the signal. The magnitude and relative phase of the A and Z signals is preserved in the I-Q demodulation process and the ratio A/E can be computed in the data processor or at a higher level of the data processing system to determine beam position. REFERENCES 1. Kahana, E. and Chung, Y., "Test Results of a Monopulse Beam Position Monitor for the Advanced Photon Source," Proceeding of Accelerator Instrumentation Fourth Annual Workshop, Berkeley, CA 1992, pp. 271- 278. 2. Ma, H. and Moore, C., " A Beam Position Monitor for Low Intensity Beams," Proceeding of Accelerator Instrumentation Workshop, Santa Fe, NM, 1993. 3. Skolnik, M.I., Introduction to Radar Systems. 2nd Ed., McGraw-Hill, 1980. 4. Ma, H., "Report III on Switchyard Beam Position Monitor," Fermilab-TM1904, January 1993. 5. Ma, H., "The Effect of Beam Intensity on the Estimation Bias of Beam Position," Fermilab-TM-1905, December 1991. 6. Shanmugan, K.S., Breipohl, A.M., Random Signals: Detection. Estimation and Data Analysis. John Wiley & Sons, 1988, New York. 7. Shanmugan, K.S., Digital and analog communication systems. John Wiley & Sons, 1979, New York. 8. Kerns, Q., et. al., " Tuned Beam Position Detector for the Fermilab Switchyard," PAC 1987, Vol. 1, pp. 661-663. 9. HI5800 Data Sheet, Harris Semiconductor, FN 2938.4, December 1993. 10. HSP50016 Data Sheet, Harris Semiconductor, FN 3288.3, June 1994. 11. Crochiere, R.E. and Rabiner, L.R., "Multirate Digital Signal Processing", Prentice Hall, 1983.

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